JPH0258861B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a high frequency half bridge type switching circuit. In recent years, low-voltage, high-current power supplies have been required for electronic computers, their peripheral equipment, and general communications equipment, and switching power supplies, which have the advantages of being small, lightweight, and highly efficient, have been widely used. Conventionally, various circuits have been devised as switching circuits using switching elements such as bipolar transistors for the switching power supply. Among these, so-called single-ended circuits that use one switching element have a simple circuit configuration and are suitable for miniaturization, but as the output power increases, a large current element is required as the switching element, and the power Since direct current is superimposed on the transmission transformer, the transformer is inefficiently utilized and can only be used in switching type power supplies with relatively low output power. In addition, a bridge-type switching circuit that uses four switching elements is suitable for switching power supplies with high output power, but for low output power, as many as four transistors are used, which increases the size of the power supply. , is not suitable because the circuit configuration is complicated. However, bush-pull type and half-bridge type switching circuits that use two switching elements have simpler circuit configurations than bridge type switching circuits, and the use of the transformer is more efficient because direct current is not superimposed on the power transmission transformer. If a switching element is used, twice the output power of a single-ended type can be obtained, so it can be used for switching type power supplies ranging from low output power to high output power. However, the push-pull type circuit has a drawback in that a voltage twice the input voltage is applied between both ends of the switching element, such as the collector-emitter of a bipolar transistor, so a switching element with a high withstand voltage is required. Furthermore, in order to make switching type power supplies more compact and to improve response speed to load fluctuations and input fluctuations, it is desired to increase the frequency of switching circuits. However, in the above conventional circuit, as the switching frequency is increased, the time required for the rise and fall of the voltage across the switching element and the current flowing through the switching element cannot be ignored compared to the switching cycle. The larger the loss, the more the so-called switching loss increases. This is unavoidable due to the characteristics of switching elements such as bipolar transistors, and due to this increased switching loss, the power transfer efficiency of switching power supplies deteriorates, and a large heat sink is required to dissipate the heat generated by this loss. The advantages of a switching power supply are lost, such as the size of the power supply becoming larger.
It was practically impossible to increase the switching frequency above 50KHz to 100KHz (when using bipolar transistors). Additionally, noise generated from the switching circuit affects the load, which has been a problem. The present invention has been made in view of the above points, and its object is to provide a half-bridge type high frequency switching circuit which has extremely low switching loss and low noise even when the switching frequency is approximately 100 KHz or higher. The present invention will be explained below with reference to the drawings. FIG. 1 shows the basic configuration of a conventional half-bridge type switching circuit, and FIG. 2 shows the basic configuration of a switching circuit according to an embodiment of the present invention. Capacitors 13 and 14 are connected in parallel to both ends of each switching element 3 and 4, and switching is performed using the inductance seen from the primary side of the transformer 5, the inductance of the chiyoke coil 6, and the resonance operation of the capacitors 13 and 14. By preventing the voltage across elements 3 and 4 from changing rapidly compared to the switching period, the time during which the voltage and the current flowing through the switching element overlap is extremely shortened, and even when the switching frequency becomes high, This is a half-bridge type switching circuit that can extremely reduce switching loss. The switching circuit shown in FIG. 2 will be explained in detail below. As shown in the figure, two input DC power supplies 1 and 2 are connected in series, and a series connection circuit of two switching elements 3 and 4 made of, for example, bipolar transistors is connected in parallel to both ends of the input DC power supplies 1 and 2. Point to power transmission transformer 5
The respective one ends of the primary windings are connected to each other. Resonant capacitors 13 and 14 and damper diodes 11 and 12 are connected in parallel to both ends of switching elements 3 and 4, respectively, in the direction shown. Further, the anodes of rectifying diodes 7 and 8 are connected to both ends of the secondary winding of the transformer 5 , respectively, the cathodes of the diodes 7 and 8 are both connected to one end of the chiyoke coil 6, and the cathodes of the diodes 7 and 8 are connected to one end of the chiyoke coil 6. One end is connected to the midpoint of the secondary winding of the transformer 5 via a smoothing capacitor 9. Further, both ends of the capacitor 9 are connected in parallel to a load end 10, respectively. In addition, control terminals of switching elements 3 and 4 (not shown), such as bipolar
A drive circuit for turning on or off the switching element is connected to the base of the transistor. In addition, in order to make the explanation of the circuit operation below easier to understand, the voltages of the input DC power supplies 1 and 2 are assumed to be the same voltage and are respectively referred to as Ein, the capacitance values of the resonance capacitors 13 and 14 are also assumed to be the same and are respectively referred to as C, and the transformer 5 The number of turns from one end to the midpoint of the secondary winding of is N ₂ , and the number of turns of the primary winding is N ₁
shall be. Furthermore, the switching elements 3 and 4 and the diodes 7, 8, 11, and 12 are assumed to be ideal switches. Next, the capacitance value C of the resonance capacitors 13 and 14, the inductance value L ₁ seen from the primary side of the transformer 5,
The value L ₂ is obtained by multiplying the inductance value of the chiyoke coil 6 by the square of the turns ratio of the transformer 5 (N1/N2) ² , and the constant DC output voltage generated across the capacitor 9 after rectification is applied to the windings of the transformer 5. Setting conditions such as the value Eout multiplied by the linear ratio N1/N2 are such that the resonance frequency of C and _L1 is approximately matched to the switching frequency of the switching elements 3 and 4. Further, C is selected to be sufficiently larger than the value of the stray capacitance equivalently existing between both ends of the switching elements 3 and 4. Make Eout smaller than Ein. Also, L ₂ is
Choose around 0.1 to 10 times L ₁ . The circuit operation of this embodiment will be explained based on the above setting conditions. 3a to 3h are equivalent circuit diagrams showing each section of one cycle of the circuit operation of the embodiment shown in FIG. 2. In FIG. In addition, in FIGS. 3a to 3h, switching elements 3, 4 and diodes 7, 8, 11, 12
is shown as an ideal switch. Furthermore, capacitor 9 is shown as an ideal battery. Also, the fourth
Figures a to j are voltage and current waveform diagrams of important parts of the circuit operation of this embodiment, where a is a waveform diagram of the voltage i ₃ flowing through the switching element 3, and b is a diagram of the voltage v ₃ across the switching element 3. c is a waveform diagram of the current i ₁₁ flowing through the damper diode 11,
d is a waveform diagram of the current i ₄ flowing through the switching element 4, e is a waveform diagram of the voltage v ₄ across the switching element 4, f is a waveform diagram of the current i ₁₂ flowing through the damper diode 12, and g is a power transmission transformer. A waveform diagram of the current i ₁ flowing through the inductance seen from the primary side of the transformer 5 , h is a waveform diagram of the current i 7 which is the current flowing through the output rectifying diode 7 multiplied by the turns ratio N2/ _N1 of the transformer 5 , i is the waveform diagram of current i8, which is the current flowing through the output rectifying diode ₈ multiplied by the turns ratio N2/N1 of the transformer 5, and j is the waveform diagram of the current flowing through the chiyoke coil 6 multiplied by the turns ratio N2/N1 of the transformer 5. 2 is a waveform diagram of current i ₂ multiplied by . The equivalent circuit for the section t=0 _{to t1} shown in FIG. 4 a to j is shown in _FIG .
Figure b, the section from t = t ₂ to _{t 3} is shown in Figure 3 c, t = t ₃ to _{t 4}
The section from t= _t4 to _t5 is shown in FIG. 3e, the section from t= _t5 to _t6 is shown in FIG. 3f, and the section from t= _t6 to _t7 is shown in FIG. , t= _t7 to _t8 correspond to FIG. 3h, respectively. Then, the cycle begins at t=0, and t=
One cycle ends at t ₈ . Next, from t=0 to t=t ₈
I will explain step by step. First, the initial conditions of the circuit at t=0 are as shown in FIGS. [ _V ], the capacitor 14 is charged to 2×Ein [ _V ], and i ₁ =−i ₂ . For the following explanation, for example, i ₃ at t=t ₁ is i ₃
(t ₁ ), and v ₄ at t=t ₃ is v ₄ (t ₈ ). In FIG. 2, the interval t=0 to _t1 is the interval in which the switching element 3 is in a conductive state and the voltage Ein [ _V ] of the power supply 1 is applied to the primary winding of the transformer 5 . In addition, the capacitor 14 is charged with a voltage of 2×Ein [ _V ], and the voltage Ein [ _V ] of the power supply 2 and the transformer 5
In this section, the closed circuit consisting of the power supply 2, the primary winding of the transformer 5, and the capacitor 14 can be ignored in order to maintain a state of equilibrium with the sum of the primary winding voltages Ein [ _V ]. Furthermore, since the switching element 4 and the damper diode 12 are always in a damped state, they are ignored. Further, a voltage is applied to the secondary winding of the transformer 5 according to the turns ratio, and the diode 7 is in a conductive state and the diode 8 is in a slightly off state due to the above-mentioned initial conditions. From the above points, the equivalent circuit shown in FIG. 3a is obtained. At this time, v ₃ =0 [ _V ] v ₄ =2×Ein [ _V ] and does not change, and i ₁ and i ₂ change according to the following equation. i ₁ = −i ₂ (t ₀ ) + Ein/L ₁ (tθ − t ₀ ) i ₂ = i ₂ (t ₀ ) + Ein − Eout/L ₂ (tθ − t ₀ ) tθ can be any value between 0 and t ₁ time t0=0 Also, from FIG. 3a, the current i ₃ flowing through the switching element 3 is i ₃ =i ₁ +i ₂ . Further, the time t ₁ in this section is arbitrarily changed from the control terminal of the switching element 3. During the interval t= _t1 to _t2 , the switching element 3 is briefly disconnected, and the capacitor 13 is disconnected due to the resonance phenomenon between the inductance seen from the primary side of the transformer 5 , the inductance of the choke coil 6, and the capacitors 13 and 14.
is slowly charged, the capacitor 14 is slowly discharged, and from v ₃ = 0 [ _V ] v ₄ = 2 × Ein [ _V ], v ₃ =
This is the period in which it changes up to v ₄ = Ein [ _V ]. This section includes switching elements 3, 4, diodes 8, 1
1 and 12 are all in the damped state and can therefore be ignored, resulting in the equivalent circuit shown in FIG. 3b. The time t ₂ −t ₁ in this interval is obtained by the following equation. Also, v ₃ , v ₄ , i ₁ , and i ₂ change according to the following formula: v ₃ = i ₁ (t ₁ ) + i ₂ (t ₁ )/2Cωsin(ω(tθ−t ₁ ))
+Ein(1−Cos(ω(tθ−t ₁ ))) EoutL ₁ /L ₁ +L ₂ (1−Cos(ω(tθ−t ₁ ))) v ₄ =2Ein−v ₃ i ₁ =i ₁ (t ₁ ) − (i ₁ (t ₁ ) + i ₂ (t ₁ )) L ₂ /L ₁ +L ₂ (
1−cos(ω(tθ−t ₁ ))) +(Ein/ωL ₁ −Eout/2Cω) sin(ω(tθ−t ₁ )
) + Eout / L ₁ + L ₂ (tθ−t ₁ ) i ₂ = i ₂ (t ₁ ) − (i ₁ (t ₁ ) + i ₂ (t ₁ )) L ₁ /L ₁ + L ₂ (
1−cos(ω(tθ−t ₁ ))) +(Ein/ωL ₂ −L ₁ Eout/ωL ₂ (L ₁ +L ₂ )) sin(ω
(tθ− _t1 ))−1/ _L1 + _L2Eout (tθ− _t1 ) tθ is any time between _t1 and _t2 . Therefore, at t= _t2 , v3=v4=Ein and transformer 5 The voltage across the primary winding becomes 0 [ _V ], and the voltage generated in the secondary winding also becomes 0 [ _V ], so the diode 7 is soon cut off, but the inductance of the chiyoke coil 6 has a certain finite value. Since a current flows through the diode 8 during the next period, the current flows through the diode 8. The interval t = t ₂ - _{t 3} is the same as the interval t = t ₁ - _{t 2} , where the inductance seen from the primary side of the transformer 5, the inductance of the chiyoke coil 6, and the capacitor 1
Due to the resonance phenomenon of 3 and 14, the capacitor 13 is slowly charged and the capacitor 14 is slowly discharged, and from v ₃ = v ₄ = Ein [ _V ], v ₃ = 2 × Ein [ _V ], v ₄ =
This is the period during which the voltage changes up to 0 [ _V ]. In this section, the voltage of the primary winding of the transformer 5 is reversed from that in the section t= _t1 to _t2 , so the diode 7 is cut off and the diode 8 is turned on. Furthermore, the switching elements 3 and 4 and the diodes 11 and 12 are always in a depleted state. Thus, the equivalent circuit shown in FIG. 3c is obtained.
The time t ₃ to _{t 2} in this section is obtained by the following equation. Also, v ₃ , v ₄ , i ₁ , and i ₂ change according to the following formula: v ₃ = i ₁ (t ₂ )−i ₂ (t ₂ )/2Cωsin(ω(tθ−t ₂ ))
+L ₁ /L ₁ +L ₂ (1-cos (ω(tθ-t ₂ ))) Eout + Ein v ₄ = 2Ein-v ₃ i ₁ = (L ₁ /L ₁ +L ₂ +L ₂ /L ₁ +L ₂ cos (ω (tθ− _t2 ))
) i ₁ (t ₂ ) + L ₂ /L ₁ + L ₂ (1−cos(ω(tθ−t ₂ )))
i ₂ (t ₂ ) Eout/L ₁ +L ₂ (tθ−t ₂ )+Eout/ω(L ₁ +L ₂ ) sin
(ω(tθ− _t2 )) _i2 = _L1 / _L1 + _L2 ( _i1 ( _t2 ) _−i2 ( _t2 ))(1−cos(
ω(tθ− _t2 )))+ _i2 ( _t2 )−Eout/ _L1 + _L2 (tθ− _t2 ) _−L1 / _ωL2 ( _L1 + _L2 )E
out sin(ω(tθ−t ₂ )) tθ is any time from t ₂ to t ₃ Therefore, at t = t ₃ , v ₃ = 2Ein [ _V ], v ₄ = 0 [ _V ], but i ₁ and i ₂ still remains, which is v ₃ 2Ein [ _V ]
As described above, an attempt is made to charge _v4 to below 0 [ _V ], but the damper diode 12 is positively biased and becomes conductive, and does not change thereafter. The interval t=t ₃ to _{t 4} is a period from when the current first flows through the damper diode 12 until the current reaches 0 [ _A ]. The diode 12 is conductive and a voltage of Ein [ _V ] is applied to the primary winding of the transformer 5 by the power supply ₂ in the opposite direction to the interval t=0 to t1. Also, the capacitor 13 has 2Ein [ _V ]
is charged, and the voltage of power supply 1 is Ein [ _V ]
In order to maintain a balanced state with the sum of the voltage Ein [ _V ] of the primary winding of the transformer 5 and the voltage Ein [V] of the primary winding of the transformer 5, in this section, the closed circuit consisting of the power supply 1, the primary winding of the transformer 5 , and the capacitor 13 is Can be ignored. Moreover, the switching element 3 and the diodes 11 and 7 are always in a damped state. Thus, the equivalent circuit shown in FIG. 3d is obtained. From Figure 3d, the current flowing through the diode 12 is i ₁
~i ₂ , so the current in diode 12 is 0 [ _A ]
This means that i ₁ = i ₂ . Therefore, the time t ₄ −t ₃ in this section can be obtained from the following equation. t ₄ − t ₃ = L ₁ L ₂ (i ₁ (t ₃ ) − i ₂ (t ₃ )/(L ₁ + L ₂ ) Ei
In this interval, v ₃ = 2Ein [ _V ] and v ₄ = 0 [ _V ], which do not change, and i _{1 and} i ₂ change according to the following equation. i ₁ = -Ein/L ₁ (tθ - t ₃ ) + i ₁ (t ₃ ) i ₂ = Ein - Eout/L ₂ (tθ - t ₃ ) + i ₂ (t ₃ ) tθ is any value between t ₃ and t ₄ During the period t= _t4 to _t5 , the switching element 4 is in a conductive state, and the voltage of the power supply 2 is applied to the primary winding of the transformer 5.
This is a section in which Ein[ _V ] is applied in the opposite direction to the section t= _t0 to _t1 . Also, in capacitor 13
The voltage of 2Fin [ _V ] remains charged, and for the same reason as in the other section t= _t3 to _t4 , the equivalent circuit shown in FIG. 3e is obtained. In the figure, at t= _t4 , the direction of i _L1 is reversed, but during this section it reverses to the direction shown. Further, the time t ₅ -t ₄ in this section is arbitrarily changed from the control terminal of the switching element 4. In this interval, v ₃ = 2Ein [ _V ] v ₄ = 0 [ _V ] and does not change, i ₁ ,
i ₂ and i ₃ change using the same formula as the interval t=0 to t ₁ when i ₁ is positive in the opposite direction to the interval t=0 to _{t 1} . however
tθ is any time between t ₄ and t ₅ , and t ₀ is replaced with t ₄ . In the section t= _t5 to _t6 , the switching element is suddenly disconnected, and due to the resonance phenomenon of the inductance seen on the primary side of the transformer 5, the inductance of the chiyoke coil 6, and the capacitors 13 and 14, the section t= _t1 to _t2 On the contrary, capacitor 13 is slowly discharged, capacitor 14 is slowly charged, and v3=Ein [ _V ], _v4
This is the period in which the voltage changes from = 0 [ _V ] to v ₃ = v ₄ = Ein [ _V ]. In this section, the switching elements 3 and 4 and the diodes 7, 11, and 12 are all in a damped state, so they can be ignored. Thus, the equivalent circuit shown in FIG. 3f is obtained. i ₁
If the opposite direction of the flow direction in the interval t=t ₁ to _{t 2} is positive, then the time t ₆ −t ₅ in this interval can be obtained from the same equation as the equation for t ₂ −t ₁ . Also, v ₃ is v ₄ in the interval t = t ₁ to _{t 2} ,
v ₄ is v ₃ in the interval t = t ₁ - _{t 2} , i ₁ and i ₂ are in the interval t = t ₁ - _{t 2}
It changes according to the same formula as . However, tθ in the formula is t ₅ to _{t 6}
Set it to any time in between, and replace t ₁ with t ₅ . Therefore, at t=t ₆ , v ₃ = v ₄ = Ein, the voltage across the primary winding of transformer 5 becomes 0 [ _V ], and the voltage generated in the secondary winding also becomes 0 [ _V ]. The diode 8 is soon cut off, but since current is flowing through the inductance of the chiyoke coil 6, the diode 8 becomes conductive the next moment, and a current flows. The interval t = t ₆ - _{t 7} is the same as the interval t = t ₅ - _{t 6} , and the inductance seen from the primary side of the transformer 5 , the inductance of the chiyoke coil 6, and the capacitor 1
3 and 14, the capacitor 13 is slowly discharged and the capacitor 14 is slowly charged, contrary to the interval t=t ₂ to _{t 3} , and from v ₃ = v ₄ = Ein [ _V ]
This is the period in which it changes from v ₃ = 0 [ _V ] to v ₄ = 2Ein [ _V ]. In this section, the voltage of the primary winding of the transformer 5 is reversed from that in the section t= _t5 to _t6 , so the diode 8 is cut off and the diode 7 is turned on. Also,
Switching elements 3, 4 and diodes 11, 12
is always in a state of decline. Thus, the equivalent circuit shown in FIG. 3g is obtained. The flow direction of i ₁ is defined as the interval t=t ₂ ~
If the opposite direction of t ₃ is positive, then the time t ₇ − of this interval
t ₆ is obtained from the same formula as t ₃ −t ₂ . Also, v ₃ is v ₄ in the interval t = t ₂ - _{t 3} , and v ₄ is v ₃ , i ₁ in the interval t = t ₂ - t ₃ .
and i ₂ change according to the same formula as the interval t=t ₂ to _{t 3} .
However, in the formula, tθ is any time between t ₆ and t ₇ , and t ₂ is t ₆
Change it to a new one. Therefore, at t = t ₇ , v ₃ = 0 [ _V ] and v ₄ = 2Ein [ _V ], but i ₁ and i ₂ still remain, which makes v ₃
Below 0 [ _V ], an attempt is made to charge _v4 to more than 2Ein [ _V ], but the damper diode 11 is positively biased and becomes conductive, and does not change thereafter. The interval t=t ₇ to _{t 8} is a period from when the current first flows to the damper diode 11 until the current reaches 0 [ _A ]. The diode 11 is in a conductive state, and a voltage of Ein [ _V ] is applied to the primary winding of the transformer 5 by the power supply 1 in the same way as in the interval t=t ₀ -t ₁ . Further, for the same reason as for the other section t= _t0 to _t1 , the equivalent circuit shown in FIG. 3h is obtained. If the flow direction of i ₁ is positive in the interval t = t ₃ to _{t 4} , then
The time t ₈ -t ₇ in this interval is obtained from the same equation as the equation for t ₄ -t ₃ . Further, v ₃ =0 [ _V ] and v ₄ =2Ein [ _V ], which do not change, and i ₁ and i ₂ change using the same formula as in the interval t = t ₃ to t ₄ .
However, tθ is any time between _t7 and _t8 , and _t3 is replaced with _t7 . Therefore, i ₁ = i ₂ at t = t ₈ , and otherwise this t = t ₈
The state at becomes the initial condition for t=t ₀ above, and a new cycle begins. From the above points, as can be seen from FIG. Since the voltage across the switching element does not rise suddenly when the switching element goes from a closed state to an off state, the time during which the voltage and the current flowing through the switching element increase can be extremely shortened. Next, when this embodiment is applied to a switching type power supply, there is a problem with the output power, which is t.
The value obtained by integrating the equation of i ₂ from =t ₀ to _{t 8} and multiplying it by Eout is the output power value supplied to the load from the load end 10 in FIG. In addition, in order to change the output power, the conduction time of the switching elements 3 and 4, that is, t ₁ −t ₀ , t ₅ −t ₄
All you have to do is change the . However, in that case, as can be seen from some of the formulas above, the
Since i ₁ and i ₂ are the initial conditions for the next section and influence the time etc. of the next section, t ₁ − t ₀ , t ₅ −
When t ₄ changes, the entire period t ₀ to t ₈ also changes. When changing the output power due to this, it is necessary to consider the switching period as well as the conduction time of the switching element. For example, in Figure 5, L1 is 40 [〓 _H) , L2 is 120 [〓 _H ], and C is
22000 [ _PF ], Ein is 65 [ _V ], Eout is 56 [ _V ], switching elements and diodes are all ideal switches, power transmission transformer is ideal transformer, t ₁ − t ₀ = t ₅
FIG. ₇ is a diagram showing the relationship between t ₁ −t ₀ and t ₅ −t ₄ , output power Pout, and switching frequency when −t 4 =ton. As you can see from Figure 5, for example
To change Pout from 20 [ _W ] to 100 [ _W ], change ton from 2.26 [〓 _s ] to 3.5 [〓 _S ] and from 91.1 [ _K Hz] to 81.4 [ _K Hz]. Good. Furthermore, although the above operation is the basic operation of this embodiment, when the time during which the switching elements 3 and 4 are conductive in the above operation, that is, t ₁ - _{t 0} and t ₅ - _{t 4} are shortened, t The circuit may operate in such a way that t ₂ becomes 0 [ _A ] from a certain point between ₁ and t ₃ until t ₃ , and from a certain point between t ₅ and t ₇ until t ₇ , but even in that case, the basic There are no particular problems with this. As described above, according to the present invention, the time during which the current flowing through the switching element and the voltage across it increase can be extremely shortened, and switching loss can be extremely reduced, resulting in a drastic reduction in the heat generation of the switching element. , the life of the element itself is extended, and reliability and safety are improved. In addition, when applied to switching power supplies, etc., the heat sink becomes smaller and the switching frequency can be increased, so inductance and capacitors can be reduced, and direct current is not superimposed on the transformer, so the effect of its use is good. As a result, the power supply device can be made smaller and lighter. Furthermore, the response speed to input fluctuations and load fluctuations is also improved. Furthermore, as can be seen from FIGS. 4a and 4e, since only 2Ein [ _V ] is applied to both ends of the switching element, a relatively low breakdown voltage transistor can be used. Furthermore, since the voltage across the switching element and the voltage and current in the primary winding of the power transmission transformer do not change abruptly, noise generated therefrom can be extremely reduced. Note that the present invention is not limited to the above-described embodiments, and can be implemented with various modifications without changing the gist. FIG. 6 shows an example in which the Chiyoke coil 6 is connected to the diodes 7 and 8 via the capacitor 9 from the middle point of the secondary winding of the power transmission transformer 5. This is an example in which the coil is connected in series to the primary winding of the coil. In these two examples, substantially the same circuit operation and effects as in the above embodiment can be obtained. If the output power does not need to be changed much, v ₃ = v 3 at t ₃ and t ₇ in Figure 4 a to j, respectively.
If the circuit constants and the conduction times of the switching elements 3 and 4 are adjusted so that i ₁ =i ₂ at the same time as 2Ein [ _V ], the damper diode can be omitted as shown in FIG. In addition, if the leakage isoductance generated equivalently between the primary and secondary sides of the transformer 5 is large, it will act as a chiyoke coil, so as shown in FIG. can be omitted. In Fig. 10, two power supplies 1 and 2 are used in Fig. 2, but the voltage of one power supply 15 is divided by capacitors 16 and 17, so that two power supplies are equivalently used.
It was designed so that it would become one. However, in order to perform the same operation as the circuit shown in FIG. , 14 must be set to a large value so as to have almost no effect on the resonance motion. FIG. 11 shows a special example in which damper diodes 11 and 12 are connected between the base and emitter of bipolar transistors used as the switching elements 3 and 4. This is equivalent to the diode and damper diode 11 or 1 that exist between the collector and base of the bipolar transistor.
2 are connected in series, and when the two diodes in series are in a damped state, the withstand voltage is borne by the diode between the collector and base of the transistor, so damper diodes 11 and 12 have a comparative Diodes with low breakdown voltage can be used. FIGS. 12 to 15 show circuits for taking out two outputs. Figures 12 and 13 show load end 1.
Both ends are the same polarity with respect to the midpoint of 8, Fig. 14, 1
Figure 5 shows different polarities. In addition, in FIGS. 12 and 14, since the chiyoke coils 19 and 20 are inserted at the respective outputs relative to the midpoint of the load end 18, they are not easily affected by the output on the opposite side, but the chiyoke coils 19 and 20 Because current flows in only one direction, the core used in the chiyoke coil is inefficiently utilized. In Fig. 15, a chiyoke coil 6 is connected to the middle point of the secondary winding of the transformer 5 , and current flows in both directions, so the core used for the chiyoke coil 6 is used efficiently. easily affected by side output. In addition, in Figures 12 to 15, by shifting the tap at the midpoint of the secondary winding of transformer 5 to either side, or by setting the conduction times of switches 3 and 4 to different times instead of the same, the load end Viewed from the midpoint of 18, the absolute value of the voltage on both sides can be changed. As a result, in FIGS. 12 and 13, if the load is connected to both ends of the load end 18 without using the midpoint, the voltage will be the same as that of the capacitor 21.
Since the voltage is the difference between the voltages at both ends of and 22, by changing the conduction time of switches 3 and 4, an output in which the voltage changes smoothly from positive to negative or from negative to positive can be obtained. In Figures 12 and 13, the outputs on both sides are the midpoint of the secondary winding of the transformer 5 and the load end 1.
However, if the secondary winding of the transformer 5 is cut off, two midpoints of the load end 18 are provided, and two lines are connected to each other, the outputs on both sides can be isolated. can do. In this case, the number of outputs is not limited to two, and it is also possible to obtain a plurality of outputs from only one side or a plurality of outputs from both ends. Of course, you are free to connect or isolate each output. Although the connection method of the primary winding of the transformer 5 in FIG . 16 is different from the circuit in FIG. 2, it is a circuit that can obtain almost the same circuit operation and effect. Double. 17 to 21 show circuits in which the connection positions of the resonance capacitors are different, and the capacitors 13, 1
4 and 23 are resonance capacitors. Although the circuit operation is slightly different, the same effect can be obtained. As described in the explanation of the embodiment of the circuit shown in FIG. 2, when controlling the output by changing the conduction time of the switching elements 3 and 4, the switching period must be taken into consideration. However, as can be seen from the above explanation, after the current flows through the diode 11,
After switch 3 becomes conductive and current flows through diode 12, switch 4 only needs to become conductive, so as shown in FIG. You can do a forced synchronization. In this example, current transformers 24 and 25 are used for current detection, but other methods may of course be used. Several embodiments and modifications have been described above, but
Although most of the diagrams use so-called double-wave rectification for output rectification, it is of course possible to use a rectification method such as bridge rectification. In addition, the direction of the rectifier diode is as follows in all embodiments and modifications:
It is possible to perform it in the opposite direction. Further, the number of outputs is not limited to one or two, and by providing a plurality of secondary winding walls of the transformer 5 , it is possible to obtain a plurality of outputs. It is also possible to use a combination of some of the above embodiments and modifications.

[Brief explanation of the drawing]

Fig. 1 is a diagram showing the configuration of a conventional half-bridge type switching circuit, Fig. 2 is a circuit diagram of an embodiment of the present invention, Fig. 3 is an equivalent circuit diagram of each period in the embodiment, and Fig. 4 is a diagram of the same embodiment. In the voltage and current waveform diagrams of important parts, a and d are current waveform diagrams flowing through each switching element, b and e are voltage waveform diagrams at both ends thereof, c and f are current waveform diagrams flowing through each damper diode, g is the current waveform diagram flowing through the inductance seen from the primary side of the power transmission transformer,
h and i are current waveform diagrams flowing through each output rectifier diode, j is a current waveform diagram flowing through the choke coil, and Figure 5 shows the relationship between the conduction time of the switching element, output power, and switching frequency in the same example. 6 to 22 are circuit configuration diagrams showing other modified examples of the embodiment. 1, 2, 15... Input DC power supply, 3, 4... Switching element, 5... Power transmission transformer, 6,
19, 20... Chiyoke coil, 7, 8... Output rectifier diode, 9, 21, 22... Output smoothing capacitor, 10, 18... Load end, 11, 12
...damper diode, 13, 14, 23...
...Resonance capacitor, 16, 17...Input voltage dividing capacitor.

Claims (1)

[Claims] 1. Two input DC power supplies are connected in series, and a circuit in which two switching elements are connected in series is connected in parallel to this series circuit, and power transmission is performed between the connection point of the two DC power supplies and the connection point of the two switching elements. A damper diode and a resonant capacitor are connected in parallel to both ends of each switching element, and a rectifier circuit and a choke coil are connected to the secondary winding of the transformer. A half-bridge type high-frequency switching circuit characterized in that a smoothing capacitor is connected through the capacitor, and a load is connected to both ends of the smoothing capacitor.