JPH067744B2 - Current resonance type converter - Google Patents

Description

The present invention relates to improvement of control characteristics of a discontinuous mode current resonant converter having a period in which no current flows.

<Prior Art> A conventional current resonance type converter utilizes LC resonance to
It is a method of switching with a sinusoidal waveform that smoothly changes the current of switching elements such as MOSFET, and the current waveform during switching is sinusoidal (resonance waveform), and the resonance current is zero. Since it can be switched when it becomes, noise and switching loss during switching are small. Therefore, this current resonance type converter is considered to be an effective method for reducing the noise and the frequency of the switching power supply (related to downsizing of the device).

The current resonance type includes a discontinuous mode in which a current does not flow and a continuous mode in which the current does not flow. The discontinuous mode is roughly classified into a half waveform and a full waveform. , A current resonant converter in a discontinuous mode that flows only when switching is on.

The output voltage control in this current resonance type converter is performed by the ratio (fs / f) of the switching frequency fs and the resonance frequency fn.
This is done by changing n). Especially in the case of full-wave current resonance type, if the circuit loss is neglected, the output voltage has no load dependency and is affected only by the input voltage fluctuation, and the input / output voltage relationship is as shown in the following equation. It is determined by the ratio of the frequency to the resonant frequency.

Vout / Vin = fs / fn ... where Vin is the input voltage (DC) and Vout is the output voltage (DC).

<Problems to be Solved by the Invention> However, in the current resonance type converter shown in the above-mentioned prior art, since the fixed inductor or the fixed capacitor is used as the resonance element, the resonance frequency fn shown in the above formula is constant. In order to control the output voltage Vout, it is necessary to change the switching frequency fs. Therefore, since the switching frequency fluctuates, a beat occurs when operating in parallel. Also, noise countermeasures are complicated. Further, there are problems such as hindering miniaturization of capacitors, choke coils, transformers and the like.

The present invention has been made in view of the above problems of the prior art, and an object of the present invention is to provide a current resonance type converter in which the output voltage can be controlled while keeping the switching frequency fixed. Is.

<Means for Solving the Problems> The configuration of the present invention for solving the above problems is a variable inductor in a discontinuous mode current resonance converter that controls an output voltage to a constant value according to an input fluctuation or a load fluctuation. And a capacitor, and an output voltage control circuit that increases or decreases the control current of the variable inductor based on the difference between the output voltage and the reference voltage. The control current changes the inductance value of the variable inductor to change the resonance frequency. It is characterized in that the output voltage is controlled to a constant value by changing it.

<Operation> According to the present invention, the control current is increased or decreased depending on the level of the output voltage, the inductance value of the variable inductor is changed, and the resonance frequency is changed, so that the switching frequency remains constant and the output voltage remains constant. You can control.

<Examples> Hereinafter, the present invention will be described with reference to the drawings.

FIG. 1 is a block diagram showing an embodiment of the current resonance type converter according to the present invention, which is a full waveform of a half bridge circuit system.

In FIG. 1, Vin is an input voltage, 1 and 2 are input voltages V
A dividing capacitor for dividing in into two, 3 and 4 are input voltage Vin
A switching element such as MOSFET connected in series at both ends of
Gate drive circuits 5 and 6 are connected between the gate and source terminals of the MOSFETs 3 and 4 to drive the MOSFETs 3 and 4, 7 and 8
Is a parasitic diode of MOSFETs 3 and 4, 9 is a variable inductor for resonance in which one end of an inductor winding is connected to the connection point of split capacitors 1 and 2, 10 is the connection point of semiconductor switches 3 and 4 and the inductor of variable inductor 9. A transformer in which both ends of the primary winding are connected to the other end of the use winding, 11 is a transformer 1
A resonance capacitor connected to both ends of the primary winding of 0,
Reference numerals 12 and 13 denote rectifying diodes whose anodes are connected to both ends of the secondary winding of the transformer 10, 14 is a choke coil whose one end is connected to cathodes of the rectifying diodes 12 and 13, and 15 and 16 are chokes. A filter capacitor and a load resistor connected between the other end of the coil 14 and the middle point of the transformer 10, and 14 and 15 form an output filter. Vout is the output voltage applied across the load resistance. Further, 17 and 18 are output voltage detection resistors for detecting the voltage across the load resistor 16, 19 is an output reference voltage value, and 20 is a divided output voltage by the output voltage detection resistors 17 and 18 and an output reference voltage value 20. An operational amplifier for taking the difference and amplifying, 21 for a base drive circuit driven by the voltage across the load resistor 16, and 22 for current control for amplifying the output of the base drive circuit 21 and driving the control winding of the variable inductor 9. And the transistors 17 to 22 form an output voltage control circuit. Although MOSFETs are used as the switching elements in the above configuration diagram, the present invention is not limited to this.

2 is a principle diagram of the variable inductor 9 used in FIG.

In FIG. 2, two cores 23, 2 having the same material shape are used.
4 is wound, and the inductor side is connected to the resonance circuit and the control side is connected to the control circuit. That is, the resonance current IL flows through the inductor winding 25, and the control current Ict flows through the control winding 26.

Next, the operating principle of this variable inductor will be described. Note that FIG. 3 is a diagram showing the movement of the magnetic flux on the magnetization (B-H) curve, and FIG. 4 is a characteristic diagram of the variable inductor.

In FIG. 2, when the resonance current IL flows and the ampere-turn on the control side is larger than the ampere-turn on the inductor side while the direct-current control current Ict1 is flowing, the magnetization characteristic of the core 23 is as shown in FIG. Like the first quadrant, the vehicle has a minor loop drawn by the resonance current IL around the operating point determined by the control current Ict1 on the initial magnetization curve.
The slope of the minor loop is called incremental permeability. If this control current Ict1 changes, the minor loop due to the resonance current IL moves on the initial magnetization curve, so the incremental permeability changes. On the other hand, the core 24 has the control current Ict in the opposite direction, and has a characteristic symmetrical to the core 23 as in the third quadrant of FIG. The inductance Lr of the variable inductor at this time is expressed by the following equation.

Lr = ² μ0 · μd · N ² · A / l, where μ 0: Permeability of vacuum μ d: Incremental permeability of core N: Number of turns on the inductor side A: Cross-sectional area 1: Magnetic path length. Therefore, the inductance Lr of the variable inductor can be changed by changing the control current Ict, and as shown in FIG. 4, the inductance Lr increases as the control current Ict decreases.

FIG. 5 is an operation waveform diagram for explaining the operation of FIG. It is assumed that the output filter on the secondary side is sufficiently large and the load current I0 is always constant.

In FIG. 5, (period of T0 ≦ t ≦ T1), immediately before time T0, the two MOSFETs 3 and 4 are both off, and the load current I0 flows back through the rectifying diodes 12 and 13 on the secondary side. There is. At time T0, when the MOSFET 3 turns on, the drain-
The source current Ids1 linearly increases through the primary winding of the transformer 10 and the variable inductor 9.

(Period of T1 ≦ t ≦ T2) When the rectifying diode 13 is turned off at time T1, the resonance by the resonance capacitor 11 and the variable inductor 9 is started. The resonance current IL of the variable inductor 9 and the voltage Vc across the resonance capacitor 11 are expressed by the following equation.

IL (t) = I0 / N + (Vin / 2 / Zn) .sin.omega. (T-T1) = Ids1 + Id1 ... Vc (t) = Vin / 2- (Vin / 2) .cos.omega. (T-T1) .. However, N : Transformer 10 turns ratio Id1: Parasitic diode (7) current.

(Period of T2 ≦ t ≦ T3) The resonance current IL becomes negative and the parasitic diode 7 of the MOSFET 3
A current flows in the direction of regenerating electric power to the input side through.
If the MOSFET 3 is turned off during this time (time Ta), the MOSFET 3
It can be turned off when no current is flowing to it (zero current switching), so no switching loss occurs.

(Period of T3 ≦ t ≦ T4) At time T3, the resonance current IL stops flowing,
The resonance capacitor 11 continues to discharge linearly until the rectifying diode 13 is turned on, and the resonance capacitor 11 V
As shown in the waveform chart c, the discharge ends at time T4.

In the period of T1 ≦ t ≦ T3, only the rectifying diode 12 is turned on on the secondary side.

After that, when the MOSFET 4 is turned on, the same operation as described above is performed, but the resonance current I flowing through the variable inductor 9 is changed.
The direction of L is opposite to that of MOSFET 3.

Thus, the input energy is sent to the secondary side while the MOSFETs 3 and 4 are alternately turned on.

Next, the output voltage control by the variable inductor will be described with reference to the comparison diagrams of the resonance waveform with respect to the input voltage fluctuation shown in FIG. 1 and FIG.

As shown in FIG. 6, when operating at the input voltage Vin1 (FIG. 6 (a)), the input voltage dropped to Vin2 (the sixth voltage).
(B)), the relationship between the input / output voltage ratio and fs / fn can be calculated from the above equation as follows: Vout / Vin = fs / fn Therefore, the output voltage Vout also decreases. Output voltage Vout
Is decreased, the negative input of the operational amplifier 20 becomes smaller than the output reference voltage value 19, and the output of the operational amplifier 20 becomes positive. Base drive circuit 21 and current control transistor 22
Since the control current Ict is reduced when the output of the operational amplifier 20 is positive, the inductance Lr of the variable inductor 9 increases.

When the inductance of the variable inductor 9 at the input voltage Vin1 is Lr1 and the inductance of the variable inductor 9 at the input voltage Vin2 is Lr2, the characteristic impedance Zn
And the resonance frequency fn are respectively expressed by the following equations.

(When input voltage Vin1) However, Zn1 <Zn2, fn1> fn2, Lr1 <Lr2, and the output voltage before and after the input voltage fluctuation is given by the following equation.

(Before change) Vout1 = (fs / fn1) Vin1 (after change) Vout2 = (fs / fn2) Vin2 ... ″ However, Vin1> Vin2 and fs are constant. , The resonance frequency drops to fn2, and the output voltage becomes Vout1 = Vou.
Stabilizes at t2. In this way, by controlling the variable inductor, the output voltage can be kept constant against input fluctuations.

Also, in the actual circuit, the output voltage changes even when the load fluctuates, but it is much smaller than the effect of the input fluctuation, and it can be handled by changing the resonance frequency and controlling the input / output voltage ratio as described above. I can.

In the above embodiment, the full-waveform current resonance type is explained, but even in the half-waveform which is the current resonance type converter in the mode where the current is discontinuous, the output voltage is controlled by the ratio between the switching frequency and the resonance frequency. Since the change is performed, the method of the present invention can be applied.

<Effects of the Invention> As described above in detail with the embodiments, according to the present invention, since the output voltage can be controlled to a constant value while the switching frequency remains constant, noise countermeasures are facilitated. Also, by increasing the switching frequency, it is possible to reduce the size of capacitors, transformers, choke coils, etc., and to realize a current resonance type converter that can eliminate the occurrence of beats that occur when operating in parallel. .

[Brief description of drawings]

FIG. 1 is a block diagram showing an embodiment of a current resonance converter according to the present invention, FIG. 2 is a block diagram showing an example of a variable inductor used in FIG. 1, and FIG. 3 is a diagram showing a magnetic flux on a magnetization curve. FIG. 4 is a characteristic diagram of the variable inductor, FIG. 5 is an operation waveform diagram for explaining the operation of FIG. 1, and FIG. 6 is a comparison diagram of resonance waveform with respect to input voltage fluctuation. 9 ... Variable inductor for resonance, 11 ... Capacitor for resonance, 17, 18 ... Output voltage detection resistance, 19 ... Output reference voltage value, 20 ... Operational amplifier, 21 ... Base drive circuit, 22
... transistor, Vin ... input voltage, Vout ... output voltage.

Front page continuation (56) References JP-A-57-182217 (JP, A) JP-A-1-114367 (JP, A) JP-B-62-36466 (JP, B2) Jovanovi c, Lee, Ch en "A ZERO-CORENT-S WITCHED OFF-LINE QUA ASI-RESONANT CONVERTER TER WITH WIT REDUCED FR EQUENCY RANGE" HFPC MAY 1988 PROCEEDINGS P. 15-24

Claims (1)

[Claims] 1. A current resonance type converter of a discontinuous mode in which an output voltage is controlled to a constant value according to an input fluctuation or a load fluctuation, and an LC resonance circuit comprising a variable inductor and a capacitor connected in series to the variable inductor. An output voltage control circuit that increases or decreases the control current of the variable inductor based on the difference between the output voltage and the reference voltage, and changes the inductance value of the variable inductor by the control current to change the resonance frequency, thereby making the output voltage constant. A current resonance type converter characterized by being configured to be controlled to a value.