WO2020063331A1 - 感应电机的控制装置和控制方法 - Google Patents

感应电机的控制装置和控制方法 Download PDF

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Publication number
WO2020063331A1
WO2020063331A1 PCT/CN2019/105160 CN2019105160W WO2020063331A1 WO 2020063331 A1 WO2020063331 A1 WO 2020063331A1 CN 2019105160 W CN2019105160 W CN 2019105160W WO 2020063331 A1 WO2020063331 A1 WO 2020063331A1
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WIPO (PCT)
Prior art keywords
stator
angle
command
control device
correction amount
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Ceased
Application number
PCT/CN2019/105160
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English (en)
French (fr)
Inventor
汪宇涛
刘少枫
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Omron Shanghai Co Ltd
Omron Corp
Original Assignee
Omron Shanghai Co Ltd
Omron Corp
Omron Tateisi Electronics Co
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Omron Shanghai Co Ltd, Omron Corp, Omron Tateisi Electronics Co filed Critical Omron Shanghai Co Ltd
Priority to EP19866437.7A priority Critical patent/EP3859968B1/en
Priority to US17/266,621 priority patent/US11271506B2/en
Priority to JP2021506554A priority patent/JP7074256B2/ja
Publication of WO2020063331A1 publication Critical patent/WO2020063331A1/zh
Anticipated expiration legal-status Critical
Ceased legal-status Critical Current

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/24—Vector control not involving the use of rotor position or rotor speed sensors
    • H02P21/28—Stator flux based control
    • H02P21/30—Direct torque control [DTC] or field acceleration method [FAM]
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H02P21/141—Flux estimation
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/14—Estimation or adaptation of machine parameters, e.g. flux, current or voltage
    • H02P21/18—Estimation of position or speed
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/22—Current control, e.g. using a current control loop
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P21/00—Arrangements or methods for the control of electric machines by vector control, e.g. by control of field orientation
    • H02P21/24—Vector control not involving the use of rotor position or rotor speed sensors
    • H02P21/28—Stator flux based control
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02P—CONTROL OR REGULATION OF ELECTRIC MOTORS, ELECTRIC GENERATORS OR DYNAMO-ELECTRIC CONVERTERS; CONTROLLING TRANSFORMERS, REACTORS OR CHOKE COILS
    • H02P2207/00—Indexing scheme relating to controlling arrangements characterised by the type of motor
    • H02P2207/01—Asynchronous machines

Definitions

  • the present application relates to the technical field of motors, and in particular, to a control device and a control method for an induction motor.
  • FIG. 1 is a system block diagram of a speed sensorless vector control of a motor in the prior art. As shown in Figure 1:
  • the magnetizing current instruction calculation means 101 generates a magnetizing current instruction according to the input magnetic flux instruction
  • the magnetizing current adjusting means 103 obtains the magnetizing voltage command according to the magnetizing current command and the magnetizing current fed back by the park transforming means 110;
  • the torque current command calculation means 116 generates a torque current command according to the input speed command value and the speed estimation value fed back by the speed estimation means 111;
  • the torque current adjusting means 102 obtains a torque voltage command according to the torque current fed back by the torque current command and the park conversion means 110;
  • the anti-Park-Clark transform means 104 performs an inverse Park-Clark transform on the magnetizing voltage command and the torque voltage command based on the magnetic chain command angle to obtain U, V, and W three-phase stator voltage commands;
  • the inverter unit 200 generates U, V, and W three-phase stator voltages for input to the stator of the motor M according to the stator voltage command, thereby controlling the motor M.
  • the speed estimation value is generated as follows: the actual output U, V, and W three-phase stator voltages and the current flowing through the stator of the motor M are sampled, and the ⁇ - ⁇ coordinates are obtained by Clark transform means 107 and 108 The stator voltage and current signals of the system; the current and flux estimation means 109 generates the estimated current based on the speed estimation value fed back by the speed estimation means 111 and the stator voltage and current signals of the ⁇ - ⁇ coordinate system output by Clark transform means 107 and 108. And the estimated magnetic flux; the speed estimation means 111 calculates a new velocity estimation value based on the estimated current, the estimated magnetic flux, and the current signal output by the Clark transform means 108.
  • the Park transform means 110 performs Park transform on the current signal output by the Clark transform means 108 based on the magnetic flux command angle to obtain the magnetizing current and the torque current.
  • the generation method of the magnetic flux command angle is as follows: slip frequency calculation means 112 generates slip frequency; stator frequency calculation means 113 calculates the stator frequency according to the speed estimation value and slip frequency fed back by speed estimation means 111. ⁇ 1 ; the integrator 117 integrates the stator frequency ⁇ 1 to obtain the flux command angle.
  • the stator voltage input to the motor M can be controlled, thereby performing vector control on the motor M.
  • the speed estimation value ⁇ 1 calculated by the speed estimation means 111 may be located in an unstable region, that is, ⁇ 1 ⁇ ( ⁇ 1th , ⁇ 1th ).
  • the control system will be out of control.
  • Patent Document 1 discloses a control system for a motor, which can cause the speed estimation value ⁇ 1 calculated by the speed estimation means to be located outside the unstable region, thereby preventing the control system from losing control.
  • FIG. 2 is a schematic diagram of a motor control system described in Patent Literature 1.
  • FIG. 2 is different from the control system shown in FIG. 1 in that the control system of FIG. 2 adds a stator frequency lower limit limit determination method 114 and a speed command correction amount calculation method 115.
  • the determination method 114 for the stator frequency lower limit limit can discriminate based on the speed command value input to the control system, the slip frequency calculated by the slip frequency calculation means 112, and the lower limit value ⁇ 1th of the stator frequency. And output the discrimination result.
  • the stator frequency calculation means 113 performs processing based on the determination result of 114 so that the speed estimation value output by the stator frequency calculation means 113 is outside the unstable region.
  • the speed command correction amount calculation means 115 calculates the speed command correction amount according to the stator frequency lower limit value ⁇ 1th , the speed command value, and the slip frequency, and outputs different speed command correction amounts according to the determination result of 114. Therefore, the torque current command calculation means 116 in FIG. 2 calculates the torque current command based on the speed command value, the speed estimation value, and the speed command correction amount.
  • Patent Document 1 Japanese Patent Application Laid-Open No. 2010-22096A
  • the inventor of the present application has found that in Patent Document 1, the stator frequency calculation means 113, the stator frequency lower limit determination means 114, and the speed command correction amount calculation means 115 all need to be calculated based on the slip frequency.
  • the slip frequency ⁇ s is usually calculated by the following formula (1):
  • i q is the torque current
  • ⁇ r is the rotor flux linkage
  • L m is the mutual inductance of the rotor and the stator
  • R r is the resistance of the rotor.
  • the calculation accuracy of the slip frequency ⁇ s is greatly affected by the rotor resistance R r .
  • the error of the actual R r compared to the nominal value will even increase to more than 20%.
  • the calculation deviation of ⁇ s will increase significantly. Therefore, the calculation results of the stator frequency calculation method 113, the stator frequency lower limit determination method 114, and the speed command correction amount calculation method 115 will all have deviations. Therefore, the control system is still possible.
  • the operation is in an unstable area, which causes the control system to run out of control.
  • the present application provides a control device and a control method for an induction motor.
  • the speed sensorless vector control of the motor by using the control device calculation using a slip frequency ⁇ s that is subject to errors due to resistance changes is avoided. It can make the control system run outside the unstable area and improve the stability of control by the control device.
  • a control device for an induction motor including:
  • Magnetizing current adjustment unit for calculating magnetizing voltage command
  • Torque current adjustment unit used to calculate the torque voltage command
  • Flux instruction angle generating unit which estimates the angle based on a preset lower limit value of the stator frequency ⁇ 1th , the stator frequency ⁇ 1 , and the flux linkage Calculate the flux command angle;
  • a motor stator voltage instruction calculation unit calculates a stator voltage instruction for controlling a stator of a motor according to the magnetized voltage instruction, the torque voltage instruction, and the flux linkage instruction angle.
  • the magnetic flux instruction angle generating unit includes:
  • a frequency difference calculation unit that calculates a difference between the preset lower limit value ⁇ 1th of the stator frequency and the stator frequency ⁇ 1 ;
  • An integrating unit that integrates the difference to obtain the flux command angle correction amount
  • a correction unit that estimates an angle of the magnetic flux using the magnetic flux command angle correction amount Correction is performed to obtain the magnetic chain command angle.
  • the minimum value of the flux linkage command angle correction amount is 0, that is, the lower limit value of the integral is 0.
  • control device further includes:
  • Flux estimation angle calculation unit which is based on the ⁇ - ⁇ coordinate system based on the stator voltage signal, stator current signal, and speed estimation value of the motor. with Perform arctangent calculation to obtain the estimated angle of the flux linkage
  • control device further includes:
  • Stator frequency calculation unit which estimates the magnetic flux angle Derivative with respect to time to obtain the stator frequency ⁇ 1 .
  • control device further includes:
  • a stator frequency calculation unit that performs arc tangent on the ⁇ - ⁇ coordinate system-based currents I a and I b obtained based on the stator current flowing into the motor to obtain a current vector angle ⁇ I, and then the current vector angle ⁇ I Differential operation is performed to obtain the stator frequency ⁇ 1 .
  • control device further includes:
  • a speed command correction amount calculation unit configured to calculate a speed command correction amount according to the preset lower limit value ⁇ 1th of the stator frequency and the stator frequency ⁇ 1 ,
  • the speed command correction amount is used to adjust a speed command value, thereby calculating the torque voltage command.
  • the speed command correction amount calculation unit includes:
  • a first calculation unit configured to calculate a difference between the preset lower limit value ⁇ 1th of the stator frequency and the stator frequency ⁇ 1 ;
  • the first determination unit determines the speed command correction amount based on the difference calculated by the first calculation unit. Also includes:
  • a first PI adjustment unit which performs a PI operation according to a difference calculated by the first calculation unit, and outputs a PI operation result
  • the first determining unit determines that the speed command correction amount is 0 when the PI operation result is less than 0,
  • the first determination unit determines the speed command correction amount as the PI operation result.
  • a method for controlling an induction motor including:
  • a stator voltage command for controlling the stator operation of the motor is calculated based on the magnetizing voltage command, the torque voltage command, and the flux linkage command angle.
  • the beneficial effect of the present application is: avoid using the slip frequency ⁇ s for errors caused by resistance changes for calculation, thereby enabling the control system to run outside the unstable region and improving the stability of control performed by the control device.
  • FIG. 1 is a system block diagram of speed sensorless vector control of a motor in the prior art
  • FIG. 2 is a schematic diagram of a motor control system described in Patent Literature 1.
  • FIG. 2 is a schematic diagram of a motor control system described in Patent Literature 1.
  • FIG. 3 is a schematic diagram of a control device according to Embodiment 1 of the present application.
  • FIG. 4 is a schematic diagram of a flux instruction angle generating unit 204 in the control device of Embodiment 1 of the present application;
  • FIG. 5 is a schematic diagram of the stator frequency calculation unit 202a
  • FIG. 6 is a schematic diagram of a speed command correction amount calculation unit 201 according to Embodiment 1 of the present application.
  • FIG. 7 is a schematic diagram of the torque current command calculation unit 116.
  • FIG. 9 is a schematic diagram of a method for controlling an induction motor according to Embodiment 2 of the present application.
  • FIG. 3 is a schematic diagram of the control device of this embodiment.
  • the control device 30 of the induction motor controls the stator voltage input to the motor M according to the flux command, the speed command value, and the lower limit value of the stator frequency input to the control device 30 to perform vector control on the motor M. .
  • the control device 30 for an induction motor includes a magnetizing current adjusting unit 103, a torque current adjusting unit 102, a magnetic flux command angle generating unit 204, and a motor stator voltage command calculating unit 104.
  • the magnetizing current adjustment unit 103 may calculate a magnetizing voltage command according to the magnetizing current command and the magnetizing current.
  • the torque current adjusting unit 102 may calculate a torque voltage command according to the torque current and the torque current command; the flux linkage command angle generating unit 204 may calculate a preset lower limit value of the stator frequency ⁇ 1th , the stator frequency ⁇ 1 , and Estimated flux angle Calculate the flux command angle.
  • the motor stator voltage command calculation unit 104 may calculate a stator voltage command for controlling the stator of the motor based on the magnetized voltage command, the torque voltage command, and the flux linkage command angle. In one embodiment, the motor stator voltage command calculation unit 104 may perform an inverse Park-Clark transformation on the magnetized voltage command and the torque voltage command based on the magnetic flux command angle, thereby obtaining a stator voltage command.
  • stator voltage instruction obtained by the motor stator voltage instruction calculation unit 104 may be input to the inverter unit 200, thereby forming U, V, and W three-phase stator voltages output to the motor.
  • the magnetic flux command angle generating unit 204 can calculate the magnetic flux command angle without calculating the slip frequency ⁇ s of the motor, so the resistance of the motor rotor can be avoided. The effect of the change on the calculation result, so that the control device is operated outside the unstable area, and the stability of the control performed by the control device is improved.
  • FIG. 4 is a schematic diagram of the flux instruction angle generating unit 204 in the control device according to the first embodiment of the present application.
  • the magnetic flux command angle generation unit 204 includes a frequency difference calculation unit 204 a, an integration unit 204 b, and a correction unit 204 c.
  • the frequency difference calculation unit 204 a calculates a difference between a preset lower limit value ⁇ 1th of the stator frequency and the stator frequency ⁇ 1 .
  • the frequency difference calculation unit 204a may be a subtractor, and the calculated difference value is ⁇ 1th - ⁇ 1 .
  • the integration unit 204b may integrate the difference to obtain a flux linkage command angle correction amount.
  • the minimum value of the magnetic chain command angle correction amount is 0, for example, the anti-saturation lower limit value of the integration unit 204b is 0, that is, the integration result of the integration unit 204b is greater than or equal to 0.
  • the correction unit 204c estimates the angle of the magnetic flux using the magnetic flux command angle correction amount. Make corrections to get the flux command angle.
  • the correction unit 204c may be an adder that converts the magnetic flux command angle correction amount and the magnetic flux estimation angle. Add up to get the commanded angle of the flux linkage.
  • the flux command angle generation unit 204 can accurately control the flux command angle, and thus is equivalent to ensuring that the stator frequency ⁇ 1 corresponding to the flux angle under the control of the flux command angle is always greater than or equal to the lower limit value. ⁇ 1th , thereby preventing the control device 30 from operating in an unstable region.
  • the control device 30 may further include a magnetic flux estimation angle calculation unit 203.
  • the magnetic flux estimation angle calculation unit 203 may calculate the magnetic flux based on the ⁇ - ⁇ coordinate system for the stator voltage signal, the stator current signal, and the speed estimation value of the motor M. with Calculate the arc tangent to get the estimated angle of the flux linkage
  • the magnetic flux estimation angle calculation unit 203 can obtain the magnetic flux estimation angle according to the following formula (2)
  • the estimated flux angle obtained by the flux estimation angle calculation unit 203 It can be input to the magnetic flux instruction angle generating unit 204 for calculation.
  • the magnetic flux estimation angle calculation unit 203 calculates the magnetic flux estimation angle In this case, the use of the slip frequency ⁇ s of the motor for calculation is avoided, so the influence of the resistance change of the motor rotor on the calculation result can be avoided.
  • control device 30 may further include a current / flux estimation unit 109, Clark conversion units 107 and 108, and a speed estimation unit 111.
  • the voltage detection unit 105 detects the U, V, and W three-phase stator voltages actually output to the motor M to obtain a stator voltage signal; the current detection unit 106 detects the current flowing through the stator of the motor M to obtain a stator current signal.
  • Clark transform means 107 and 108 perform Clark transform on the stator voltage signal and the stator current signal to obtain the stator voltage and the stator current of the ⁇ - ⁇ coordinate system.
  • the current / flux estimation unit 109 generates an estimated current and an estimated flux based on the speed estimation value fed back by the speed estimation unit 111 and the stator voltage signal and the stator current signal of the ⁇ - ⁇ coordinate system output by the Clark transformation units 107 and 108.
  • the estimated magnetic flux is expressed as with among them, Is the estimated magnetic flux of the ⁇ axis, Estimated flux linkage for ⁇ axis.
  • the estimated magnetic flux is expressed as with Is input to the magnetic flux estimation angle calculation unit 203 for calculating the magnetic flux estimation angle
  • the speed estimation unit 111 may estimate the magnetic flux ( with ) And the stator current output from Clark transform unit 108 to calculate a new speed estimate.
  • the control device 30 may further include a stator frequency calculation unit 202.
  • the stator frequency calculation unit 202 may not calculate the stator frequency ⁇ 1 based on the slip frequency ⁇ s of the motor. Therefore, the influence of the resistance change of the motor rotor on the calculation result can be avoided, thereby allowing the control device to operate outside the unstable region and improve The control device performs control stability.
  • the stator frequency calculation unit 202 may estimate the magnetic flux angle. Derivative with respect to time, the stator frequency ⁇ 1 is obtained . For example, the stator frequency calculation unit 202 may calculate the stator frequency ⁇ 1 according to the following formula (3):
  • stator frequency calculation unit 202 may also use other methods to calculate the stator frequency ⁇ 1 .
  • the stator frequency calculation unit 202a may perform arc tangent on the ⁇ - ⁇ coordinate-based currents I a and I b based on the stator current flowing into the motor M to obtain the current vector angle ⁇ I, and then Differential operation is performed on the current vector angle ⁇ I to obtain the stator frequency ⁇ 1 .
  • FIG. 5 is a schematic diagram of the stator frequency calculation unit 202a. As shown in FIG. 5, the stator currents I U , I V , and I W detected by the current detection unit 106 are transformed into the ⁇ - ⁇ coordinate system based on the Clark transform unit 108. Currents I a and I b , where I a is the current on the ⁇ axis and I b is the current on the ⁇ axis.
  • the currents I a and I b are input to the stator frequency calculation unit 202 a.
  • the stator frequency calculation unit 202a includes a current vector angle calculation unit 205 and a frequency calculation unit 202a1.
  • the current vector angle calculation unit 205 obtains the current vector angle ⁇ I based on the currents I a and I b .
  • the current vector angle calculation unit 205 can calculate the current vector angle ⁇ I by the following formula (4):
  • the frequency calculation unit 202a1 may derive the current vector angle ⁇ I calculated by the current vector angle calculation unit 205 to calculate the stator frequency ⁇ 1.
  • the frequency calculation unit 202a1 may calculate according to the following formula (5) The stator frequency ⁇ 1 .
  • the control device 30 may further include a speed command correction amount calculation unit 201.
  • the speed command correction amount calculation unit 201 may calculate a speed command correction amount based on a preset lower limit value of the stator frequency ⁇ 1th and the stator frequency ⁇ 1 .
  • the speed command correction amount is used to adjust the speed command value input to the control device 30 to calculate the torque voltage command.
  • FIG. 6 is a schematic diagram of a speed command correction amount calculation unit 201 according to an embodiment of the present application. As shown in FIG. 6, the speed command correction amount calculation unit 201 includes a first calculation unit 201 a and a first determination unit 201 c.
  • the first calculation unit 201a is configured to calculate a difference between the preset lower limit value of the stator frequency ⁇ 1th and the stator frequency ⁇ 1 ; the first determination unit 201c determines the difference based on the difference calculated by the first calculation unit 201a. Speed command correction amount.
  • the first calculation unit 201 a may be a subtractor, and the calculated difference is, for example, ⁇ 1th - ⁇ 1 .
  • the difference is less than 0, it means that ⁇ 1 is outside the unstable region.
  • the first determination unit 201c determines that the output speed command correction amount is 0.
  • the difference is greater than or equal to 0, it means that ⁇ 1 is in Within the unstable region, at this time, the first decision unit 201c decides to output the difference as a speed command correction amount.
  • the speed command correction amount calculation unit 201 may further include: a first PI adjustment unit 201 b.
  • the first PI adjustment unit 201b can perform proportional integral (PI) calculation based on the difference calculated by the first calculation unit 201a, and output a PI calculation result.
  • PI proportional integral
  • the first decision unit 201c can output the speed command correction amount according to the PI operation result. For example, when the PI operation result is less than 0, the first decision unit 201c determines that the output speed command correction amount is 0; when the PI operation result is greater than or When it is equal to 0, the first determination unit 201c determines that the output speed command correction amount is the result of the PI operation.
  • the speed command correction amount calculation unit 201 in the present application can calculate the speed command correction amount without depending on the slip frequency ⁇ s of the motor, so the influence of the resistance change of the motor rotor on the calculation result can be avoided.
  • control device 30 may further include a torque current command calculation unit 116.
  • the torque current command calculation unit 116 may calculate a torque current command based on the input speed command value, the speed estimation value output by the speed estimation value calculation unit 111, and the speed command correction amount output by the speed command correction amount calculation unit 201.
  • FIG. 7 is a schematic diagram of the torque current command calculation unit 116.
  • the torque current command calculation unit 116 includes an adder 116a, a subtractor 116b, and a second PI regulator 116c.
  • the speed command finger is added to the speed command correction amount, and the result of the addition is subtracted from the speed estimation value to obtain a difference.
  • the difference is subjected to proportional integral (PI) calculation by the second PI regulator 116c.
  • PI proportional integral
  • the result of the PI calculation performed by the second PI regulator 116c is output to the torque current adjustment unit 102 as a torque current command value.
  • Patent Document 1 for descriptions of other units in FIG. 3, refer to Patent Document 1.
  • the torque current command calculation unit 116 is modified, a speed command correction amount adjustment unit 201, a stator frequency calculation unit 202, a magnetic flux estimation angle calculation unit 203, and a magnetic field are added.
  • Chain instruction angle calculation unit 204 Therefore, the stator frequency ⁇ 1 can be controlled within the stable region (or outside the unstable region) without relying on the slip frequency ⁇ s of the motor, that is, ⁇ 1 ⁇ 1th is guaranteed, so the control device is improved for control The stability.
  • the rated voltage of the induction motor is 200V
  • the rated power is 2.2kW
  • the motor is running forward
  • the load torque (Torque) is -27.6Nm
  • the stability zone boundary that is, the lower limit value of the preset stator frequency
  • FIG. 8 is a schematic diagram of vector control of the motor in this example.
  • the horizontal axis is time, and the unit is second (S), and the vertical axis is frequency, and the unit is hertz (Hz).
  • curve 801 is the actual rotational speed ⁇ r of the rotor of the motor M when the control device according to this embodiment performs control
  • curve 802 is the stator frequency ⁇ 1 of the motor M when the control device according to this embodiment performs control
  • curve 803 is the motor The slip frequency ⁇ s of M
  • the curve 804 is the stator frequency ⁇ 1 ′ of the motor M when controlled by the control device of Patent Document 1.
  • Embodiment 2 of the present application provides a control method of an induction motor, and the control method corresponds to the control device 30 of Embodiment 1.
  • FIG. 9 is a schematic diagram of a method for controlling an induction motor according to this embodiment. As shown in FIG. 9, the method includes:
  • Step 901 Calculate the magnetizing voltage instruction.
  • Step 902 Calculate the torque voltage instruction.
  • Step 903 Calculate the flux instruction angle according to the preset lower limit value of the stator frequency, the stator frequency, and the flux estimation angle;
  • Step 904 Calculate a stator voltage instruction for controlling the stator operation of the motor according to the magnetization voltage instruction, the torque voltage instruction, and the flux linkage instruction angle.
  • control method For the description of each step of the control method, reference may be made to the description of the corresponding unit in Embodiment 1.
  • control method may further include steps other than those described in FIG. 9, and reference may also be made to the description of each unit of the control device 30 in Embodiment 1.
  • the stator frequency ⁇ 1 can be controlled within the stable region (or outside the unstable region) without relying on the slip frequency ⁇ s of the motor, that is, ⁇ 1 ⁇ 1th is guaranteed, so the control device is improved to perform Control stability.
  • the parameter calculation device described in combination with the embodiments of the present application may be directly embodied as hardware, a software module executed by a processor, or a combination of the two.
  • These hardware modules can be implemented by using a field programmable gate array (FPGA) to cure these software modules.
  • FPGA field programmable gate array
  • a software module may reside in RAM memory, flash memory, ROM memory, EPROM memory, EEPROM memory, registers, hard disk, removable disk, CD-ROM, or any other form of storage medium known in the art.
  • a storage medium may be coupled to the processor so that the processor can read information from and write information to the storage medium; or the storage medium may be a component of the processor.
  • the processor and the storage medium may reside in an ASIC.
  • This software module can be stored in the memory of the mobile terminal or in a memory card that can be inserted into the mobile terminal. For example, if the electronic device uses a large-capacity MEGA-SIM card or a large-capacity flash memory device, the software module may be stored in the MEGA-SIM card or a large-capacity flash memory device.
  • the parameter calculation device described in this embodiment may be implemented as a general-purpose processor, a digital signal processor (DSP), an application specific integrated circuit (ASIC), or a field programmable gate array (FPGA) for performing the functions described in this application. Or other programmable logic devices, discrete gate or transistor logic devices, discrete hardware components, or any suitable combination thereof. It may also be implemented as a combination of computing devices, for example, a combination of a DSP and a microprocessor, multiple microprocessors, one or more microprocessors in conjunction with a DSP communication, or any other such configuration.
  • DSP digital signal processor
  • ASIC application specific integrated circuit
  • FPGA field programmable gate array

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  • Control Of Ac Motors In General (AREA)

Abstract

一种感应电机的控制装置和控制方法，该控制装置包括：磁化电流调节单元，用于计算磁化电压指令；转矩电流调节单元，用于计算转矩电压指令；磁链指令角度生成单元，其根据预设的定子频率的下限值ω 1th、定子频率ω 1、以及磁链估算角，计算磁链指令角度；以及电机定子电压指令计算单元，其根据所述磁化电压指令，所述转矩电压指令和所述磁链指令角度，计算用于对电机的定子进行控制的定子电压指令。能够使控制系统运行在不稳定区外，提高控制装置进行控制的稳定性。

Description

感应电机的控制装置和控制方法 技术领域
本申请涉及电机技术领域，尤其涉及一种感应电机的控制装置和控制方法。
背景技术
在对电机进行矢量控制时，有时难以在电机中设置用于检测转子速度的传感器，或者为了降低成本而不设置传感器，从而对电机进行无速度传感器的矢量控制。
图1是现有技术中对电机进行无速度传感器的矢量控制的一个系统框图。如图1所示：
磁化电流指令演算手段101根据输入的磁链指令生成磁化电流指令；
磁化电流调节手段103根据磁化电流指令与park变换手段110反馈的磁化电流得到磁化电压指令；
转矩电流指令演算手段116根据输入的速度指令值和速度估计手段111反馈的速度估算值生成转矩电流指令；
转矩电流调节手段102根据转矩电流指令与park变换手段110反馈的转矩电流得到转矩电压指令；
反Park-Clark变换手段104基于磁链指令角度，对磁化电压指令和转矩电压指令进行反Park-Clark变换，得到U、V、W三相的定子电压指令；
逆变单元200根据定子电压指令生成用于输入到电机M的定子的U、V、W三相定子电压，从而对电机M进行控制。
如图1所示，速度估算值的生成方式如下：采样实际输出的U、V、W三相定子电压、以及流过电机M的定子的电流，通过Clark变换手段107、108得到α-β坐标系的定子电压和电流信号；电流、磁链估算手段109基于速度估计手段111反馈的速度估算值，以及Clark变换手段107、108输出的α-β坐标系的定子电压和电流信号，生成估算电流和估算磁链；速度估计手段111根据该估算电流、估算磁链、以及Clark变换手段108输出的该电流信号，计算出新的速度估算值。
如图1所示，磁化电流和转矩电流的生成方式如下：Park变换手段110基于磁链指令角度，对Clark变换手段108输出的电流信号进行Park变换，得到磁化电流和转矩电流。
如图1所示，磁链指令角度的生成方式如下：滑差频率演算手段112生成滑差频率；定子频率演算手段113根据速度估计手段111反馈的速度估算值以及滑差频率，计算出定子频率ω 1；积分器117对定子频率ω1进行积分，得到磁链指令角度。
根据图1，通过向该控制系统输入磁链指令和速度指令值，能够控制输入到电机M的定子电压，从而对电机M进行矢量控制。
在图1所示的控制系统中，速度估算手段111计算出的速度估算值ω 1有可能位于不稳定区，即，ω 1∈(-ω 1th，ω 1th)。当速度估算值ω 1位于不稳定区时，将导致该控制系统失控。
专利文献1公开了一种电机的控制系统，能够使速度估算手段计算出的速度估算值ω 1位于不稳定区之外，由此，避免控制系统失控。
图2是专利文献1记载的电机的控制系统的一个示意图。如图2所示的控制系统与图1所示的控制系统相比，区别在于：图2的控制系统增加了定子频率下限limit实行判别手段114以及速度指令补正量演算手段115。
在图2中，定子频率下限limit实行判别手段114能够基于输入到该控制系统的速度指令值，滑差频率演算手段112所计算出的滑差频率，以及定子频率的下限值ω 1th进行判别并输出判别结果。在图2中，定子频率演算手段113根据114的判别结果进行处理，以使定子频率演算手段113输出的速度估算值位于不稳定区之外。
在图2中，速度指令补正量演算手段115根据定子频率下限值ω 1th，速度指令值，以及滑差频率计算速度指令补正量，并根据114的判别结果输出不同的速度指令补正量。从而，图2的转矩电流指令演算手段116基于速度指令值、速度估算值、以及速度指令补正量，计算转矩电流指令。
专利文献1：日本特开2010-22096A
应该注意，上面对技术背景的介绍只是为了方便对本发明的技术方案进行清楚、完整的说明，并方便本领域技术人员的理解而阐述的。不能仅仅因为这些方案在本发明的背景技术部分进行了阐述而认为上述技术方案为本领域技术人员所公知。
发明内容
本申请的发明人发现，在专利文献1中，定子频率演算手段113、定子频率下限limit实行判别手段114、以及速度指令补正量演算手段115都需要根据滑差频率进行计算。滑差频率ω s通常采用下式(1)来进行计算得到：
Figure PCTCN2019105160-appb-000001
其中，i q为转矩电流，ψ r为转子磁链，L m为转子和定子的互感，R r为转子的电阻。
由上述式(1)可知，滑差频率ω s的计算精度受转子电阻R r的影响很大。当电机运行发热时，实际R r的相比于标称值的误差甚至会增大到20％以上。ω s的计算偏差会显著增大，因此，定子频率演算手段113、定子频率下限limit实行判别手段114、以及速度指令补正量演算手段115的运算结果都将出现偏差，因此，控制系统仍有可能运行在不稳定区内，从而导致该控制系统失控。
本申请提供一种感应电机的控制装置和控制方法，在利用该控制装置对电机进行无速度传感器矢量控制中，避免使用受电阻变化而产生误差的滑差频率ω s来进行计算，由此，能够使控制系统运行在不稳定区外，提高控制装置进行控制的稳定性。
根据本申请实施例的一个方面，提供一种感应电机的控制装置，包括：
磁化电流调节单元，用于计算磁化电压指令；
转矩电流调节单元，用于计算转矩电压指令；
磁链指令角度生成单元，其根据预设的定子频率的下限值ω 1th、定子频率ω 1、以及磁链估算角
Figure PCTCN2019105160-appb-000002
计算磁链指令角度；以及
电机定子电压指令计算单元，其根据所述磁化电压指令，所述转矩电压指令和所述磁链指令角度，计算用于对电机的定子进行控制的定子电压指令。
根据本申请实施例的另一个方面，其中，所述磁链指令角度生成单元包括：
频率差计算单元，其计算所述预设的定子频率的下限值ω 1th与所述定子频率ω 1的差值；
积分单元，其对该差值进行积分以得到磁链指令角补正量；以及
补正单元，其使用所述磁链指令角补正量对所述磁链估算角
Figure PCTCN2019105160-appb-000003
进行补正，以得到所述磁链指令角度。
根据本申请实施例的另一个方面，其中，所述磁链指令角补正量的最小值是0，即，所述积分的下限值为0。
根据本申请实施例的另一个方面，其中，所述控制装置还包括：
磁链估算角计算单元，其针对基于电动机的定子电压信号、定子电流信号和速度估算值所计算的基于α-β坐标系的磁链
Figure PCTCN2019105160-appb-000004
和
Figure PCTCN2019105160-appb-000005
进行反正切计算，得到所述磁链估算角
Figure PCTCN2019105160-appb-000006
根据本申请实施例的另一个方面，其中，所述控制装置还包括：
定子频率计算单元，其将所述磁链估算角
Figure PCTCN2019105160-appb-000007
相对于时间求导，得到所述定子频率ω 1。
根据本申请实施例的另一个方面，其中，所述控制装置还包括：
定子频率计算单元，其对基于流入所述电机的定子电流而得到的基于α-β坐标系的电流I a和I b做反正切得到电流矢量角∠I，再对所述电流矢量角∠I进行微分运算得到所述定子频率ω 1。
根据本申请实施例的另一个方面，其中，所述控制装置还包括：
速度指令补正量计算单元，其用于根据所述预设的定子频率的下限值ω 1th和所述定子频率ω 1，计算速度指令补正量，
其中，所述速度指令补正量用于对速度指令值进行调整，从而计算所述转矩电压指令。
根据本申请实施例的另一个方面，其中，速度指令补正量计算单元包括：
第一计算单元，用于计算所述预设的定子频率的下限值ω 1th和所述定子频率ω 1的差值；
第一决定单元，根据所述第一计算单元计算出的差值，决定所述速度指令补正量。还包括：
第一PI调节单元，其根据所述第一计算单元计算出的差值进行PI运算，输出PI运算结果；
所述第一决定单元在所述PI运算结果小于0时，决定所述速度指令补正量为0，
所述第一决定单元在所述PI运算结果大于或等于0时，决定所述速度指令补正量为所述PI运算结果。
根据本申请实施例的另一个方面，提供一种感应电机的控制方法，包括：
计算磁化电压指令；
计算转矩电压指令；
根据预设的定子频率的下限值、定子频率、以及磁链估算角，计算磁链指令角度；以及
根据所述磁化电压指令，所述转矩电压指令和所述磁链指令角度，计算用于对电机的定子运转进行控制的定子电压指令。
本申请的有益效果在于：避免使用受电阻变化而产生误差的滑差频率ωs来进行计算，由此，能够使控制系统运行在不稳定区外，提高控制装置进行控制的稳定性。
参照后文的说明和附图，详细公开了本申请的特定实施方式，指明了本申请的原理可以被采用的方式。应该理解，本申请的实施方式在范围上并不因而受到限制。在所附权利要求的精神和条款的范围内，本申请的实施方式包括许多改变、修改和等同。
附图说明
所包括的附图用来提供对本申请实施例的进一步的理解，其构成了说明书的一部分，用于例示本申请的实施方式，并与文字描述一起来阐释本申请的原理。显而易见地，下面描述中的附图仅仅是本申请的一些实施例，对于本领域普通技术人员来讲，在不付出创造性劳动性的前提下，还可以根据这些附图获得其他的附图。在附图中：
图1是现有技术中对电机进行无速度传感器的矢量控制的一个系统框图；
图2是专利文献1记载的电机的控制系统的一个示意图
图3是本申请实施例1的控制装置的一个示意图；
图4是本申请实施例1的控制装置中磁链指令角度生成单元204的一个示意图；
图5是该定子频率计算单元202a的一个示意图；
图6是本申请实施例1的速度指令补正量计算单元201的一个示意图；
图7是转矩电流指令计算单元116的一个示意图；
图8是对该实例中的电机进行矢量控制的一个示意图；
图9是本申请实施例2的感应电机的控制方法的一个示意图。
具体实施方式
参照附图，通过下面的说明书，本申请的前述以及其它特征将变得明显。在说明书和附图中，具体公开了本申请的特定实施方式，其表明了其中可以采用本申请的原则的部分实施方式，应了解的是，本申请不限于所描述的实施方式，相反，本申请包括落入所附权利要求的范围内的全部修改、变型以及等同物。
实施例1
本申请实施例提供了一种感应电机的控制装置。图3是本实施例的控制装置的一个示意图。
如图3所示，感应电机的控制装置30根据向该控制装置30输入的磁链指令、速度指令值和定子频率下限值，控制输入到电机M的定子电压，从而对电机M进行矢量控制。
如图3所示，感应电机的控制装置30包括：磁化电流调节单元103、转矩电流调节单元102、磁链指令角度生成单元204、以及电机定子电压指令计算单元104。
如图3所示，磁化电流调节单元103可以根据磁化电流指令和磁化电流，计算磁化电压指令。
转矩电流调节单元102可以根据转矩电流和转矩电流指令，计算转矩电压指令；磁链指令角度生成单元204可以根据预设的定子频率的下限值ω 1th、定子频率ω 1、以及磁链估算角
Figure PCTCN2019105160-appb-000008
计算磁链指令角度。
电机定子电压指令计算单元104可以根据该磁化电压指令，该转矩电压指令和该磁链指令角度，计算用于对电机的定子进行控制的定子电压指令。在一个实施方式中，电机定子电压指令计算单元104可以基于磁链指令角度，对磁化电压指令和该转矩电压指令进行反Park-Clark变换，从而得到定子电压指令。
在本实施例中，电机定子电压指令计算单元104得到的定子电压指令可以被输入到逆变单元200，从而形成输出到电动机的U、V、W三相定子电压。
在本实施例中，关于磁化电流调节单元103、转矩电流调节单元102、电机定子电压指令计算单元104、以及逆变单元200的说明可以参考背景技术中提及的专利文献1。
根据本实施例，在感应电机的控制装置30中，磁链指令角度生成单元204在计算磁链指令角度时，可以不基于电机的滑差频率ωs来进行计算，所以，可以避免电 机转子的电阻变化对计算结果的影响，由此，使控制装置运行在不稳定区外，提高控制装置进行控制的稳定性。
图4是本申请实施例1的控制装置中磁链指令角度生成单元204的一个示意图。如图4所示，磁链指令角度生成单元204包括：频率差计算单元204a，积分单元204b，以及补正单元204c。
如图4所示，频率差计算单元204a计算预设的定子频率的下限值ω 1th与定子频率ω 1的差值。在一个实施方式中，频率差计算单元204a可以是减法器，其计算的差值是ω 1th-ω 1。
在本实施例中，积分单元204b可以对该差值进行积分，以得到磁链指令角补正量。在一个实施方式中，该磁链指令角补正量的最小值是0，例如，该积分单元204b抗饱和下限值为0，也就是说，该积分单元204b的积分结果大于或等于0。
在本实施例中，补正单元204c使用该磁链指令角补正量对磁链估算角
Figure PCTCN2019105160-appb-000009
进行补正，以得到该磁链指令角度。在一个实施方式中，补正单元204c可以是加法器，其将磁链指令角补正量与磁链估算角
Figure PCTCN2019105160-appb-000010
相加，得到该磁链指令角度。
在电机的实际运行过程中，实际磁链角与定子频率相对应。通过磁链指令角度生成单元204可以准确控制磁链指令角，由此，等效于保证了该磁链指令角控制下的磁链角所对应的定子频率ω 1始终大于或等于该下限值ω 1th，从而避免该控制装置30运行在不稳定区。
在本实施例中，如图3所示，该控制装置30还可以具有磁链估算角计算单元203。其中，磁链估算角计算单元203可以针对基于电机M的定子电压信号、定子电流信号和速度估算值所计算的基于α-β坐标系的磁链
Figure PCTCN2019105160-appb-000011
和
Figure PCTCN2019105160-appb-000012
进行反正切计算，得到该磁链估算角
Figure PCTCN2019105160-appb-000013
例如，磁链估算角计算单元203可以根据下式(2)得到磁链估算角
Figure PCTCN2019105160-appb-000014
Figure PCTCN2019105160-appb-000015
磁链估算角计算单元203得到的该磁链估算角
Figure PCTCN2019105160-appb-000016
可以被输入到磁链指令角度生成单元204进行计算。
根据本实施例，磁链估算角计算单元203在计算磁链估算角
Figure PCTCN2019105160-appb-000017
时，避免了使用电机的滑差频率ωs来进行计算，所以，可以避免电机转子的电阻变化对计算结果的影响。
在本实施例中，如图3所示，控制装置30还可以具有电流/磁链估算单元109，Clark变换单元107、108，以及速度估算单元111。
电压检测单元105对实际输出到电机M的U、V、W三相定子电压进行检测，得到定子电压信号；电流检测单元106对流过电机M的定子的电流进行检测，得到定子电流信号。
Clark变换手段107、108对定子电压信号和定子电流信号进行Clark变换，得到α-β坐标系的定子电压和定子电流。电流/磁链估算单元109基于速度估计单元111反馈的速度估算值，以及Clark变换单元107、108输出的α-β坐标系的定子电压信号和定子电流信号，生成估算电流和估算磁链，该估算磁链被表示为
Figure PCTCN2019105160-appb-000018
和
Figure PCTCN2019105160-appb-000019
其中，
Figure PCTCN2019105160-appb-000020
为α轴的估算磁链，
Figure PCTCN2019105160-appb-000021
为β轴的估算磁链。在本实施例中，估算磁链被表示为
Figure PCTCN2019105160-appb-000022
和
Figure PCTCN2019105160-appb-000023
被输入到磁链估算角计算单元203，用于计算磁链估算角
Figure PCTCN2019105160-appb-000024
此外，速度估算单元111可以根据该估算电流、估算磁链(
Figure PCTCN2019105160-appb-000025
和
Figure PCTCN2019105160-appb-000026
)、以及Clark变换单元108输出的定子电流，计算出新的速度估算值。
在本实施例中，如图3所示，控制装置30还可以具有：定子频率计算单元202。 定子频率计算单元202可以不基于电机的滑差频率ωs来计算定子频率ω 1，所以，可以避免电机转子的电阻变化对计算结果的影响，由此，使控制装置运行在不稳定区外，提高控制装置进行控制的稳定性。
如图3所示，在一个实施方式中，定子频率计算单元202可以将磁链估算角
Figure PCTCN2019105160-appb-000027
相对于时间求导，得到定子频率ω 1。例如，定子频率计算单元202可以根据下式(3)计算定子频率ω 1：
Figure PCTCN2019105160-appb-000028
此外，在本实施例中，定子频率计算单元202也可以采用其他的方式来计算定子频率ω 1。
例如，在另一个实施方式中，定子频率计算单元202a可以对基于流入电机M的定子电流而得到的基于α-β坐标系的电流I a和I b做反正切得到电流矢量角∠I，再对该电流矢量角∠I进行微分运算得到定子频率ω 1。
图5是该定子频率计算单元202a的一个示意图，如图5所示，电流检测单元106检测到的定子电流I U，I V，I W经过Clark变换单元108变换为基于α-β坐标系的电流I a和I b，其中，I a为α轴的电流，I b为β轴的电流。
在本实施例中，电流I a和I b被输入到定子频率计算单元202a。该定子频率计算单元202a包括：电流矢量角计算单元205和频率计算单元202a1。
如图5所示，电流矢量角计算单元205根据电流I a和I b得到电流矢量角∠I。其中，电流矢量角计算单元205例如可以通过下式(4)计算电流矢量角∠I：
Figure PCTCN2019105160-appb-000029
在本实施例中，频率计算单元202a1可以对电流矢量角计算单元205计算出的电流矢量角∠I求导，从而计算定子频率ω 1，例如，频率计算单元202a1可以根据下式(5)计算定子频率ω 1。
Figure PCTCN2019105160-appb-000030
在本实施例中，如图3所示，控制装置30还可以具有速度指令补正量计算单元201。速度指令补正量计算单元201可以根据预设的定子频率的下限值ω 1th和定子频率ω 1，计算速度指令补正量。其中，该速度指令补正量用于对输入该控制装置30的速度指令值进行调整，从而计算该转矩电压指令。
图6是本申请实施例的速度指令补正量计算单元201的一个示意图，如图6所示，速度指令补正量计算单元201包括：第一计算单元201a和第一决定单元201c。
其中，第一计算单元201a用于计算所述预设的定子频率的下限值ω 1th和定子频率ω 1的差值；第一决定单元201c根据第一计算单元201a计算出的差值，决定速度指令补正量。
如图6所示，第一计算单元201a可以是减法器，其计算出的差值例如是ω 1th-ω 1。当该差值小于0时，说明ω 1位于不稳定区之外，此时，第一决定单元201c决定输出的速度指令补正量为0；当该差值大于或等于0时，说明ω 1位于不稳定区之内，此时，第一决定单元201c决定输出该差值作为速度指令补正量。
此外，在本实施例中，如图6所述，速度指令补正量计算单元201还可以包括： 第一PI调节单元201b。该第一PI调节单元201b能够根据第一计算单元201a计算出的差值进行比例积分(PI)运算，输出PI运算结果。
其中，第一决定单元201c能够根据PI运算结果输出速度指令补正量，例如：在该PI运算结果小于0时，第一决定单元201c决定输出的速度指令补正量为0；在PI运算结果大于或等于0时，第一决定单元201c决定输出的速度指令补正量为该PI运算结果。
在图6中，通过设置第一PI调节单元201b，能够提高输出的速度指令补正量的稳定性。
根据本实施例，在本申请的速度指令补正量计算单元201能够不依赖于电机的滑差频率ωs来计算速度指令补正量，所以，可以避免电机转子的电阻变化对计算结果的影响。
在本实施例中，如图3所示，该控制装置30还可以具有转矩电流指令计算单元116。该转矩电流指令计算单元116可以基于输入的速度指令值、速度估算值计算单元111输出的速度估算值、以及速度指令补正量计算单元201输出的速度指令补正量，计算转矩电流指令。
图7是转矩电流指令计算单元116的一个示意图，如图7所示，该转矩电流指令计算单元116包括加法器116a、减法器116b以及第二PI调节器116c，由此，在转矩电流指令计算单元116中，速度指令指与速度指令补正量相加，该相加的结果减去速度估算值从而得到差值，该差值被第二PI调节器116c进行比例积分(PI)运算，该第二PI调节器116c进行PI运算的结果被作为转矩电流指令值输出到转矩电流调节单元102。
在本实施例中，关于图3的其它单元的说明可以参考专利文献1。
在本实施例中，在现有技术的基础上，修改了转矩电流指令计算单元116，增加了速度指令补正量调节单元201，定子频率计算单元202，磁链估算角计算单元203，以及磁链指令角计算单元204。由此，无需依赖电机的滑差频率ω s就能够将定子频率ω 1控制在稳定区以内(或者，不稳定区以外)，即，保证ω 1≥ω 1th，所以，提高了控制装置进行控制的稳定性。
下面以一个实例进一步说明本申请的控制装置的效果。在该实例中，感应电机的额定电压200V，额定功率2.2kW，电机正转，负载转矩(Torque)为-27.6Nm，稳定 区边界(即，预设的定子频率的下限值)ω 1th＝0.5Hz。
图8是对该实例中的电机进行矢量控制的一个示意图。在图8中，横轴为时间，单位是秒(S)，纵轴为频率，单位是赫兹(Hz)。其中，曲线801为根据本实施例的控制装置进行控制时电机M的转子的实际转速ω r，曲线802为根据本实施例的控制装置进行控制时电机M的定子频率ω 1，曲线803为电机M的滑差频率ω s，曲线804为根据专利文献1的控制装置进行控制时电机M的定子频率ω 1’。
如图8所示，在负载稳定的情况下，转子电阻Rr因为长时间运转温升增大，实际的滑差频率ω s的绝对值随之增大(如曲线803所示)。在根据本实施例的控制装置进行控制时，速度指令补正量通过PI自动调节，使实际速度ωr自动升高，保证了ω 1稳定在ω 1th，即，曲线802并没有下降，由此实现控制系统全程稳定。
与之相对，在使用专利文献1的控制装置进行控制时，在Rr因温升增大的情况下，由于仍然根据标称的Rr的值计算滑差频率，所以计算得到的滑差频率相对于实际的滑差频率偏小，进而，实际得到的ω 1’会低于稳定区边界ω 1th(如曲线804所示)，导致无法对电机进行稳定地控制。
实施例2
本申请实施例2提供一种感应电机的控制方法，该控制方法与实施例1的控制装置30对应。
图9是本实施例的感应电机的控制方法的一个示意图，如图9所示，该方法包括：
步骤901、计算磁化电压指令；
步骤902、计算转矩电压指令；
步骤903、根据预设的定子频率的下限值、定子频率、以及磁链估算角，计算磁链指令角度；以及
步骤904、根据所述磁化电压指令，所述转矩电压指令和所述磁链指令角度，计算用于对电机的定子运转进行控制的定子电压指令。
关于该控制方法的各步骤的说明，可以参考实施例1对相应单元的说明。此外，该控制方法还可以包括图9所述的步骤之外的步骤，也可以参考实施例1对控制装置30的各单元的说明。
根据本实施，无需依赖电机的滑差频率ω s就能够将定子频率ω 1控制在稳定区以 内(或者，不稳定区以外)，即，保证ω 1≥ω 1th，所以，提高了控制装置进行控制的稳定性。
结合本申请实施例描述的参数计算装置可直接体现为硬件、由处理器执行的软件模块或二者组合。这些硬件模块例如可利用现场可编程门阵列(FPGA)将这些软件模块固化而实现。
软件模块可以位于RAM存储器、闪存、ROM存储器、EPROM存储器、EEPROM存储器、寄存器、硬盘、移动磁盘、CD-ROM或者本领域已知的任何其它形式的存储介质。可以将一种存储介质耦接至处理器，从而使处理器能够从该存储介质读取信息，且可向该存储介质写入信息；或者该存储介质可以是处理器的组成部分。处理器和存储介质可以位于ASIC中。该软件模块可以存储在移动终端的存储器中，也可以存储在可插入移动终端的存储卡中。例如，若电子设备采用的是较大容量的MEGA-SIM卡或者大容量的闪存装置，则该软件模块可存储在该MEGA-SIM卡或者大容量的闪存装置中。
针对本实施例所描述的参数计算装置，可以实现为用于执行本申请所描述功能的通用处理器、数字信号处理器(DSP)、专用集成电路(ASIC)、现场可编程门阵列(FPGA)或其它可编程逻辑器件、分立门或晶体管逻辑器件、分立硬件组件、或者其任意适当组合。也可以实现为计算设备的组合，例如，DSP和微处理器的组合、多个微处理器、与DSP通信结合的一个或多个微处理器或者任何其它这种配置。
以上结合具体的实施方式对本申请进行了描述，但本领域技术人员应该清楚，这些描述都是示例性的，并不是对本申请保护范围的限制。本领域技术人员可以根据本申请的精神和原理对本申请做出各种变型和修改，这些变型和修改也在本申请的范围内。

Claims (10)

  1. 一种感应电机的控制装置，包括：
    磁化电流调节单元，用于计算磁化电压指令；
    转矩电流调节单元，用于计算转矩电压指令；
    磁链指令角度生成单元，其根据预设的定子频率的下限值ω 1th、定子频率ω 1、以及磁链估算角
    Figure PCTCN2019105160-appb-100001
    计算磁链指令角度；以及
    电机定子电压指令计算单元，其根据所述磁化电压指令，所述转矩电压指令和所述磁链指令角度，计算用于对电机的定子进行控制的定子电压指令。
  2. 如权利要求1所述的控制装置，其中，所述磁链指令角度生成单元包括：
    频率差计算单元，其计算所述预设的定子频率的下限值ω 1th与所述定子频率ω 1的差值；
    积分单元，其对该差值进行积分以得到磁链指令角补正量；以及
    补正单元，其使用所述磁链指令角补正量对所述磁链估算角
    Figure PCTCN2019105160-appb-100002
    进行补正，以得到所述磁链指令角度。
  3. 如权利要求2所述的控制装置，其中，
    所述磁链指令角补正量的最小值是0。
  4. 如权利要求1所述的控制装置，其中，所述控制装置还包括：
    磁链估算角计算单元，其针对基于电动机的定子电压信号、定子电流信号和速度估算值所计算的基于α-β坐标系的磁链
    Figure PCTCN2019105160-appb-100003
    和
    Figure PCTCN2019105160-appb-100004
    进行反正切计算，得到所述磁链估算角
    Figure PCTCN2019105160-appb-100005
  5. 如权利要求1所述的控制装置，其中，所述控制装置还包括：
    定子频率计算单元，其将所述磁链估算角
    Figure PCTCN2019105160-appb-100006
    相对于时间求导，得到所述定子频率ω 1。
  6. 如权利要求1所述的控制装置，其中，所述控制装置还包括：
    定子频率计算单元，其对基于流入所述电机的定子电流而得到的基于α-β坐标系的电流I a和I b做反正切得到电流矢量角∠I，再对所述电流矢量角∠I进行微分运算得到所述定子频率ω 1。
  7. 如权利要求1所述的控制装置，其中，所述控制装置还包括：
    速度指令补正量计算单元，其用于根据所述预设的定子频率的下限值ω 1th和所述定子频率ω 1，计算速度指令补正量，
    其中，所述速度指令补正量用于对速度指令值进行调整，从而计算所述转矩电压指令。
  8. 如权利要求7所述的控制装置，其中，速度指令补正量计算单元包括：
    第一计算单元，用于计算所述预设的定子频率的下限值ω 1th和所述定子频率ω 1的差值；
    第一决定单元，根据所述第一计算单元计算出的差值，决定所述速度指令补正量。
  9. 如权利要求8所述的控制装置，其中，所述速度指令补正量计算单元还包括：
    第一PI调节单元，其根据所述第一计算单元计算出的差值进行比例积分运算，输出比例积分运算结果；
    所述第一决定单元在所述比例积分运算结果小于0时，决定所述速度指令补正量为0，
    所述第一决定单元在所述PI运算结果大于或等于0时，决定所述速度指令补正量为所述比例积分运算结果。
  10. 一种感应电机的控制方法，包括：
    计算磁化电压指令；
    计算转矩电压指令；
    根据预设的定子频率的下限值、定子频率、以及磁链估算角，计算磁链指令角度；以及
    根据所述磁化电压指令，所述转矩电压指令和所述磁链指令角度，计算用于对电机的定子运转进行控制的定子电压指令。
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