WO2022261902A1 - 确定cllc变流器的同步整流导通时间的方法 - Google Patents

确定cllc变流器的同步整流导通时间的方法 Download PDF

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Publication number
WO2022261902A1
WO2022261902A1 PCT/CN2021/100691 CN2021100691W WO2022261902A1 WO 2022261902 A1 WO2022261902 A1 WO 2022261902A1 CN 2021100691 W CN2021100691 W CN 2021100691W WO 2022261902 A1 WO2022261902 A1 WO 2022261902A1
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current
resonant
diode
converter
coefficient
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French (fr)
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廖华
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Siemens Ltd China
Siemens AG
Siemens Corp
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Siemens Ltd China
Siemens AG
Siemens Corp
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Priority to US18/570,265 priority Critical patent/US12184188B2/en
Priority to EP21945488.1A priority patent/EP4336721B1/en
Priority to PCT/CN2021/100691 priority patent/WO2022261902A1/zh
Publication of WO2022261902A1 publication Critical patent/WO2022261902A1/zh
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/22—Conversion of DC power input into DC power output with intermediate conversion into AC
    • H02M3/24—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters
    • H02M3/28—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC
    • H02M3/325—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33573—Full-bridge at primary side of an isolation transformer
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/22—Conversion of DC power input into DC power output with intermediate conversion into AC
    • H02M3/24—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters
    • H02M3/28—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC
    • H02M3/325—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33576—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer
    • H02M3/33592—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer having a synchronous rectifier circuit or a synchronous freewheeling circuit at the secondary side of an isolation transformer
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0009—Devices or circuits for detecting current in a converter
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/01—Resonant DC/DC converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/22—Conversion of DC power input into DC power output with intermediate conversion into AC
    • H02M3/24—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters
    • H02M3/28—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC
    • H02M3/325—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal
    • H02M3/335—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/33569—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements
    • H02M3/33576—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only having several active switching elements having at least one active switching element at the secondary side of an isolation transformer
    • H02M3/33584—Bidirectional converters
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the present disclosure generally relates to the field of circuit technology, and more particularly, relates to a method for determining a synchronous rectification conduction time of a CLLC converter.
  • Bidirectional DC/DC (direct current/direct current) converters are used in many applications, such as energy storage, V2G, etc.
  • LLC converters are widely used due to their soft switching and high efficiency performance.
  • Figure 1 is a typical bidirectional LLC (CLLC) converter topology circuit, which can work in both forward and reverse directions.
  • synchronous rectification (Synchronous Rectification, SR) is generally used to reduce the body diode power loss of the power switch on the inactive side.
  • the body diode refers to the parasitic diode inside the power switch.
  • the CLLC converter topology circuit 100 in FIG. 1 includes the first to fourth power switches on the primary side, and the fifth to eighth power switches on the secondary side.
  • Each power switch can be regarded as including a MOSFET (Q1-Q8) and anti-parallel diodes (D1-D8).
  • the body diode is simply referred to as a diode.
  • the CLLC converter topology circuit 100 in Fig. 1 also includes the input capacitance C in of the primary side, the resonant capacitor C r1 and the resonant inductance L r1 , the excitation inductance L m , the transformer T 1 , the resonant inductance L r2 of the secondary side, and the resonant capacitor C r2 and the output capacitor C out .
  • a dedicated synchronous rectification integrated circuit can be used to automatically generate a drive pulse for the MOSFET by detecting the change in voltage or current when the diode starts to conduct.
  • CLLC due to the conflict with MOSFET gate drive, synchronous rectification IC cannot be used directly. Therefore, other methods must be adopted for bidirectional CLLC converters.
  • a current common method is shown in the simplified circuit structure diagram 200 of FIG. 2 , which includes a MOSFET current sensor 201 , a zero-crossing comparator 202 , a DSP timing register 203 and a gate driver 204 .
  • a current sensor 201 and a zero-crossing comparator circuit 202 are added.
  • the current sensor 201 samples the current 205 passing through the diode, and generates a synchronous rectification trigger signal through the zero-crossing comparator circuit 202 .
  • the DSP timing register 203 is then configured to generate a synchronous rectification pulse 206 according to the trigger signal.
  • This solution requires a high-speed current sensor and an additional zero-crossing comparator circuit, which increases the cost of the system.
  • FIG. 3 shows a structure diagram of a simplified circuit 300 of this solution.
  • the diode conduction time is measured under different operating frequencies and load conditions, and the diode conduction time look-up table is established.
  • the MCU samples the required parameters, such as the working frequency f s , the output current I o , the input voltage V in and the output voltage V out , etc., and finds the adjacent working point in the look-up table 301, and Use interpolation to get the required on-time.
  • the timing register 302 of the controller such as DSP, is configured to generate the synchronous rectification pulse 303 according to the trigger signal, and then generate the gate driving signal.
  • This method does not require a current sensor and a zero-crossing detection circuit, but requires pre-measurement under different working conditions to obtain a look-up table, and the process of measuring and obtaining the table is relatively cumbersome.
  • the present invention proposes a method for determining the conduction time of the synchronous rectification of the CLLC converter, without adding additional hardware, and is simple and easy to implement.
  • a method for determining a synchronous rectification on-time of a CLLC converter operating in an under-resonant operating mode comprising:
  • Step 1 Determine the transformer turns ratio and resonant frequency of the converter
  • Step 2 Use the simulation method to determine the peak current and output current of the secondary side diode, the excitation current of the primary side excitation inductance when the secondary side diode is turned off, and the excitation current of the primary side excitation inductance when the primary side drive signal is turned off. data set;
  • Step 3 measuring the current operating frequency and current output current of the converter
  • Step 4 According to the operating frequency and the output current, determine the first coefficient and the second coefficient by linear interpolation, wherein the first coefficient is equal to the ratio of the output current to the peak current multiplied by The second coefficient is equal to the ratio of the excitation current of the primary side excitation inductance when the secondary side diode is turned off to the excitation current of the primary side excitation inductance when the primary side drive signal is turned off;
  • Step 5 Measuring the initial value of the resonant current of the primary side resonant inductance
  • Step 6 Calculate the diode conduction time according to the transformer turns ratio, the resonant frequency, the initial value of the resonant current, the current output current, the first coefficient and the second coefficient.
  • the method further includes step seven: the method for determining the conduction time of synchronous rectification of the CLLC converter further includes: using the determined conduction time of the diode to configure the Timing registers in the controller of the converter.
  • step 3 to step 7 are repeatedly executed in each switching cycle of the CLLC converter to determine the diode conduction time of the current switching cycle.
  • the diode conduction time is:
  • a method for determining the conduction time of synchronous rectification is provided for CLLC operating in an under-resonance mode. This method can approximate the conduction time of the synchronous rectification by measuring the resonant current and the output current value.
  • the method of the invention does not need a high-bandwidth current sensor and a zero-crossing detection circuit, thus saving hardware cost and PCB space.
  • the method of the present invention does not need to measure the conduction time of the diode under different frequencies and loads in advance, thus saving a lot of work.
  • Figure 1 is a topological circuit diagram of a CLLC converter
  • Fig. 2 is a simplified circuit structure diagram capable of realizing synchronous rectification of a CLLC converter in the prior art
  • Fig. 3 is a simplified circuit structure diagram of synchronous rectification of a CLLC converter realized by using a look-up table in the prior art
  • FIGS. 4A-4C are working waveform diagrams of CLLC converters working in under-resonance mode
  • 5A-5C are working waveform diagrams of CLLC converters working in super-resonance mode
  • Fig. 6 is an equivalent circuit diagram of a resonant cavity of a CLLC converter
  • Figure 9 shows an enlarged waveform diagram of the resonant current and the excitation current
  • FIG. 10 is a flowchart of an exemplary process of a method for determining a synchronous rectification conduction time of a CLLC converter in an under-resonance mode of operation.
  • FIG. 11A and FIG. 11B respectively represent the simulation curve and the calculation result curve of the conduction time of the diode under different operating frequencies.
  • D1-D8 Diode C in : Input capacitance
  • MOSFET current sensor 202 Zero-crossing comparator
  • DSP timing register 204 Gate driver
  • Timing register 303 Synchronous rectification pulse
  • V ab Input voltage
  • V 1 Primary voltage
  • V 2 Secondary side voltage
  • N Transformer turns ratio
  • I Lmp Excitation current when the diode is turned off
  • ⁇ Diode conduction angle
  • T(ns) time (nanoseconds)
  • Po(W) output power (watts)
  • I Lmpp excitation current when the primary drive signal is turned off
  • the term “comprising” and its variants represent open terms meaning “including but not limited to”.
  • the term “based on” means “based at least in part on”.
  • the terms “one embodiment” and “an embodiment” mean “at least one embodiment.”
  • the term “another embodiment” means “at least one other embodiment.”
  • the terms “first”, “second”, etc. may refer to different or the same object. The following may include other definitions, either express or implied. Unless the context clearly indicates otherwise, the definition of a term is consistent throughout the specification.
  • a method for determining the conduction time of synchronous rectification of a bidirectional CLLC converter in an under-resonant operating mode is proposed.
  • 4A-4C are working waveforms of the bidirectional CLLC converter of FIG. 1 working in the under-resonant mode, and its working frequency fs ⁇ resonant frequency fr.
  • the solid line represents the driving waveform of MOSFET Q1
  • the dotted line represents the driving waveform of MOSFET Q2. It can be understood that the driving waveforms of Q1 and Q4 are the same, and the driving waveforms of Q2 and Q3 are the same.
  • the solid line represents the current waveform of the resonant inductance L r1 on the primary side
  • the dotted line represents the current waveform of the magnetizing inductance L m .
  • the solid line represents the current waveform of the secondary diode D5
  • the dashed line represents the current waveform of the secondary diode D7.
  • D5 and D7 work alternately, and each conducts for half a period. That is to say, the current waveforms of D5-D8 are all the same, therefore, the following derivation takes D5 as an example to calculate its conduction time.
  • D5 For forward work, it is to calculate the conduction time of diodes D5-D8 on the secondary side; for reverse work, it is to calculate the conduction time of primary side D1-D4.
  • 5A-5C are working waveforms of a bidirectional CLLC converter working in a super-resonant mode, where the working frequency fs>resonant frequency fr.
  • the curves in Figures 5A-5C and the curves in Figures 4A-4C respectively represent the same meanings, and will not be described in detail here.
  • the bidirectional LLC circuit is used as an example for illustration, that is, the input is on the left and the output is on the right.
  • the bidirectional LLC circuit is working, in principle, it is only necessary to provide a drive pulse G1-G4 for the first MOSFET tube to the fourth MOSFET tube Q1-Q4 of the first power switch to the fourth power switch on the primary side, respectively, and the second MOSFET tube on the secondary side
  • the fifth to eighth MOSFETs Q5 to Q8 of the fifth to eighth power switches do not need to provide pulse work, and only the fifth to eighth diodes D5 to D8 are required to work.
  • the present invention provides a method for determining the turn-on time of D5-D8 during synchronous rectification.
  • the on-time of synchronous rectification is determined by the driving signal of the primary side, but the off-time should be determined by the conduction time of the diode. It can be seen from Figures 5A-5C that when the operating frequency of the CLLC converter is higher than the normal resonant frequency fr, and the CLLC converter circuit operates in super-resonant mode, the conduction of the fifth diode D5 on the secondary side
  • the on-time is basically synchronous with the driving pulse G1 of the first transistor Q1 on the primary side. In this case, it is sufficient to directly supply the driving pulse G1 on the primary side to the fifth transistor Q5 on the secondary side.
  • the present invention mainly aims at In the resonant working mode, how to determine the turn-off time of the diode, so as to determine the turn-on time of the diode.
  • Figure 6 is the resonant cavity equivalent circuit of the bidirectional CLLC converter shown in Figure 1, where L r1 , C r1 and L r2 , C r2 are the resonant inductance and capacitor of the primary side and secondary side respectively, and L m is the transformer
  • the excitation inductance N is the turns ratio of the transformer
  • V ab is the input voltage of the circuit
  • V 1 and V 2 are the voltages of the primary side and the secondary side of the transformer respectively
  • i Lr1 is the resonant current
  • i Lm is the excitation current.
  • the resonator input voltage can be expressed as:
  • ⁇ s is the current operating angular frequency of the CLLC circuit (in rad/s)
  • ⁇ r is the resonant angular frequency of the CLLC circuit (in rad/s)
  • ⁇ is a variable representing the shape of the diode current wave.
  • 7A-7D are detailed working waveform diagrams of the CLLC converter working in the under-resonance mode.
  • Fig. 7A the solid line represents the driving waveform G1 of MOSFET Q1, and the dotted line represents the driving waveform G2 of MOSFET Q2;
  • Fig. 7B shows the waveform of the input voltage V ab ; in Fig.
  • the solid line represents the primary side The waveform of the resonant current i Lr1 of the resonant inductance L r1 , the dotted line represents the waveform of the excitation current i Lm of the excitation inductance L m ; in Figure 7D, the solid line represents the current i D5 of the secondary diode D5, and the dotted line represents the current i D5 of the secondary diode D7
  • the current i D7 , I peak is the peak current of i D5 .
  • the current i D5 of diode D5 during conduction can be expressed as:
  • I peak is the peak current of the diode i D5
  • ⁇ is the conduction angle of the diode.
  • the excitation current i Lm of the excitation inductance L m of the transformer can be expressed as:
  • i Lm represents the excitation current
  • i Lm0 is the initial value of the excitation current
  • ⁇ I Lm is the variation of the excitation current at the end of the ⁇ angle.
  • V ab and V 2 are considered constant during the on-time, and taking the derivative of the above equation, we get:
  • the conduction angle duty cycle can be expressed as:
  • I o is the output current (also known as the load current)
  • I Lmp is the excitation current of the primary side magnetizing inductance Lm when the secondary side diode is turned off
  • I Lmpp is the excitation current of the primary side magnetizing inductance Lm when the primary side driving signal is turned off current.
  • the coefficients can be obtained by interpolation with
  • Figure 8C is the normalized frequency point fs1 of 6 different operating frequencies fs under the minimum load, read 6 groups of I o and I peak through simulation, and calculate Six k ipeak curves are obtained, as shown in FIG. 8C ; for the maximum load, the k ipeak curve shown in FIG. 8D is also obtained through simulation. Then, k ipeak under any load and any frequency must be a certain value in the middle of these two curves, and this k ipeak can be calculated by linear interpolation.
  • Fig. 9 shows enlarged waveform diagrams of the resonant current (solid line) and the excitation current (dashed line).
  • the diode conduction time T cndc can be expressed as:
  • I Lm0 in the formula is the initial value of the excitation current at the beginning of each cycle, and the initial value of the excitation current and the resonance current at the beginning of each cycle are the same, as can be seen in Figure 9, resonance current (solid line) and excitation The currents (dotted line) all vary from the same initial value, which is characteristic of LLC circuits. Therefore, the initial value of the resonant current of the primary side resonant inductance can be measured as the value of I Lm0 .
  • Sampling the average output current Io and the resonant current at the falling edge of the primary drive signal can approximate the diode conduction time.
  • FIG. 10 is a flowchart of an exemplary process of a method 1000 for determining a synchronous rectification conduction time in an under-resonance operation mode of a CLLC converter.
  • step S1001 the turns ratio N and the resonant frequency f r of the transformer of the converter are determined.
  • step S1002 using the method of simulation, the peak current Ipeak and the output current Io of the secondary side diode, the excitation current I Lmp of the excitation inductance Lm when the secondary side diode is turned off, and the primary side drive signal off can be obtained.
  • step S1003 measure the current operating frequency f s and the current output current I o of the converter
  • step S1004 according to the current operating frequency f s and the current output current I o , the first coefficient k ipeak and the second coefficient k iLmp are determined by linear interpolation.
  • the first coefficient k ipeak is defined as the ratio of the output current to the peak current multiplied by which is
  • the second coefficient k iLmp is defined as the ratio of the excitation current I Lmp of the excitation inductance Lm when the secondary side diode is turned off and the excitation current I Lmpp of the excitation inductance Lm when the primary side drive signal is turned off, namely
  • the current first and second coefficients can be determined in the data sets (I peak , I o ) and (I Lmp , I Lmpp ) determined in step S1002 by means of linear interpolation.
  • step S1005 the initial value of the resonant current of the primary side resonant inductance is measured.
  • the initial value I Lm0 of the exciting current is equal to the initial value of the resonant current.
  • step S1006 according to the transformer turns ratio N, the resonant frequency f r , the initial value of the resonant current I Lm0 , the output current I o , the first coefficient k ipeak and the second Coefficient k iLmp to calculate the diode conduction time.
  • the conduction time can be used to configure the timing register in the controller of the converter, so that the MOSFET can be controlled to conduct synchronously with the diode.
  • steps S1003 , S1004 , S1005 , S1006 and S1007 are repeatedly executed in each switching cycle of the CLLC converter to determine the diode conduction time of each switching cycle.
  • the conduction time of the diode in each switching cycle can be obtained, so as to ensure that the MOSFET can be turned on synchronously with the diode in each switching cycle.
  • the horizontal axis represents the output power Po in watts (W)
  • the vertical axis represents the time T in nanoseconds (ns)
  • the solid line represents the actual conduction time of the diode
  • the dotted line represents the simulation of the diode on-time. It can be seen from FIGS. 11A and 11B that as the output power Po increases, the calculation error decreases; and the closer the operating frequency is to the resonance frequency, the higher the accuracy.
  • a method for determining the conduction time of synchronous rectification is provided for CLLC operating in an under-resonance mode. This method can approximate the conduction time of the synchronous rectification by measuring the resonant current and the output current value.
  • the method of the present invention does not need high-bandwidth current and zero-crossing detection circuit, thus saving hardware cost and PCB space.
  • this method saves a lot of work by eliminating the need to pre-measure the diode conduction time at different frequencies and loads.
  • the device structures described in the above embodiments may be physical structures or logical structures, that is, some units may be implemented by the same physical entity, or some units may be respectively implemented by multiple physical entities, or may be implemented by multiple physical entities. Certain components in individual devices are implemented together.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Dc-Dc Converters (AREA)
  • Rectifiers (AREA)

Abstract

本公开涉及确定CLLC变流器的同步整流导通时间的方法，所述CLLC变流器在欠谐振工作模式下工作，所述方法包括：确定变流器的变压器匝数比和谐振频率；利用仿真的方法，确定副边二级管的峰值电流和输出电流以及原边励磁电感在副边二极管关断时和原边驱动信号关断时的励磁电流的数据集；测量所述变流器的当前工作频率和当前输出电流；根据所述工作频率和所述输出电流，通过线性插值法确定第一系数和第二系数；测量原边谐振电感的谐振电流初始值；以及根据所述变压器匝数比、所述谐振频率、所述谐振电流初始值、所述当前输出电流、所述第一系数和所述第二系数来计算二极管导通时间。

Description

确定CLLC变流器的同步整流导通时间的方法 技术领域
本公开通常涉及电路技术领域，更具体地，涉及确定CLLC变流器的同步整流导通时间的方法。
背景技术
双向DC/DC(直流/直流)变流器用在很多应用中，比如能量存储，V2G等。在不同的DC/DC拓扑中，LLC变流器由于其软开关和高效率性能得到了广泛的使用。图1是典型的双向LLC(CLLC)变流器拓扑电路，该电路可以在正向和反向两个方向工作。
为了提高电路效率，一般使用同步整流(Synchronous Rectification，SR)来减小位于未激活侧的功率开关的体二极管功率损耗。
体二极管是指功率开关内部的寄生二极管，图1中的CLLC变流器拓扑电路100包括原边的第一功率开关至第四功率开关，副边的第五功率开关至第八功率开关。每一个功率开关可以看做包括一个MOSFET管(Q1-Q8)和反向并联的二极管(D1-D8)。下文中将体二极管简称为二极管。
图1中的CLLC变流器拓扑电路100还包括原边的输入电容C in、谐振电容C r1和谐振电感L r1、励磁电感L m、变压器T 1、副边的谐振电感L r2、谐振电容C r2以及输出电容C out。
对于单向LLC变流器，可以通过检测二极管开始导通时电压或电流的变化，使用专用同步整流集成电路来自动生成用于MOSFET管的驱动脉冲。但是对于CLLC的情况，由于与MOSFET栅极驱动的冲突，不能直接使用同步整流集成电路。因此，针对双向CLLC变流器必须采用其它方法。
目前一种常见的方法如图2的简化电路结构图200所示，包括一个MOSFET电流传感器201、过零比较器202、DSP定时寄存器203和栅极驱动器204。在该电路中增加了电流传感器201和过零比较器电路202。电流传感器201对通过二极管的电流205进行采样，并且通过过零比较器电路202生成同步整流触发信号。然后DSP定时寄存器203被配置为根据触发信号生成同步整流脉冲206。这种解决方案需要高速电流传感器和额外的过零比较器电路，这样增加了系统的成本。
另一种方法是在MCU中使用查询表，图3示出了这种解决方案的简化电路300的结构图。首先在不同工作频率和负载条件下测量二极管导通时间，建立的二极管导通时间查询表。当在实际工作条件下，MCU对需要的参数进行采样，例如工作频率f s、输出电流I o、输入电压V in和输出电压V out等，并且在查询表301中找到临近的工作点，并且使用内插法得到需要的导通时间。根据所得到的导通时间，控制器，例如DSP，的定时寄存器302被配置为根据触发信号生成同步整流脉冲303，进而生成栅极驱动信号。这种方法不需要电流传感器和过零检测电路，但是需要预先在不同工作条件下进行测量来获得查询表，并且测量获取表的过程也较为繁琐。
发明内容
在下文中给出关于本发明的简要概述，以便提供关于本发明的某些方面的基本理解。应当理解，这个概述并不是关于本发明的穷举性概述。它并不是意图确定本发明的关键或重要部分，也不是意图限定本发明的范围。其目的仅仅是以简化的形式给出某些概念，以此作为稍后论述的更详细描述的前序。
有鉴于此，本发明提出了一种用于确定CLLC变流器的同步整流导通时间的方法，无需增加额外的硬件，并且简单易行。
根据本公开的一个方面，提供了一种用于确定CLLC变流器的同步整流导通时间的方法，所述变流器在欠谐振工作模式下工作，所述方法包括：
步骤一：确定变流器的变压器匝数比和谐振频率；
步骤二：利用仿真的方法，确定副边二级管的峰值电流和输出电流以及副边二极管关断时原边励磁电感的励磁电流和原边驱动信号关断时原边励磁电感的励磁电流的数据集；
步骤三：测量所述变流器的当前工作频率和当前输出电流；
步骤四：根据所述工作频率和所述输出电流，通过线性插值法确定第一系数和第二系数，其中，所述第一系数等于所述输出电流与所述峰值电流的比值再乘以
Figure PCTCN2021100691-appb-000001
所述第二系数等于所述副边二极管关断时所述原边励磁电感的励磁电流和所述原边驱动信号关断时所述原边励磁电感的励磁电流的比值；
步骤五：测量所述原边谐振电感的谐振电流初始值；以及
步骤六：根据所述变压器匝数比、所述谐振频率、所述谐振电流初始值、所述当前输出电流、所述第一系数和所述第二系数来计算二极管导通时间。
可选地，在上述方面的一个示例中，所述方法还包括步骤七：用于确定CLLC变流器的同步整流导通时间的方法还包括：用所确定的二极管导通时间来配置所述变流器的控制器中的定时寄存器。
可选地，在上述方面的一个示例中，在所述CLLC变流器的每个开关周期重复执行所述步骤三至所述步骤七来确定当前开关周期的二极管导通时间。
可选地，在上述方面的一个示例中，所述二极管导通时间为：
Figure PCTCN2021100691-appb-000002
根据本发明的实施例，针对CLLC工作在欠谐振模式下，提供了一种确定同步整流导通时间的方法。本方法通过测量谐振电流和输出电流值，可以近似的计算同步整流导通时间。
本发明的方法不需要高带宽电流传感器和过零检测电路，这样节省了硬件成本和PCB空间。
本发明的方法无需预先测量不同频率和负载下的二极管导通时间，因此节省了大量工作。
附图说明
参照下面结合附图对本发明实施例的说明，会更加容易地理解本发明的以上和其它目的、特点和优点。附图中的部件只是为了示出本发明的原理。在附图中，相同的或类似的技术特征或部件将采用相同或类似的附图标记来表示。附图中：
图1是CLLC变流器的拓扑电路图；
图2是现有技术中能够实现CLLC变流器的同步整流的简化电路结构图；
图3是现有技术中使用查询表实现CLLC变流器的同步整流的简化电路结构图；
图4A-4C是CLLC变流器在欠谐振模式下工作的工作波形图；
图5A-5C是CLLC变流器在超谐振模式下工作的工作波形图；
图6是CLLC变流器的谐振腔等效电路图；
图7A-7D是CLLC变流器在欠谐振模式下工作的详细工作波形图；
图8A-8D是在不同工作频率下，分别在最小负载和最大负载的情况下的第一系数 k ipeak和第二系数k iLmp的仿真曲线；
[根据细则91更正 31.08.2021]　
图9示出了谐振电流和励磁电流的放大波形图；
[根据细则91更正 31.08.2021]　
图10为用于确定CLLC变流器在欠谐振工作模式下同步整流导通时间的方法的示例性过程的流程图；以及
[根据细则91更正 31.08.2021]　
图11A和图11B分别表示在不同工作频率下二极管导通时间的仿真曲线和计算结果曲线。
其中，附图标记如下：
100：CLLC变流器拓扑电路              Q1-Q8：MOSFET管
D1-D8：二极管                        C in：输入电容
C r1：谐振电容                        L r1：谐振电感
L m：励磁电感                         T 1：变压器
L r2：谐振电感                         C r2：谐振电容
C out：输出电容                       200：同步整流电路结构图
201：MOSFET电流传感器                202：过零比较器
203：DSP定时寄存器                   204：栅极驱动器
205：电流                            206：同步整流脉冲
300：同步整流电路结构图              301：查询表
302：定时寄存器                      303：同步整流脉冲
V ab：输入电压                        V 1：原边电压
V 2：副边电压                         N：变压器匝数比
G1、G2：驱动波形                     i Lr1：谐振电流
i Lm：励磁电流                        i D5：二极管D5的电流
i D7：二极管D7的电流                  I peak：副边二极管峰值电流
i Lm0：励磁电流的初始值               ΔI Lm：励磁电流的变化量
I Lmp：二极管关断时的励磁电流         Φ：二极管导通角
k ipeak：第一系数                      k iLmp：第二系数
fs1：归一化工作频率                 ΔI Lmp：励磁电流的变化量
1000：确定同步整流导通时间的方法    S1001、S1002、S1003、S1004、S1005、
                                    S1006、S1007：步骤
T(ns)：时间(纳秒)                   Po(W)：输出功率(瓦)
I Lmpp：原边驱动信号关断时的励磁电流
具体实施方式
现在将参考示例实施方式讨论本文描述的主题。应该理解，讨论这些实施方式只是为了使得本领域技术人员能够更好地理解从而实现本文描述的主题，并非是对权利要求书中所阐述的保护范围、适用性或者示例的限制。可以在不脱离本公开内容的保护范围的情况下，对所讨论的元素的功能和排列进行改变。各个示例可以根据需要，省略、替代或者添加各种过程或组件。例如，所描述的方法可以按照与所描述的顺序不同的顺序来执行，以及各个步骤可以被添加、省略或者组合。另外，相对一些示例所描述的特征在其它例子中也可以进行组合。
如本文中使用的，术语“包括”及其变型表示开放的术语，含义是“包括但不限于”。术语“基于”表示“至少部分地基于”。术语“一个实施例”和“一实施例”表示“至少一个实施例”。术语“另一个实施例”表示“至少一个其他实施例”。术语“第一”、“第二”等可以指代不同的或相同的对象。下面可以包括其他的定义，无论是明确的还是隐含的。除非上下文中明确地指明，否则一个术语的定义在整个说明书中是一致的。
根据本发明的实施例，提出了一种在欠谐振工作模式下双向CLLC变流器的同步整流导通时间的确定方法。
图4A-4C是图1的双向CLLC变流器在欠谐振模式下工作的工作波形，其工作频率fs<谐振频率fr。
其中，在图4A中，实线表示MOSFET管Q1的驱动波形，虚线表示MOSFET管Q2的驱动波形，可以理解，Q1和Q4的驱动波形相同，Q2和Q3的驱动波形相同。
在图4B中，实线表示原边的谐振电感L r1的电流波形，虚线表示励磁电感L m的电流波形。
图4C中，实线表示副边二极管D5的电流波形，虚线表示副边二极管D7的电流波形。
在电路正向工作时，二极管D5和二极管D8的电流相同，二极管D7和二极管D6的电流相同，D5和D7是交替工作，各导通半个周期。也就是说D5～D8的电流波形都是一样的，因此，在下文中的推导以D5为例计算其导通时间。对正向工作来说，是计算副边的二极管D5～D8的导通时间；对反向工作来说，是计算原边D1～D4的导通时间。
图5A-5C是双向CLLC变流器在超谐振模式下工作的工作波形，其中工作频率fs>谐振频率fr。图5A-5C中的曲线与图4A-4C中的曲线分别代表相同的含义，在此不再详 述。
在本说明书中，都以双向LLC电路工作在正向为例进行说明，即左边是输入，右边是输出。在双向LLC电路工作时，原则上只需给原边的第一功率开关至第四功率开关的第一MOSFET管至第四MOSFET管Q1～Q4分别提供一个驱动脉冲G1～G4，副边的第五功率开关至第八功率开关的第五MOSFET管至第八MOSFET管Q5～Q8无需提供脉冲工作，只让第五二极管至第八二极管D5～D8工作即可。但是实际上因为D5～D8导通的导通损耗很大，所以通常在第五MOSFET管至第八MOSFET管D5～D8导通时同步给对应的第五MOSFET管至第八MOSFET管Q5～Q8提供驱动脉冲，让Q5～Q8和D5～D8同步导通，以降低损耗，这种方法就是所说的同步整流(Synchronous Rectification，SR)。而为了保持Q5～Q8和D5～D8同步导通，本发明提供了一种如何在进行同步整流时确定D5～D8导通的时间的方法。
同步整流导通时间由原边的驱动信号决定，但是关断时间应该由二极管的导通时间决定。从图5A-5C可以看到，当CLLC变流器的工作频率高于正常的谐振频率fr时，CLLC变流器电路在超谐振模式下工作时，副边的第五二极管D5的导通时间和原边的第一晶体管Q1的驱动脉冲G1基本是同步的，在这种情况下，把原边的驱动脉冲G1直接给副边的第五晶体管Q5就可以了。但是，在图4A-4C所示的欠谐振工作模式下，副边的第五二极管D5电流在原边的G1驱动脉冲关断前就已经下降到0了，因此，本发明主要针对在欠谐振工作模式下，如何确定二极管的关断时间，从而可以确定二极管的导通时间。
图6是图1所示的双向CLLC变流器的谐振腔等效电路，其中，L r1，C r1和L r2、C r2分别是原边和副边的谐振电感和电容器，L m是变压器励磁电感，N是变压器匝数比，V ab是电路的输入电压，V 1和V 2分别是变压器原边和副边的电压，i Lr1是谐振电流，i Lm是励磁电流。
则谐振腔输入电压可以表示为：
Figure PCTCN2021100691-appb-000003
其中，ω s是CLLC电路当前的工作角频率(单位rad/s)，ω r是CLLC电路的谐振角频率(单位rad/s)，θ是表示二极管电流波形状的变量。
图7A-7D是CLLC变流器在欠谐振模式下工作的详细工作波形图。
其中，在图7A中，实线表示MOSFET管Q1的驱动波形G1，虚线表示MOSFET管Q2的驱动波形G2；图7B示出了输入电压V ab的波形；在图7C中，实线表示原边的谐振电感L r1的谐振电流i Lr1的波形，虚线表示励磁电感L m的励磁电流i Lm的波形；图7D 中，实线表示副边二极管D5的电流i D5，虚线表示副边二极管D7的电流i D7，I peak是i D5的峰值电流。
在导通期间二极管D5的电流i D5可以表示为：
Figure PCTCN2021100691-appb-000004
其中，I peak是二极管i D5的峰值电流，Φ是二极管导通角。变压器的励磁电感L m的励磁电流i Lm可以表示为：
Figure PCTCN2021100691-appb-000005
其中，i Lm表示励磁电流，i Lm0是励磁电流的初始值，ΔI Lm是到Φ角结束时励磁电流的变化量。
由上可以得到如下公式来表示V ab：
Figure PCTCN2021100691-appb-000006
V ab和V 2在导通时间期间被认为是常数，对上面的等式求导，可以得到：
Figure PCTCN2021100691-appb-000007
由上式，导通角占空比可以表示为：
Figure PCTCN2021100691-appb-000008
现在选择时间位于
Figure PCTCN2021100691-appb-000009
处，可以得到导通角占空比的简化形式：
Figure PCTCN2021100691-appb-000010
因为I peak和ΔI Lm不能直接得到，因此引入两个系数
Figure PCTCN2021100691-appb-000011
和
Figure PCTCN2021100691-appb-000012
Figure PCTCN2021100691-appb-000013
Figure PCTCN2021100691-appb-000014
其中，I o是输出电流(也称为负载电流)，I Lmp是副边二极管关断时原边励磁电感Lm的励磁电流，I Lmpp是原边驱动信号关断时原边励磁电感Lm的励磁电流。
通过在不同的频率和负载下进行仿真比较，可以采用插值法得到系数
Figure PCTCN2021100691-appb-000015
和
Figure PCTCN2021100691-appb-000016
具体来说，首先针对最小负载，取6个不同工作频率fs的归一化的频率点fs1，通过仿真读取6组I Lmp和I Lmpp，计算I Lmp/I Lmpp得到6个
Figure PCTCN2021100691-appb-000017
如图8A的曲线所示；再针对最大负载，同样通过仿真获得图8B所示的k iLmp曲线。那么，在任何负载、任意频率下的k iLmp，一定是位于这两个曲线中间的某个值，通过线性插值可以计算出来这个k iLmp。
类似地，图8C是在最小负载下，取6个不同工作频率fs的归一化的频率点fs1，通过仿真读取6组I o和I peak，计算
Figure PCTCN2021100691-appb-000018
得到6个k ipeak，如图8C所示的曲线；在针对最大负载，同样通过仿真获得图8D所示的k ipeak曲线。那么，在任何负载、任意频率下的k ipeak，一定是位于这两个曲线中间的某个值，通过线性插值可以计算出来这个k ipeak。
可以理解，仿真的过程中，可以根据需要选取适当数量的频率点，而不限于6个频率点。
图9示出了谐振电流(实线)和励磁电流(虚线)的放大波形图。
从图9中虚线所表示的励磁电感L m的电流波形可以理解，励磁电感L m在工作过程中，电流必须是正负对称的波形，总平均值为0，否则电感就不是工作在稳定状态。因此，可以近似得到I Lmpp＝-I Lm0，即I Lmpp和I Lm0具有相同的绝对值。
通过以上分析，可以将二极管导通时间T cndc表示为：
Figure PCTCN2021100691-appb-000019
需要说明的是，变压器的励磁电流是不可测的，能测的只是谐振电流。公式中的I Lm0是在每个周期开始时励磁电流的初始值，而励磁电流和谐振电流在每个周期开始时的初始值是相同的，可以参见图9，谐振电流(实线)和励磁电流(虚线)都从同一个初始值开始变化，这是LLC电路的特点。所以，可以通过测量原边谐振电感的谐振电流初始值来作为I Lm0的值。
对平均输出电流Io和谐振电流在原边驱动信号的下降边沿进行采样，则可以近似计算出二极管导通时间。
下面将结合图10来具体描述根据本公开的实施例的用于确定CLLC变流器在欠谐振工作模式下同步整流导通时间的方法。图10为用于确定CLLC变流器在欠谐振工作模式下同步整流导通时间的方法1000的示例性过程的流程图。
首先，在步骤S1001中，确定变流器的变压器的匝数比N和谐振频率f r。
接着，在步骤S1002中，利用仿真的方法，可以获得副边二级管的峰值电流I peak和输出电流I o以及副边二极管关断时励磁电感Lm的励磁电流I Lmp和原边驱动信号关断时励磁电感Lm的励磁电流I Lmpp的数据集(I peak，I o)以及(I Lmp，I Lmpp)。
具体的仿真过程，可以参照上面结合图8A-8D所描述的方法，在此不再详述。
接着，在步骤S1003中，测量所述变流器的当前工作频率f s和当前输出电流I o；
接着，在步骤S1004中，根据当前工作频率f s和当前输出电流I o，通过线性插值法确定第一系数k ipeak和第二系数k iLmp。
所述第一系数k ipeak定义为所述输出电流与所述峰值电流的比值再乘以
Figure PCTCN2021100691-appb-000020
即
Figure PCTCN2021100691-appb-000021
所述第二系数k iLmp定义为副边二极管关断时励磁电感Lm的励磁电流I Lmp和原边驱动信号关断时励磁电感Lm的励磁电流I Lmpp的比值，即
Figure PCTCN2021100691-appb-000022
通过线性插值法，可以在步骤S1002中所确定的数据集(I peak，I o)以及(I Lmp，I Lmpp)中确定当前的第一系数和第二系数。
接下来，在步骤S1005中，测量原边谐振电感的谐振电流初始值，如上所述，励磁电流的初始值I Lm0等于这个谐振电流初始值。
接下来，在步骤S1006中，根据变压器匝数比N、所述谐振频率f r、所述谐振电流初始值I Lm0、所述输出电流I o、所述第一系数k ipeak和所述第二系数k iLmp来计算二极管导通时间。
在确定了二极管的导通时间之后，在步骤S1007中，可以用这个导通时间来配置变流器的控制器中的定时寄存器，从而可以控制MOSFET管与二极管同步导通。
优选地，在CLLC变流器的每个开关周期重复执行步骤S1003、S1004、S1005、S1006和S1007，来确定每一个开关周期的二极管导通时间。
通过这样的方式，可以得到每个开关周期的二极管导通时间，确保在每个开关周期MOSFET管都可以与二极管同步导通。
下面参照图11说明一个具体示例，来验证根据本发明的方法确定二极管导通时间的 准确性。
一个双向CLLC电路的参数为：L r1＝15uH，L m＝75uH，C r1＝42nF，L r2＝3.75uH，C r2＝169nF，N＝2，谐振频率fr＝200kHz，V in＝800V。利用本发明的方法，可以得到在不同工作频率下二极管导通时间的仿真和计算结果曲线。
图11A是在工作频率fs＝180kHz下的曲线，图11B是在工作频率fs＝190kHz下的曲线。在图11A和11B中，横轴表示输出功率Po，单位是瓦(W)，纵轴表示时间T，单位是纳秒(ns)，实线表示二极管的实际导通时间，虚线表示二极管的仿真导通时间。从图11A和11B可以看到，随着输出功率Po的增加，计算误差减小；并且工作频率越接近谐振频率，准确度越高。
根据本发明的实施例，针对CLLC工作在欠谐振模式下，提供了一种确定同步整流导通时间的方法。本方法通过测量谐振电流和输出电流值，可以近似的计算同步整流导通时间。
与传统的电流传感器方案相比，本发明的方法不需要高带宽电流和过零检测电路，这样节省了硬件成本和PCB空间。
与传统的查询表方案相比，这种方法无需预先测量不同频率和负载下的二极管导通时间，因此节省了大量工作。
上文对本说明书特定实施例进行了描述。其它实施例在所附权利要求书的范围内。在一些情况下，在权利要求书中记载的动作或步骤可以按照不同于实施例中的顺序来执行并且仍然可以实现期望的结果。另外，在附图中描绘的过程不一定要求示出的特定顺序或者连续顺序才能实现期望的结果。在某些实施方式中，多任务处理和并行处理也是可以的或者可能是有利的。
上述各流程和各系统结构图中不是所有的步骤和单元都是必须的，可以根据实际的需要忽略某些步骤或单元。上述各实施例中描述的装置结构可以是物理结构，也可以是逻辑结构，即，有些单元可能由同一物理实体实现，或者，有些单元可能分别由多个物理实体实现，或者，可以由多个独立设备中的某些部件共同实现。
上面结合附图阐述的具体实施方式描述了示例性实施例，但并不表示可以实现的或者落入权利要求书的保护范围的所有实施例。在整个本说明书中使用的术语“示例性”意味着“用作示例、实例或例示”，并不意味着比其它实施例“优选”或“具有优势”。出于提供对所描述技术的理解的目的，具体实施方式包括具体细节。然而，可以在没有这些具体细节的情况下实施这些技术。在一些实例中，为了避免对所描述的实施例的概 念造成难以理解，公知的结构和装置以框图形式示出。
本公开内容的上述描述被提供来使得本领域任何普通技术人员能够实现或者使用本公开内容。对于本领域普通技术人员来说，对本公开内容进行的各种修改是显而易见的，并且，也可以在不脱离本公开内容的保护范围的情况下，将本文所定义的一般性原理应用于其它变型。因此，本公开内容并不限于本文所描述的示例和设计，而是与符合本文公开的原理和新颖性特征的最广范围相一致。
以上所述仅为本发明的较佳实施例而已，并不用以限制本发明，凡在本发明的精神和原则之内，所作的任何修改、等同替换、改进等，均应包含在本发明的保护范围之内。

Claims (4)

  1. 用于确定CLLC变流器的同步整流导通时间的方法，所述CLLC变流器在欠谐振工作模式下工作，所述方法包括：
    步骤一：确定变流器的变压器匝数比和谐振频率；
    步骤二：利用仿真的方法，确定副边二级管的峰值电流和输出电流以及副边二极管关断时的原边励磁电感的励磁电流和原边驱动信号关断时原边励磁电感的励磁电流的数据集；
    步骤三：测量所述变流器的当前工作频率和当前输出电流；
    步骤四：根据所述工作频率和所述输出电流，通过线性插值法确定第一系数和第二系数，其中，所述第一系数等于所述输出电流与所述峰值电流的比值再乘以
    Figure PCTCN2021100691-appb-100001
    所述第二系数等于所述副边二极管关断时所述原边励磁电感的励磁电流和所述原边驱动信号关断时所述原边励磁电感的励磁电流的比值；
    步骤五：测量原边谐振电感的谐振电流初始值；以及
    步骤六：根据所述变压器匝数比、所述谐振频率、所述谐振电流初始值、所述当前输出电流、所述第一系数和所述第二系数来计算二极管导通时间。
  2. 如权利要求1所述的方法，还包括步骤七：用所确定的二极管导通时间来配置所述变流器的控制器中的定时寄存器。
  3. 如权利要求1或2所述的方法，其中，在所述CLLC变流器的每个开关周期重复执行所述步骤三至所述步骤七来确定每一个开关周期的二极管导通时间。
  4. 如权利要求1-3中任意一项所述的方法，其中，所述二极管导通时间为：
    Figure PCTCN2021100691-appb-100002
PCT/CN2021/100691 2021-06-17 2021-06-17 确定cllc变流器的同步整流导通时间的方法 Ceased WO2022261902A1 (zh)

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