WO2023093216A1 - 开关电路及电子设备 - Google Patents

开关电路及电子设备 Download PDF

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Publication number
WO2023093216A1
WO2023093216A1 PCT/CN2022/117882 CN2022117882W WO2023093216A1 WO 2023093216 A1 WO2023093216 A1 WO 2023093216A1 CN 2022117882 W CN2022117882 W CN 2022117882W WO 2023093216 A1 WO2023093216 A1 WO 2023093216A1
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WO
WIPO (PCT)
Prior art keywords
voltage
electron mobility
high electron
mobility transistor
gate
Prior art date
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Ceased
Application number
PCT/CN2022/117882
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English (en)
French (fr)
Inventor
张紫淇
孙霓
黄松
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Honor Device Co Ltd
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Honor Device Co Ltd
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Filing date
Publication date
Application filed by Honor Device Co Ltd filed Critical Honor Device Co Ltd
Priority to EP22857104.8A priority Critical patent/EP4210226B1/en
Priority to US18/022,633 priority patent/US12549170B2/en
Publication of WO2023093216A1 publication Critical patent/WO2023093216A1/zh
Anticipated expiration legal-status Critical
Ceased legal-status Critical Current

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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—ELECTRIC POWER NETWORKS; CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J7/00—Circuit arrangements for charging or discharging batteries or for supplying loads from batteries
    • H02J7/34—Parallel operation in networks using both storage and other DC sources, e.g. providing buffering
    • H02J7/342—The other DC source being a battery actively interacting with the first one, i.e. battery to battery charging
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03K—PULSE TECHNIQUE
    • H03K17/00—Electronic switching or gating, i.e. not by contact-making and –breaking
    • H03K17/06—Modifications for ensuring a fully conducting state
    • H03K17/063—Modifications for ensuring a fully conducting state in field-effect transistor switches
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03K—PULSE TECHNIQUE
    • H03K17/00—Electronic switching or gating, i.e. not by contact-making and –breaking
    • H03K17/51—Electronic switching or gating, i.e. not by contact-making and –breaking characterised by the components used
    • H03K17/56—Electronic switching or gating, i.e. not by contact-making and –breaking characterised by the components used by the use, as active elements, of semiconductor devices
    • H03K17/687—Electronic switching or gating, i.e. not by contact-making and –breaking characterised by the components used by the use, as active elements, of semiconductor devices the devices being field-effect transistors
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02J—ELECTRIC POWER NETWORKS; CIRCUIT ARRANGEMENTS OR SYSTEMS FOR SUPPLYING OR DISTRIBUTING ELECTRIC POWER; SYSTEMS FOR STORING ELECTRIC ENERGY
    • H02J7/00—Circuit arrangements for charging or discharging batteries or for supplying loads from batteries
    • H02J7/90—Regulation of charging or discharging current or voltage
    • H02J7/96—Regulation of charging or discharging current or voltage in response to battery voltage
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03K—PULSE TECHNIQUE
    • H03K17/00—Electronic switching or gating, i.e. not by contact-making and –breaking
    • H03K17/08—Modifications for protecting switching circuit against overcurrent or overvoltage
    • H03K2017/0806—Modifications for protecting switching circuit against overcurrent or overvoltage against excessive temperature
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03K—PULSE TECHNIQUE
    • H03K2217/00—Indexing scheme related to electronic switching or gating, i.e. not by contact-making or -breaking covered by H03K17/00
    • H03K2217/0009—AC switches, i.e. delivering AC power to a load
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03K—PULSE TECHNIQUE
    • H03K2217/00—Indexing scheme related to electronic switching or gating, i.e. not by contact-making or -breaking covered by H03K17/00
    • H03K2217/0036—Means reducing energy consumption
    • H—ELECTRICITY
    • H03—ELECTRONIC CIRCUITRY
    • H03K—PULSE TECHNIQUE
    • H03K2217/00—Indexing scheme related to electronic switching or gating, i.e. not by contact-making or -breaking covered by H03K17/00
    • H03K2217/0081—Power supply means, e.g. to the switch driver
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the present application relates to the field of circuit technology, in particular to a switch circuit and electronic equipment.
  • Two-way charging means that mobile phones, tablets, laptops and other terminals can be charged through chargers, mobile power supplies, etc., that is, forward charging; mobile phones, tablets, laptops and other electronic devices can also be turned into mobile power supplies to directly charge other terminals.
  • Charging electronic devices to be charged for example, can charge electronic devices such as mobile phones, tablets, laptops, and smart wearable devices, that is, reverse charging.
  • Electronic devices with a bidirectional charging function are attracting more and more attention from users because they can effectively and timely provide emergency power to other electronic devices.
  • MOSFETs Metal Oxide Semiconductor field effect transistors
  • the present application provides a switch circuit and electronic equipment. On-resistance can be reduced.
  • an embodiment of the present application provides a switch circuit
  • the switch circuit includes: an external node, an internal node, a first switch module, and a drive module;
  • the first switch module includes an enhanced gallium nitride high electron mobility transistor, and the enhanced The gallium nitride high electron mobility transistor includes a first gate, a first pole and a second pole;
  • the driving module includes a control terminal; the first gate is coupled to the control terminal, the first pole is coupled to an external node, and the internal node is coupled to a second pole.
  • the external node is used to receive the charging voltage and transmit the charging voltage to the enhanced GaN high electron mobility transistor
  • the driving module is used to control the enhanced GaN high electron mobility transistor to turn on or off, When the enhancement mode GaN high electron mobility transistor is turned on, the charging voltage is transmitted to the internal node to be transmitted to the charging conversion chip through the internal node; or, the internal node is used to receive the charging voltage and transmit the charging voltage to the enhanced Gallium nitride high electron mobility transistor
  • the drive module is used to control the enhancement mode gallium nitride high electron mobility transistor to turn on or off, when the enhancement mode gallium nitride high electron mobility transistor is turned on, the charging voltage is transmitted to the outside node to be transmitted to the electronic device to be charged through an external node.
  • the driving module controls the enhanced gallium nitride high electron mobility transistor to turn on or off, and realizes the bidirectional turn-on and turn-off functions of the switch circuit. And because the enhancement mode GaN high electron mobility transistor itself does not have a body diode, there will be no current leakage problem when the enhancement mode GaN high electron mobility transistor is turned off. In this way, through an enhancement GaN-type high electron mobility transistors can realize bidirectional turn-on and turn-off functions. Compared with setting two back-to-back metal-oxide-semiconductor field-effect transistors, the layout size of components is reduced.
  • the setting of the enhanced gallium nitride high electron mobility transistor can reduce the loss of the switching circuit and reduce the power consumption of the switching circuit , which can alleviate the problem of device heating.
  • the external node when the external node receives the charging voltage, the external node can be coupled with the charger, for example, and the internal node can be coupled with the charging conversion chip; when the internal node receives the charging voltage, the internal node can be coupled with the charging conversion chip, for example, the external A node may, for example, be coupled to an electronic device to be charged.
  • the electronic device to be charged is, for example, a mobile phone to be charged.
  • the driving module further includes a reference terminal, and the second pole of the enhanced gallium nitride high electron mobility transistor and the reference terminal of the control module are coupled to the reference point; when the external node receives the charging voltage, the driving module Based on the voltage of the reference point, the first driving voltage is sent to the first gate, and the enhanced GaN high electron mobility transistor is turned on in response to the first driving voltage, and the charging voltage is transmitted to the internal node; the driving module is also based on the reference point The voltage at the first gate sends the second driving voltage to the first gate, and the enhancement-mode gallium nitride high electron mobility transistor is turned off in response to the second driving voltage; or, when the internal node receives the charging voltage, the driving module sends the voltage to the first gate based on the voltage of the reference point.
  • the first gate sends the third driving voltage, and the enhanced GaN high electron mobility transistor is turned on in response to the third driving voltage, and transmits the charging voltage to the external node; the driving module also sends the fourth driving voltage to the first gate voltage, the enhancement mode GaN high electron mobility transistor is turned off in response to the fourth driving voltage.
  • the driving module when charging forward, the driving module outputs the first driving voltage based on the voltage of the reference point, so that the difference between the voltage of the first gate and the voltage of the second pole is greater than The threshold voltages of the first gate and the second pole of the enhanced gallium nitride high electron mobility transistor, so as to keep the first switch module in the on state, and ensure the normal charging; when the positive direction is turned off (complete charging or When encountering an abnormal situation), the driving module outputs the second driving voltage based on the voltage of the reference point, so that the difference between the voltage of the first gate and the voltage of the second electrode is smaller than the second driving voltage of the enhanced GaN high electron mobility transistor.
  • the drive module when reverse charging, the drive module outputs a third drive voltage based on the voltage of the reference point, so that the first gate The difference between the voltage of the first electrode and the voltage of the first electrode is greater than the threshold voltage of the first gate and the first electrode of the enhancement-mode gallium nitride high electron mobility transistor, thereby keeping the first switch module in an on state and ensuring the charging Normal operation; when the reverse cut-off (complete charging or encountered abnormal conditions), in order to prevent the first switch module from opening when the voltage at the first pole drops, the drive module outputs the fourth drive voltage through the control terminal, so that the first The difference between the voltage of the gate and the voltage of the first electrode is always smaller than the threshold voltage of the first gate and the first electrode of the enhanced gallium nitride high electron mobility transistor, so that the first switch module is always in the off state, That is to realize the bidirectional on and off function.
  • the driving module when forward charging, forward shutdown and reverse charging, the driving module outputs a third drive voltage based on the voltage of the reference point,
  • the switch circuit further includes a second switch module, the second switch The module is used to prevent the internal node from transmitting the received charging voltage to the reference point when the driving module sends the fourth driving voltage to the first grid; the driving module sends the fourth driving voltage to the first grid based on the voltage of the reference point, enhancing The GaN-type high electron mobility transistor is turned off in response to the fourth driving voltage. That is, when the driving module outputs the driving voltages (the first driving voltage, the second driving voltage, the third driving voltage and the fourth driving voltage), they all change based on the change of the reference point. In this way, the calculation of the driving module can be further simplified .
  • the second switch module includes an N-type metal-oxide-semiconductor field-effect transistor;
  • the N-type metal-oxide-semiconductor field-effect transistor includes a second gate, source and drain; the source is coupled to the second pole, the drain is coupled to the internal node, and the signal obtained by the second gate can control whether the NMOS field effect transistor is turned on or not.
  • the reference point can be separated from the internal node, so that not only the positive
  • the driving module outputs a driving voltage based on the voltage of the reference point, so that the first switch module is turned on during forward charging and reverse charging and turned off when forward turning off ; and when turning off in the reverse direction, the driving module can output the driving voltage based on the voltage of the reference point, so that the first switch module is completely turned off when turning off in the reverse direction, that is, in different stages, the driving module is based on the reference point
  • the voltage output drive voltage so that the calculation process of the drive module is further simplified.
  • the switch circuit in this embodiment includes an enhanced bidirectional GaN high electron mobility transistor and an N-type metal oxide semiconductor field effect transistor
  • the enhanced bidirectional GaN high electron mobility transistor itself has an on-resistance Low characteristic, therefore, compared with two metal-oxide-semiconductor field-effect transistors, the on-resistance of the switching circuit of the present embodiment is reduced, and it is verified that it includes an enhancement-mode bidirectional gallium nitride high electron mobility transistor and an N-type metal
  • the on-resistance of the switching circuit of the oxide semiconductor field effect transistor can be reduced by 25%, so that the switching loss can be reduced, the power consumption can be reduced, and the heating can be alleviated.
  • the second switch module includes an N-type metal-oxide-semiconductor field-effect transistor;
  • the N-type metal-oxide-semiconductor field-effect transistor includes a second gate, source and drain; the drain is coupled to the second pole and the internal node respectively, the source is coupled to the reference point, and the signal obtained by the second gate can control whether the NMOS field effect transistor is turned on or not.
  • the reference point can be separated from the internal node, so that not only the positive
  • the driving module outputs a driving voltage based on the voltage of the reference point, so that the first switch module is turned on during forward charging and reverse charging and turned off when forward turning off ; and when turning off in the reverse direction, the driving module can output the driving voltage based on the voltage of the reference point, so that the first switch module is completely turned off when turning off in the reverse direction, that is, in different stages, the driving module is based on the reference point
  • the voltage output drive voltage so that the calculation process of the drive module is further simplified.
  • the switch circuit in this embodiment includes an enhanced bidirectional GaN high electron mobility transistor and an N-type metal oxide semiconductor field effect transistor
  • the enhanced bidirectional GaN high electron mobility transistor itself has an on-resistance Therefore, compared with two MOSFETs, the on-resistance of the switching circuit of this embodiment is reduced, and since the second switching module is not located on the charging path, the second switching module can For low-power devices, it has been verified that the switching circuit (not in the charging path) including the enhancement-mode bi-directional GaN high electron mobility transistor and the N-type MOSFET is compared to two MOSFETs
  • the switching circuit of the crystal, the switching circuit of the embodiment of the present application can reduce the on-resistance by more than 40%, so that the switching loss can be further reduced, the power consumption can be reduced, and the heating can be alleviated.
  • the range of the on-resistance of the NMOS field effect transistor is greater than or equal to 1 ohm, and less than or equal to 100 ohms; the withstand voltage range of the second gate to the source of the N-type metal oxide semiconductor field effect transistor is greater than or equal to 5V, and less than or equal to 60V; the N-type metal oxide semiconductor field effect transistor The drain-source breakdown voltage range of the effect transistor is greater than or equal to 10V and less than or equal to 120V. That is, the second switch module is a low-power metal-oxide-semiconductor field effect transistor, and the loss of the second switch module can be relatively low, which can further reduce the power consumption of the switch circuit.
  • the first gate is coupled to the second gate.
  • the first gate can obtain the signal alone, or it can be coupled with the second gate, and the signal output from the control terminal of the driving module can simultaneously control the on or off of the first switch module and the second switch module, thus simplifying the switching The structure of the circuit.
  • the switch circuit further includes a resistor; the first end of the resistor is coupled to the reference point, and the second end of the resistor is grounded.
  • the resistor By setting the resistor, it can be turned off in the forward direction and in the reverse direction.
  • the voltage of the reference point drops to 0V quickly, and the voltage of the driving module based on the reference point also drops to 0V quickly, that is, the charge in the enhanced bidirectional GaN high electron mobility transistor and the N-type metal-oxide-semiconductor field-effect transistor
  • the rapid discharge makes the enhanced bidirectional GaN high electron mobility transistor and the NMOS field effect transistor rapidly turned off.
  • the resistance value of the resistor is greater than 100 kohms, that is, the current of the reference point will not be too large when the reference point is discharged to the ground, and the reference point can also be made The voltage drops to 0V relatively quickly.
  • the enhancement mode GaN high electron mobility transistor can be an ordinary enhancement mode bidirectional GaN high electron mobility transistor, that is, the gate-source threshold voltage of the enhancement mode GaN high electron mobility transistor and The gate-drain threshold voltages are different; it can also be an enhanced bidirectional gallium nitride high electron mobility transistor, that is, the threshold voltage of the gate and the second pole is equal to the threshold voltage of the gate and the first pole.
  • an embodiment of the present application provides an electronic device, the electronic device includes the switching circuit described in any one of the foregoing, and can realize all the effects of the foregoing switching circuit.
  • FIG. 1 is a schematic structural diagram of an electronic device provided in an embodiment of the present application.
  • FIG. 2 is one of a schematic diagram of an application scenario of an electronic device provided in an embodiment of the present application
  • FIG. 3 is one of a schematic diagram of an application scenario of an electronic device provided in an embodiment of the present application.
  • FIG. 4 is a schematic structural diagram of a switch circuit provided in an embodiment of the present application.
  • FIG. 5 is a schematic structural diagram of another electronic device provided in the embodiment of the present application.
  • FIG. 6 is a schematic structural diagram of another switch circuit provided by the embodiment of the present application.
  • FIG. 7 is a schematic structural diagram of another switch circuit provided by the embodiment of the present application.
  • FIG. 8 is a schematic structural diagram of another electronic device provided in the embodiment of the present application.
  • FIG. 9 is a simulation diagram during forward charging provided by the embodiment of the present application.
  • FIG. 10 is a simulation diagram during forward shutdown provided by the embodiment of the present application.
  • FIG. 11 is a simulation diagram during reverse charging provided by the embodiment of the present application.
  • FIG. 12 is a simulation diagram during reverse turn-off provided by the embodiment of the present application.
  • FIG. 13 is a schematic structural diagram of another switch circuit provided by the embodiment of the present application.
  • FIG. 14 is a schematic structural diagram of another switch circuit provided by the embodiment of the present application.
  • FIG. 15 is a schematic structural diagram of another electronic device provided by the embodiment of the present application.
  • Fig. 16 is another simulation diagram during forward charging provided by the embodiment of the present application.
  • FIG. 17 is another simulation diagram during forward shutdown provided by the embodiment of the present application.
  • Fig. 18 is another simulation diagram during reverse charging provided by the embodiment of the present application.
  • FIG. 19 is another simulation diagram of reverse turn-off provided by the embodiment of the present application.
  • first and second in the description and claims of the embodiments of the present application are used to distinguish different objects, rather than to describe a specific order of objects.
  • first target object, the second target object, etc. are used to distinguish different target objects, rather than describing a specific order of the target objects.
  • words such as “exemplary” or “for example” are used as examples, illustrations or illustrations. Any embodiment or design scheme described as “exemplary” or “for example” in the embodiments of the present application shall not be interpreted as being more preferred or more advantageous than other embodiments or design schemes. Rather, the use of words such as “exemplary” or “such as” is intended to present related concepts in a concrete manner.
  • multiple processing units refers to two or more processing units; multiple systems refers to two or more systems.
  • the embodiment of the present application provides an electronic device.
  • the electronic device provided in the embodiment of the present application may be a mobile phone, a computer, a tablet computer, a personal digital assistant (PDA for short), a vehicle-mounted computer, a TV, a smart wearable device, a smart Household equipment, etc., the embodiment of the present application does not specifically limit the specific form of the above-mentioned electronic equipment.
  • the electronic device is a mobile phone as an example for description below.
  • a mobile phone 100 includes an external interface 10, a switch circuit 20, a charging conversion chip 30, a battery 40 and a system on chip (System on Chip , SoC) 50 and other structures.
  • the system on chip (SoC chip) 50 is integrated with, for example, a central processing unit (Central Processing Unit, CPU), an image processing unit (Graphic Processing Unit, GPU) and a modem (Modem).
  • the setting of the external interface 10 enables the mobile phone 100 to be coupled with an external device.
  • the external interface 10 is, for example, a USB interface or the like.
  • External devices may include, for example, electronic devices such as chargers, mobile power supplies, earphones, mobile phones to be charged, tablet computers, and notebook computers.
  • the system on chip (SoC chip) 50 can be coupled with the external interface 10, the system on chip (SoC chip) 50 can receive the voltage of the external interface 10, and identify the type of the external device connected to the external interface 10 based on the charging protocol, so as to Determine whether the external interface 10 is connected to the mobile phone to be charged, or the charger.
  • the switch circuit 20 is turned on.
  • the charging voltage output by the charger 200 is transmitted to the charging conversion chip 30 through the external interface 10 and the switching circuit 20 .
  • the charging conversion chip 30 converts the charging voltage, and transmits the converted voltage to the battery 40 , so as to charge the battery 40 .
  • the converted voltage is transmitted to the system on chip (SoC chip) 50 to provide power for the CPU, GPU and Modem integrated in the system on chip (SoC chip) 50 to ensure the normal operation of the CPU, GPU and Modem.
  • SoC chip system on chip
  • the switch circuit 20 is turned off, and the charging voltage output by the charger 200 cannot be transmitted to the charging conversion chip 30 .
  • the charging conversion chip 30 cannot supply power to the battery 40 and the CPU, GPU and Modem integrated in the system on chip (SoC chip) 50 .
  • the battery 40 provides working voltage for the CPU, GPU and Modem integrated in the system on chip (SoC chip) 50 to ensure the normal operation of the CPU, GPU and Modem integrated in the system on chip (SoC chip) 50 .
  • the switch circuit 20 when the external device connected to the external interface 10 is the mobile phone 300 to be charged, that is, when the mobile phone 100 needs to charge the mobile phone 300 to be charged, the switch circuit 20 is turned on.
  • the charging voltage output by the battery 40 is transmitted to the mobile phone 300 to be charged through the charging conversion chip 30 , the switch circuit 20 and the external interface 10 , so as to charge the mobile phone 300 to be charged.
  • an embodiment of the present application provides a switch circuit, where the switch circuit includes a first switch module and a drive module.
  • the first switch module is an enhanced gallium nitride high electron mobility transistor.
  • the enhanced gallium nitride high electron mobility transistor includes a first gate, a first pole and a second pole.
  • the driving module is coupled to the first gate, and is used for outputting a driving voltage to the first gate.
  • the turn-on or turn-off of the enhancement-mode GaN high electron mobility transistor is related to its threshold voltage.
  • the enhanced gallium nitride high electron mobility transistor in the case of forward charging (the voltage at the first pole needs to be transmitted to the second pole), when the voltage of the first gate and the voltage of the second pole When the difference between the voltage of the first gate and the voltage of the second electrode is less than the threshold voltage, the enhanced GaN high electron mobility transistor is turned on; rate transistor off.
  • the enhanced gallium nitride high The electron mobility transistor is turned on; when the difference between the voltage of the first gate and the voltage of the first electrode is less than the threshold voltage, the enhanced gallium nitride high electron mobility transistor is turned off.
  • the enhancement mode GaN high electron mobility transistor can be controlled to be turned on or off through the driving voltage output by the driving module, that is, in order to realize bidirectional
  • the driving voltage is a voltage output from the driving module to the first gate, and this voltage can control the turn-on or turn-off of the enhanced gallium nitride high electron mobility transistor.
  • the enhancement mode GaN high electron mobility transistor itself does not have a body diode, there will be no current leakage problem when the enhancement mode GaN high electron mobility transistor is turned off. In this way, the bidirectional turn-on and turn-off functions can be realized through the enhanced GaN high electron mobility transistor.
  • the layout size of components is reduced.
  • the enhanced gallium nitride high electron mobility transistor itself has the characteristics of low on-resistance, the setting of the enhanced gallium nitride high electron mobility transistor can reduce the loss of the switching circuit and reduce the power consumption of the switching circuit , and relieve fever.
  • the embodiment of the present application does not specifically limit the type of the enhanced gallium nitride high electron mobility transistor.
  • it may be an ordinary enhanced gallium nitride high electron mobility transistor, or an enhanced bidirectional gallium nitride high electron mobility transistor.
  • the first switch module is an ordinary enhanced gallium nitride high electron mobility transistor
  • the first pole is, for example, the drain of the enhanced gallium nitride high electron mobility transistor
  • the second pole is, for example, the enhanced gallium nitride high electron mobility transistor. Source of the electron mobility transistor.
  • the enhanced gallium nitride high electron mobility transistor is turned on; when the difference between the first gate voltage and the source voltage of the enhanced gallium nitride high electron mobility transistor is less than the gate-source threshold voltage, the enhanced gallium nitride high The electron mobility transistor is turned off.
  • the enhanced gallium nitride high electron mobility transistor is turned on; when the difference between the first gate voltage and the drain voltage of the enhanced gallium nitride high electron mobility transistor is less than the gate-drain threshold voltage, the enhanced gallium nitride high The electron mobility transistor turns on.
  • the gate-source threshold voltage and the gate-drain threshold voltage are different.
  • the so-called enhanced bidirectional GaN high electron mobility transistor means that the first pole and the second pole of the enhancement mode GaN high electron mobility transistor are equivalent. That is, the distance from the area where the second pole is located to the channel region is equal to the distance from the area where the first pole is located to the channel area, and the material of the area where the second pole is located is the same as that of the area where the first pole is located and the device structure same.
  • the difference between the first gate voltage and the second pole voltage of the enhanced bidirectional GaN high electron mobility transistor is greater than the gate
  • the enhancement mode bidirectional GaN high electron mobility transistor is turned on when the threshold voltage of the first pole and the second pole is lower than the difference between the first gate voltage and the second pole voltage When the threshold voltage of the gate and the second electrode is lower, the enhancement mode bidirectional GaN high electron mobility transistor is turned off.
  • the difference between the first gate voltage and the first pole voltage of the enhanced bidirectional GaN high electron mobility transistor is greater than the gate voltage When the threshold voltage of the first electrode and the first electrode, the enhanced bidirectional GaN high electron mobility transistor is turned on; the difference between the first gate voltage and the first electrode voltage of the enhanced bidirectional GaN high electron mobility transistor is less than When the threshold voltage of the gate and the first electrode is lower, the enhancement mode bidirectional GaN high electron mobility transistor is turned off.
  • the threshold voltage of the gate and the second electrode is equal to the threshold voltage of the gate and the first electrode.
  • the embodiment of the present application does not limit the specific structure of the driving module.
  • the driver module includes a driver chip and a Microcontroller Unit (MCU), wherein the MCU and the driver chip can be set separately, or the driver chip can be integrated with a MUC.
  • MCU Microcontroller Unit
  • the first switch module as an enhanced bidirectional gallium nitride high electron mobility transistor as an example for illustration. It can be understood that since the threshold voltages of the first gate and the second electrode of the enhanced bidirectional gallium nitride high electron mobility transistor are equal to the threshold voltages of the first gate and the first electrode, no matter whether the following embodiments are The threshold voltage of the first gate and the second electrode, or the threshold voltage of the first gate and the first electrode are both referred to as the threshold voltage of the enhanced bidirectional GaN high electron mobility transistor.
  • the following example uses the first gate of the enhanced bidirectional GaN high electron mobility transistor when charging in the forward direction
  • the received driving voltage is the first driving voltage, wherein the first driving voltage can control the enhanced bidirectional GaN high electron mobility transistor to turn on when it is forwardly charged
  • the driving voltage received by the first gate of the mobility transistor is the second driving voltage, wherein the second driving voltage can control the enhanced bidirectional gallium nitride high electron mobility transistor to turn off in the forward direction
  • the driving voltage received by the first gate of the gallium high electron mobility transistor is the third driving voltage, wherein the third driving voltage can control the enhanced bidirectional gallium nitride high electron mobility transistor to turn on when reverse charging; reverse turn off When off, the driving voltage received by the first gate of the enhanced bidirectional GaN high electron mobility transistor is the fourth driving voltage, wherein the fourth driving voltage can control the reverse turn-off of the enhanced bidirectional GaN high electron mobility transistor .
  • the following content does not
  • the switch circuit 20 includes an internal node V BUS , an external node V USB , a first switch module 21 and a drive module 22 .
  • the first switch module 21 includes an enhanced bidirectional GaN high electron mobility transistor.
  • the enhanced bidirectional GaN high electron mobility transistor includes a first gate G1, a first pole D1 and a second pole D2.
  • the driving module 22 includes a reference terminal R1 and a control terminal C.
  • the first pole D1 is coupled to the external node V USB
  • the second pole D2 is coupled to the reference point R2
  • the reference terminal R1 is coupled to the reference point R2
  • the internal node V BUS is coupled to the second pole D2 .
  • the control terminal C of the driving module 22 is coupled to the first grid G1, and the driving module 22 outputs a driving voltage to the first grid G1 through the control terminal C.
  • the switch circuit 20 is coupled to the external interface 10 through the external node V USB , and the switch circuit 20 is coupled to the charging conversion chip 30 through the internal node V BUS .
  • the driving module 22 collects the voltage at the reference point R2 (the second pole D2 ) in real time.
  • forward charging the charging voltage received by the external node V USB needs to be transmitted to the internal node V BUS
  • the driving module 22 Outputting the first driving voltage based on the voltage at the reference point R2 (second pole D2), wherein the difference between the first driving voltage and the collected voltage at the reference point R2 (second pole D2) is kept at the first difference in real time
  • the first difference is greater than the threshold voltage of the enhanced bidirectional GaN high electron mobility transistor, that is, the voltage difference between the voltage of the first gate G1 and the voltage of the second electrode D2 is greater than the enhanced bidirectional GaN high electron mobility
  • the threshold voltage of the transistor in this way, during forward charging, it can ensure that the enhanced bidirectional gallium nitride high electron mobility transistor is always in the on state, thereby ensuring the normal progress of forward charging.
  • the driving module 22 When charging is completed, etc., when it is necessary to change the enhanced bidirectional GaN high electron mobility transistor from the on state to the off state, the driving module 22 outputs the second driving voltage based on the voltage at the reference point R2 (D2), wherein, the difference between the second driving voltage and the collected voltage at the reference point R2 (D2) is kept at the second difference in real time, and the second difference is smaller than the threshold voltage of the enhanced bidirectional gallium nitride high electron mobility transistor, that is The voltage difference between the voltage of the first gate G1 and the voltage of the second electrode D2 is smaller than the threshold voltage of the enhanced bidirectional GaN high electron mobility transistor, so that the enhanced bidirectional GaN high electron mobility transistor is turned off broken.
  • the driving module 22 In the case of reverse charging (the charging voltage received by the internal node V BUS needs to be transmitted to the external node V USB ), that is, when the enhanced bidirectional GaN high electron mobility transistor needs to be turned on and kept on, the driving module 22
  • the third drive voltage is output based on the voltage at the reference point R2 (D2), wherein the difference between the third drive voltage and the collected voltage at the reference point R2 (D2) is a third difference, and the third difference is greater than the enhanced Threshold voltage of bidirectional GaN high electron mobility transistors.
  • the threshold voltage of the electron mobility transistor in this way, during reverse charging, it can ensure that the enhanced bidirectional gallium nitride high electron mobility transistor is always in the on state, thereby ensuring the normal progress of reverse charging.
  • the driving module 22 When charging is completed, etc., when it is necessary to change the enhanced bidirectional gallium nitride high electron mobility transistor from the on state to the off state, the driving module 22 outputs a fourth driving voltage to the first gate G1, and the fourth driving voltage For example, 0.
  • the fourth driving voltage is 0V
  • the voltage at the first pole D1 will be greater than or equal to 0V, that is, the first pole D1
  • the difference between the four driving voltages (0V) and the first pole D1 (greater than or equal to 0V) must be less than or equal to 0, that is, the difference between the voltage of the first grid G1 and the voltage of the first pole D1 must be less than or equal to Equal to 0, which is less than the threshold voltage of the enhanced bidirectional GaN high electron mobility transistor. Therefore, a complete turn-off of the enhancement-mode bi-directional GaN high electron mobility transistor is achieved.
  • the external device connected to the external interface 10 is the charger 200, that is, when the charger 200 needs to charge the mobile phone 100, that is, the charging voltage received by the external node V USB
  • the enhanced bidirectional GaN high electron mobility transistor needs to be turned from an off state to an on state.
  • the driving module 22 will collect the voltage of the reference point R2 (same as the voltage at the second pole D2) in real time and perform calculations based on the reference point R2, so that the output of the first A voltage difference between the driving voltage and the collected reference point R2 is kept at the first difference of 5V in real time.
  • the driving module 22 sends a first driving voltage of 5V to the first gate G1.
  • the difference between the voltage (5V) of the first gate G1 and the voltage (0V) of the second pole D2 is 5V, which is greater than the threshold voltage of 1.5V of the enhanced bidirectional gallium nitride high electron mobility transistor, and the enhanced bidirectional gallium nitride GaN high electron mobility transistor turns on.
  • the voltage of the second pole D2 gradually rises from 0V to 20V.
  • the driving module 22 will collect the voltage of the reference point R2 (second pole D2) in real time, and adjust the first driving voltage output to the first grid G1 in real time based on the collected voltage of the reference point R2 (second pole D2), wherein , the difference between the first driving voltage and the voltage at the reference point R2 (the second pole D2 ) is always kept at the first difference of 5V. Therefore, when the voltage of the reference point R2 (second pole D2) gradually rises to 20V, the driving module 22 will also adjust its output first driving voltage in real time based on the voltage of the reference point R2 (second pole D2) to gradually increase from 5V to 25V.
  • the voltage difference between the first gate G1 and the second pole D2 is kept at 5V, which is greater than the threshold voltage 1.5V of the enhanced bidirectional GaN high electron mobility transistor.
  • the enhanced bidirectional GaN GaN high electron mobility transistors are always on.
  • the charging voltage 20V of the external node V USB is transmitted to the internal node V BUS through the conduction-enhanced bidirectional GaN high electron mobility transistor, so as to be transmitted to the charging conversion chip 30 through the internal node V BUS .
  • the charging conversion chip 30 converts the charging voltage, and transmits the converted voltage to the battery 40 , so as to charge the battery 40 .
  • the converted voltage is transmitted to the system on chip (SoC chip) 50 to provide power for the CPU, GPU and Modem integrated in the system on chip (SoC chip) 50 to ensure the normal operation of the CPU, GPU and Modem.
  • SoC chip system on chip
  • the enhanced bidirectional GaN high electron mobility transistor When the charging is completed; or when an abnormal situation such as excessive voltage will cause damage to the mobile phone 100 is encountered during the charging process, it is necessary to change the enhanced bidirectional GaN high electron mobility transistor from the on state to the off state.
  • the voltages of the external node V USB , the internal node V BUS , the first pole D1, the second pole D2 and the reference point R2 are all charging voltages of 20V.
  • a grid G1 has a voltage of 25V.
  • the driving module 22 will collect the voltage of the reference point R2 (same as the voltage at the second pole D2) in real time and perform calculations based on the reference point R2, so that the output of the first A voltage difference between the driving voltage and the collected reference point R2 is kept at the second difference 0V in real time. Therefore, when the driving module 22 collects a voltage of 20V at the reference point R2, the driving module 22 sends a second driving voltage of 20V to the first grid G1.
  • the difference between the voltage of the first gate G1 and the voltage of the second electrode D2 is 0V, which is less than the threshold voltage of 1.5V of the enhanced bidirectional GaN high electron mobility transistor, and the enhanced bidirectional GaN high electron mobility rate transistor off.
  • the driving module 22 will collect the voltage of the reference point R2 (second pole D2) in real time, and adjust the second driving voltage output to the first grid G1 in real time based on the collected voltage of the reference point R2 (second pole D2), Wherein, the difference between the second driving voltage and the voltage of the reference point R2 (the second pole D2 ) is always kept at the second difference of 0V.
  • the voltage of the reference point R2 (second pole D2) will drop to 0V, and the second driving voltage output by the driving module 22 will also drop accordingly, so that the first The voltage difference between the first gate G1 and the second pole D2 of the switch module 21 is always kept at 0V, which ensures that the enhanced bidirectional GaN high electron mobility transistor is always in an off state.
  • the charging voltage output by the charger 200 cannot be transmitted to the charging conversion chip 30 .
  • the charging conversion chip 30 cannot supply power to the battery 40 and the CPU, GPU and Modem integrated in the system on chip (SoC chip) 50 .
  • the driving module 22 collects the voltage of the reference point R2 (second pole D2) in real time, and performs calculation based on the collected reference point R2 (second pole D2), The voltage difference between the output third driving voltage and the collected reference point R2 is kept at the third difference of 5V in real time.
  • the driving module 22 sends a third driving voltage of 25V to the first grid G1 through the control terminal C.
  • the difference between the voltage (25V) of the first gate G1 and the voltage (0V) of the first pole D1 is 25V, which is greater than the threshold voltage of 1.5V of the enhanced bidirectional gallium nitride high electron mobility transistor, and the enhanced bidirectional gallium nitride
  • the GaN high electron mobility transistor is turned on, and the voltage of the first pole D1 gradually increases from 0V to 20V.
  • the driving module 22 Since the voltage of the reference point R2 (the second pole D2) has been kept at 20V, the driving module 22 has been sending a third driving voltage of 25V to the first grid G1, and the third driving voltage (25V) is the same as that of the first pole D1 (maximum 20V), the voltage difference is greater than or equal to 5V, which is greater than the threshold voltage 1.5V of the enhanced bidirectional GaN high electron mobility transistor.
  • the enhanced bidirectional GaN high electron mobility transistor is always at The conduction state ensures the normal charging.
  • the enhanced bidirectional GaN high electron mobility transistor When the charging is completed; or when an abnormal situation such as excessive voltage is encountered during the charging process will cause damage to the mobile phone 300 to be charged, it is necessary to change the enhanced bidirectional GaN high electron mobility transistor from the on state to the off state.
  • the enhanced bidirectional GaN high electron mobility transistor When the enhanced bidirectional GaN high electron mobility transistor is turned on, the voltages of the internal node V BUS , the external node VUSB, the first pole D1 , the second pole D2 and the reference point R2 are all charging voltages of 20V.
  • the voltage of the first grid G1 is, for example, 25V. In order to turn off the enhancement mode bi-directional GaN high electron mobility transistor.
  • the driving module 22 cannot perform calculations based on the reference point R2, so that the difference between the output fourth driving voltage and the voltage at the reference point R2 remains at the fourth difference 0V, that is, the fourth driving voltage is equal to the voltage at the reference point R2. This is because if the driving module 22 outputs the fourth driving voltage based on the voltage of the reference point R2, since the voltage of the reference point R2 and the internal node V BUS are both 20V, the fourth driving voltage output by the driving module 22 is 20V.
  • the driving module 22 directly outputs the fourth driving voltage of 0V.
  • the driving module 22 When the driving module 22 outputs the fourth driving voltage of 0V to the first gate G1, even if the voltage at the first pole D1 drops, the voltage at the first pole D1 will be greater than or equal to 0V, that is, the first gate
  • the difference between the voltage of G1 (0V) and the voltage of the first pole D1 (greater than or equal to 0V) must be less than or equal to 0, which is less than the threshold voltage 1.5V of the enhanced bidirectional GaN high electron mobility transistor. Therefore, a complete turn-off of the enhancement-mode bi-directional GaN high electron mobility transistor is achieved.
  • the switch circuit 20 further includes a second switch module 23 .
  • the second switch module 23 is coupled between the second pole D2 and the internal node V BUS .
  • the second switch module 23 includes a MOSFET, for example, may include an NMOSFET.
  • the NMOSFET includes a second gate G2, a source S and a drain D3.
  • the source S of the second switch module 23 is coupled to the second pole D2 of the first switch module 21, the drain D3 of the second switch module 23 is coupled to the internal node V BUS , and the second gate G2 of the second switch module 23 is coupled to The first gate G1 of the first switch module 21 is coupled.
  • the second switch module 23 is set to separate the reference point R2 from the internal node V BUS .
  • the driving module 22 outputs voltage based on the reference point R2 driving voltage, so that the first switch module 21 is turned on during forward charging and reverse charging and is turned off when it is turned off in the forward direction; and when it is turned off in the reverse direction, the driving module 22 can output driving voltage, so that the first switch module 21 is completely turned off when it is turned off in reverse, that is, at different stages, the driving module 22 outputs the driving voltage based on the voltage of the reference point R2, thus further simplifying the operation process of the driving module .
  • the switch circuit 20 in this embodiment includes an enhanced bidirectional GaN high electron mobility transistor and an NMOSFET
  • the enhanced bidirectional GaN high electron mobility transistor itself has the characteristic of low on-resistance, therefore, it is relatively Compared with the two MOSFETs, the on-resistance of the switching circuit 20 of this embodiment is reduced. It has been verified that the switching circuit 20 including the enhanced bidirectional gallium nitride high electron mobility transistor and the NMOSFET is compared with the switching circuit of the two MOSFETs, The on-resistance can be reduced by 25%. In this way, switching loss can be reduced, power consumption can be reduced, and heat generation can be relieved.
  • the switch circuit 20 further includes a resistor 24 , one end of the resistor 24 is coupled to the reference point R2 , and the other end of the resistor 24 is grounded.
  • the voltage of the reference point R2 can be quickly dropped to 0V during forward shutdown and reverse shutdown, and the voltage of the driving module 22 based on the reference point R2 is also quickly dropped to 0V, that is, the enhanced The charge in the bidirectional GaN high electron mobility transistor and the NMOSFET is quickly discharged, thereby making the enhancement mode bidirectional GaN high electron mobility transistor and the NMOSFET quickly turned off.
  • the resistance value of the resistor 24 is, for example, greater than 100 kilohms.
  • the resistance value of the resistor 24 is set to be greater than 100 k ⁇ , the current of the reference point R2 will not be too large when discharging to the ground, and the voltage of the reference point R2 will drop to 0V relatively quickly.
  • the resistor 24 is set in the switch circuit 20 and one end of the resistor 24 is grounded as an example, but this does not constitute a limitation to the present application.
  • the reference terminal R1 of the driving module 22 may also be set to ground. The grounding of the reference terminal R1 makes the voltage of the reference point R2 quickly drop to 0V, and the voltage of the driving module 22 based on the reference point R2 also quickly drops to 0V.
  • the second switch module 23 is an NMOSFET, and the threshold voltage of the NMOSFET is 2V as an example for description.
  • FIG. 9 is a simulation diagram of forward charging provided in this embodiment.
  • the charger 200 when the external device connected to the external interface 10 is the charger 200, that is, the charger 200 needs to charge the mobile phone 100
  • the charging voltage received by the external node V USB needs to be transmitted to the internal node V BUS , it is necessary to turn the enhanced bidirectional GaN high electron mobility transistor and the NMOSFET from the off state to the on state.
  • the enhanced bidirectional GaN high electron mobility transistor and NMOSFET are both turned off, and the voltage at the external node V USB is the charging voltage 20V.
  • the internal node V BUS is 0V
  • the first gate G1 and the second gate G2 are also 0V
  • the voltages of the reference point R2, the second pole D2 and the source S are the same as the internal node V BUS .
  • the driving module 22 In the turn-on phase, in order to turn on the enhanced bidirectional GaN high electron mobility transistor and the NMOSFET, the driving module 22 will collect the voltage of the reference point R2 (same as the voltage at the second pole D2) in real time and perform the operation based on the reference point R2. The operation is performed so that the voltage difference between the output first driving voltage and the collected reference point R2 is maintained at the first difference of 5V in real time. Therefore, when the voltage at the reference point R2 collected by the driving module 22 is 0V, the driving module 22 gradually applies the first driving voltage to 5V to the first grid G1 and the second grid G2 in 100 ⁇ s. It can be seen from Fig.
  • the first driving voltage applied by the first gate G1 and the second gate G2 starts to be greater than 2V, that is, the voltage of the first gate G1 (greater than 2V) and the second pole D2 (0V)
  • the voltage difference is greater than the threshold voltage 1.5V of the enhanced bidirectional GaN high electron mobility transistor, the enhanced bidirectional GaN high electron mobility transistor is turned on, the second gate G1 (greater than 2V) and the source S (0V ) is greater than the threshold voltage 2V of the NMOSFET, the NMOSFET is turned on, and the internal node V BUS begins to have a voltage.
  • the voltage at the internal node V BUS rises from 0V to 20V.
  • the voltage at the reference point R2 will also rise from 0V to 20V. Since the drive module 22 collects the voltage at the reference point R2 in real time (the same as the voltage at the second pole D2) and performs calculations based on the reference point R2, the voltage difference between the first output driving voltage and the collected reference point R2 is real-time Keep at the first difference 5V. Therefore, when the voltage at the reference point R2 also rises from 0V to 20V, the first driving voltage output from the control terminal C of the driving module 22 will gradually rise to 25V.
  • the voltages at the internal node V BUS and the reference point R2 are both maintained at 20V, and the first driving voltage output by the control terminal C of the driving module 22 is also maintained at 25V.
  • the first gate The voltage difference between G1 (25V) and the second pole D2 (20V) is kept at 5V, which is greater than the threshold voltage 1.5V of the enhanced bidirectional gallium nitride high electron mobility transistor.
  • the second gate G2 (25V) and the source S (20V) The voltage difference is kept at 5V, which is greater than the threshold voltage of the NMOSFET 2V.
  • the enhanced bidirectional GaN high electron mobility transistor and the NMOSFET are always in the on state.
  • the charging voltage 20V of the external node V USB is transmitted to the internal node V BUS through the conduction-enhanced bidirectional GaN high electron mobility transistor, so as to be transmitted to the charging conversion chip 30 through the internal node V BUS .
  • the charging conversion chip 30 converts the charging voltage, and transmits the converted voltage to the battery 40 , so as to charge the battery 40 .
  • the converted voltage is transmitted to the system on chip (SoC chip) 50 to provide power for the CPU, GPU and Modem integrated in the system on chip (SoC chip) 50 to ensure the normal operation of the CPU, GPU and Modem.
  • SoC chip system on chip
  • Fig. 10 is the simulation diagram of the forward shutdown provided by this embodiment, combined with Fig. 2, Fig. 8 and Fig. 10, when the charging is completed; or, the mobile phone 100 will be damaged if it encounters an excessive voltage during the charging process, etc. In abnormal situations, it is necessary to change the enhanced bidirectional GaN high electron mobility transistor and NMOSFET from the on state to the off state.
  • both the enhanced bidirectional GaN high electron mobility transistor and the NMOSFET are turned on, the external node V USB , the internal node V BUS , the second pole D2, the source S, and the drain
  • the voltages of D3 and the reference point R2 are both charging voltages of 20V.
  • the voltages of the first grid G1 and the second grid G2 are 25V.
  • the drive module 22 In the turn-off phase, in order to turn off the enhanced bi-directional GaN high electron mobility transistor and the NMOSFET, the drive module 22 will collect the voltage of the reference point R2 in real time (same as the voltage at the second pole D2) and perform the operation based on the reference point R2. operation, so that the voltage difference between the output second driving voltage and the collected reference point R2 is kept at the second difference 0V in real time. Therefore, at 305 ⁇ s, the second driving voltage output by the driving module 22 gradually drops to 20V.
  • the second driving voltage applied by the first gate G1 and the second gate G2 drops to 21.5V, that is, the first gate G1 (21.5V) and the second gate D2 (20V)
  • the voltage difference is equal to the threshold voltage 1.5V of the enhanced bidirectional gallium nitride high electron mobility transistor, and the voltage difference between the second gate G1 (21.5V) and the source S (20V) is less than the threshold voltage 2V of the NMOSFET.
  • the bidirectional GaN high electron mobility transistor and the NMOSFET start to turn off, and the voltage at the internal node V BUS starts to drop.
  • the voltage at the internal node V BUS gradually drops from 20V to 0V.
  • the voltage at the reference point R2 gradually drops from 20V to 0V. Since the driving module 22 collects the voltage of the reference point R2 in real time and performs calculations based on the reference point R2, the difference between the output second driving voltage and the collected voltage of the reference point R2 is kept at the second difference 0V in real time. Therefore, when the voltage at the reference point R2 gradually drops from 20V to 0V, the second driving voltage output from the control terminal C of the driving module 22 will gradually drop to 0V accordingly.
  • the voltages at the internal node V BUS and the reference point R2 are both kept at 0V, and the second driving voltage output by the control terminal C of the driving module 22 is also kept at 0V.
  • the first gate The voltage difference between the pole G1 (0V) and the second pole D2 (0V) is kept at 0V, which is less than the threshold voltage 1.5V of the enhanced bidirectional gallium nitride high electron mobility transistor, the second gate G2 (0V) and the source
  • the voltage difference of S(0V) is kept at 0V, which is less than the threshold voltage of the NMOSFET 2V, so that the enhanced bidirectional GaN high electron mobility transistor and the NMOSFET are always in the off state.
  • the charging voltage output by the charger 200 cannot be transmitted to the charging conversion chip 30 .
  • the charging conversion chip 30 cannot supply power to the battery 40 and the CPU, GPU and Modem integrated in the system on chip (SoC chip) 50 .
  • Fig. 11 is a simulation diagram during reverse charging provided by this embodiment.
  • the external device connected to the external interface 10 when the external device connected to the external interface 10 is the mobile phone 300 to be charged, that is, the mobile phone 100 needs to be a mobile phone to be charged
  • the 300 is charging that is, when the charging voltage received by the internal node V BUS needs to be transmitted to the external node V USB , it is necessary to turn the enhanced bidirectional GaN high electron mobility transistor and the NMOSFET from the off state to the on state.
  • the enhanced bidirectional GaN high electron mobility transistor and NMOSFET are both turned off, and the voltage at the internal node V BUS is the charging voltage 20V.
  • the external node V USB is 0V
  • the first gate G1 and the second gate G2 are also 0V
  • the voltages of the reference point R2, the second pole D2 and the source S are the same as the external node V USB .
  • the driving module 22 In the turn-on phase, in order to turn on the enhanced bidirectional GaN high electron mobility transistor and the NMOSFET, the driving module 22 will collect the voltage of the reference point R2 (same as the voltage at the second pole D2) in real time and perform the operation based on the reference point R2. operation, so that the voltage difference between the output third driving voltage and the collected reference point R2 is kept at the third difference 5V in real time. Therefore, when the voltage at the reference point R2 collected by the driving module 22 is 0V, the driving module 22 gradually applies the first driving voltage to 5V to the first grid G1 and the second grid G2 in 101 ⁇ s.
  • the first driving voltage applied by the first gate G1 and the second gate G2 starts to be greater than 2V, that is, the first gate G1 (greater than 2V) and the first electrode D1 (0V)
  • the voltage difference is greater than the threshold voltage 1.5V of the enhanced bidirectional GaN high electron mobility transistor, the enhanced bidirectional GaN high electron mobility transistor is turned on, the second gate G1 (greater than 2V) and the source S (0V ) is greater than the threshold voltage 2V of the NMOSFET, the NMOSFET is turned on, and the external node V USB begins to have a voltage.
  • the voltage at the external node V USB rises from 0V to 20V.
  • the voltage at the reference point R2 will also rise from 0V to 20V. Since the driving module 22 collects the voltage of the reference point R2 in real time and performs calculations based on the reference point R2, the difference between the output third driving voltage and the collected voltage of the reference point R2 is maintained at a third difference of 5V in real time. Therefore, when the voltage at the reference point R2 also rises from 0V to 20V, the third driving voltage output from the control terminal C of the driving module 22 will gradually rise to 25V accordingly.
  • the voltages at the external node V USB and the reference point R2 are both maintained at 20V, and the third driving voltage output by the control terminal C of the driving module 22 is also maintained at 25V.
  • the first gate The voltage difference between G1 (25V) and the first pole D1 (20V) is kept at 5V, which is greater than the threshold voltage 1.5V of the enhanced bidirectional gallium nitride high electron mobility transistor.
  • the second gate G2 (25V) and the source S (20V) The voltage difference is kept at 5V, which is greater than the threshold voltage of the NMOSFET 2V.
  • the enhanced bidirectional GaN high electron mobility transistor and the NMOSFET are always in the on state to ensure the normal charging.
  • Fig. 12 is a simulation diagram of the reverse shutdown provided by this embodiment, combined with Fig. 3, Fig. 8 and Fig. 12, when the charging is completed; or, encountering an excessive voltage during the charging process will cause damage to the mobile phone 100, etc. In abnormal situations, it is necessary to change the enhanced bidirectional GaN high electron mobility transistor and NMOSFET from the on state to the off state.
  • both the enhanced bidirectional GaN high electron mobility transistor and the NMOSFET are turned on, the external node V USB , the internal node V BUS , the second pole D2, the source S, and the drain
  • the voltages of D3 and the reference point R2 are both charging voltages of 20V.
  • the voltages of the first grid G1 and the second grid G2 are 25V.
  • the drive module 22 In the turn-off phase, in order to turn off the enhanced bi-directional GaN high electron mobility transistor and the NMOSFET, the drive module 22 will collect the voltage of the reference point R2 in real time (same as the voltage at the second pole D2) and perform the operation based on the reference point R2. operation, so that the voltage difference between the output fourth driving voltage and the collected reference point R2 is kept at the fourth difference 0V in real time. Therefore, at 310 ⁇ s, the fourth driving voltage output by the driving module 22 gradually drops to 20V.
  • the fourth driving voltage applied by the first gate G1 and the second gate G2 drops to 21.5V, that is, the first gate G1 (21.5V) and the first pole D1 (20V)
  • the voltage difference is equal to the threshold voltage 1.5V of the enhanced bidirectional gallium nitride high electron mobility transistor, and the voltage difference between the second gate G1 (21.5V) and the source S (20V) is less than the threshold voltage 2V of the NMOSFET.
  • the bidirectional GaN high electron mobility transistor and the NMOSFET start to turn off, and the voltage at the internal node V BUS starts to drop from 20V.
  • the voltage at the internal node V BUS gradually drops from 20V to 0V.
  • the voltage at the reference point R2 gradually drops from 20V to 0V. Since the driving module 22 collects the voltage of the reference point R2 in real time and performs calculations based on the reference point R2, the difference between the output fourth driving voltage and the collected voltage of the reference point R2 remains at the fourth difference 0V in real time. Therefore, when the voltage at the reference point R2 gradually drops from 20V to 0V, the fourth driving voltage output from the control terminal C of the driving module 22 will gradually drop to 0V accordingly.
  • the voltages at the internal node V BUS and the reference point R2 are both kept at 0V, and the fourth driving voltage output by the control terminal C of the driving module 22 is also kept at 0V.
  • the first gate The voltage difference between G1 (0V) and the first pole D1 (0V) is kept at 0V, which is less than the threshold voltage 1.5V of the enhanced bidirectional gallium nitride high electron mobility transistor, the second gate G2 (0V) and the source S
  • the voltage difference (0V) is kept at 0V, that is, less than the threshold voltage of the NMOSFET 2V, so that the enhanced bidirectional GaN high electron mobility transistor and the NMOSFET are always in an off state.
  • a complete turn-off of the enhancement-mode bidirectional GaN high electron mobility transistor is achieved.
  • the difference from the previous example is the setting position of the second switch module 23 .
  • the drain D3 of the second switch module 23 is respectively coupled to the second pole D2 of the first switch module 21 and the internal node V BUS
  • the source S of the second switch module 23 is coupled to the reference point R2
  • the second gate G2 of 23 is coupled to the first gate G1 of the first switch module 21 .
  • the second switch module 23 can be, for example, a low-power MOSFET, such as a low-power Si NMOSFET.
  • the on-resistance range of the second switch module 22 is greater than or equal to 1 ohm and less than or equal to 100 ohms; the withstand voltage range of the second switch module 22 from the gate to the source is greater than or equal to 5V, and Less than or equal to 60V; the range of drain-source breakdown voltage is greater than or equal to 10V and less than or equal to 120V.
  • the switch circuit 20 of this embodiment can further reduce the on-resistance because the second switch module 23 can be a low-power MOSFET. It has been verified that the switch circuit 20 of this embodiment can reduce the on-resistance by more than 40%. In this way, switching loss can be further reduced, power consumption can be reduced, and heat generation can be relieved.
  • the switch circuit 20 may include a resistor 24 . Wherein, one end of the resistor 24 is coupled to the reference point R2, and the other end of the resistor 24 is grounded. In other optional embodiments, referring to FIG. 14 , the reference end R1 of the driving module 22 may also be set to be grounded.
  • the second switch module 23 is an NMOSFET, and the threshold voltage of the NMOSFET is 2V as an example for description.
  • FIG. 16 is a simulation diagram of forward charging provided in this embodiment.
  • the charger 200 when the external device connected to the external interface 10 is the charger 200, that is, the charger 200 needs to charge the mobile phone 100
  • the charging voltage received by the external node V USB needs to be transmitted to the internal node V BUS , it is necessary to turn the enhanced bidirectional GaN high electron mobility transistor and the NMOSFET from the off state to the on state.
  • the enhanced bidirectional GaN high electron mobility transistor and NMOSFET are both turned off, and the voltage at the external node V USB is the charging voltage 20V.
  • the internal node V BUS is 0V
  • the first gate G1 and the second gate G2 are also 0V
  • the voltages of the reference point R2, the second pole D2 and the source S are the same as the internal node V BUS .
  • the driving module 22 In the turn-on phase, in order to turn on the enhanced bidirectional GaN high electron mobility transistor and the NMOSFET, the driving module 22 will collect the voltage of the reference point R2 (same as the voltage at the second pole D2) in real time and perform the operation based on the reference point R2. The operation is performed so that the voltage difference between the output first driving voltage and the collected reference point R2 is maintained at the first difference of 5V in real time. Therefore, when the voltage collected by the driving module 22 at the reference point R2 is 0V, the driving module 22 gradually applies the first driving voltage to 5V to the first grid G1 and the second grid G2 in 100 ⁇ s. It can be seen from Fig.
  • the first driving voltage applied by the first gate G1 and the second gate G2 starts to be greater than 2V, that is, the voltage of the first gate G1 (greater than 2V) and the second pole D2 (0V)
  • the voltage difference is greater than the threshold voltage 1.5V of the enhanced bidirectional GaN high electron mobility transistor, the enhanced bidirectional GaN high electron mobility transistor is turned on, the second gate G1 (greater than 2V) and the source S (0V ) is greater than the threshold voltage 2V of the NMOSFET, the NMOSFET is turned on, and the internal node V BUS begins to have a voltage.
  • the voltage at the internal node V BUS rises from 0V to 20V.
  • the voltage at the reference point R2 will also rise from 0V to 20V. Since the drive module 22 collects the voltage at the reference point R2 in real time (the same as the voltage at the second pole D2) and performs calculations based on the reference point R2, the voltage difference between the first output driving voltage and the collected reference point R2 is real-time Keep at the first difference 5V. Therefore, when the voltage at the reference point R2 also rises from 0V to 20V, the first driving voltage output from the control terminal C of the driving module 22 will gradually rise to 25V accordingly.
  • the voltages at the internal node V BUS and the reference point R2 are both maintained at 20V, and the first driving voltage output by the control terminal C of the driving module 22 is also maintained at 25V.
  • the first gate The voltage difference between G1 (25V) and the second pole D2 (20V) is kept at 5V, which is greater than the threshold voltage 1.5V of the enhanced bidirectional gallium nitride high electron mobility transistor.
  • the second gate G2 (25V) and the source S (20V) The voltage difference is kept at 5V, which is greater than the threshold voltage of the NMOSFET 2V.
  • the enhanced bidirectional GaN high electron mobility transistor and the NMOSFET are always in the on state.
  • the charging voltage 20V of the external node V USB is transmitted to the internal node V BUS through the conduction-enhanced bidirectional GaN high electron mobility transistor, so as to be transmitted to the charging conversion chip 30 through the internal node V BUS .
  • the charging conversion chip 30 converts the charging voltage, and transmits the converted voltage to the battery 40 , so as to charge the battery 40 .
  • the converted voltage is transmitted to the system on chip (SoC chip) 50 to provide power for the CPU, GPU and Modem integrated in the system on chip (SoC chip) 50 to ensure the normal operation of the CPU, GPU and Modem.
  • SoC chip system on chip
  • Fig. 17 is the simulation diagram of the forward shutdown provided by this embodiment, combined with Fig. 2, Fig. 15 and Fig. 17, when the charging is completed; or, the mobile phone 100 will be damaged if it encounters an excessive voltage during the charging process, etc. In abnormal situations, it is necessary to change the enhanced bidirectional GaN high electron mobility transistor and NMOSFET from the on state to the off state.
  • both the enhanced bidirectional GaN high electron mobility transistor and the NMOSFET are turned on, the external node V USB , the internal node V BUS , the second pole D2, the source S, and the drain
  • the voltages of D3 and the reference point R2 are both charging voltages of 20V.
  • the voltages of the first grid G1 and the second grid G2 are 25V.
  • the drive module 22 In the turn-off phase, in order to turn off the enhanced bi-directional GaN high electron mobility transistor and the NMOSFET, the drive module 22 will collect the voltage of the reference point R2 in real time (same as the voltage at the second pole D2) and perform the operation based on the reference point R2. operation, so that the voltage difference between the output second driving voltage and the collected reference point R2 is kept at the second difference 0V in real time. Therefore, at 302 ⁇ s, the second driving voltage output by the driving module 22 gradually drops to 20V. It can be seen from Figure 17 that at 303 ⁇ s, the second driving voltage applied by the first gate G1 and the second gate G2 drops to 21.5V, that is, the voltage of the first gate G1 (21.5V) and the second gate D2 (20V).
  • the voltage difference is equal to the threshold voltage 1.5V of the enhanced bidirectional gallium nitride high electron mobility transistor, and the voltage difference between the second gate G1 (21.5V) and the source S (20V) is less than the threshold voltage 2V of the NMOSFET.
  • the bidirectional GaN high electron mobility transistor and the NMOSFET start to turn off, and the voltage at the internal node V BUS starts to drop.
  • the voltage at the internal node V BUS gradually drops from 20V to 0V.
  • the voltage at the reference point R2 gradually drops from 20V to 0V. Since the driving module 22 collects the voltage of the reference point R2 in real time and performs calculations based on the reference point R2, the difference between the output second driving voltage and the collected voltage of the reference point R2 is kept at the second difference 0V in real time. Therefore, when the voltage at the reference point R2 gradually drops from 20V to 0V, the second driving voltage output from the control terminal C of the driving module 22 will gradually drop to 0V accordingly.
  • the voltages at the internal node V BUS and the reference point R2 are both kept at 0V, and the second driving voltage output by the control terminal C of the driving module 22 is also kept at 0V, so that the first gate
  • the voltage difference between G1 (0V) and the second pole D2 (0V) is kept at 0V, which is less than the threshold voltage 1.5V of the enhanced bidirectional gallium nitride high electron mobility transistor, the second gate G2 (0V) and the source S
  • the voltage difference (0V) is kept at 0V, that is, less than the threshold voltage of the NMOSFET 2V, so that the enhanced bidirectional GaN high electron mobility transistor and the NMOSFET are always in an off state.
  • the charging voltage output by the charger 200 cannot be transmitted to the charging conversion chip 30 .
  • the charging conversion chip 30 cannot supply power to the battery 40 and the CPU, GPU and Modem integrated in the system on chip (SoC chip) 50 .
  • Fig. 18 is a simulation diagram during reverse charging provided by this embodiment.
  • the external device connected to the external interface 10 when the external device connected to the external interface 10 is the mobile phone 300 to be charged, that is, the mobile phone 100 needs to be a mobile phone to be charged
  • the 300 is charging that is, when the charging voltage received by the internal node V BUS needs to be transmitted to the external node V USB , it is necessary to turn the enhanced bidirectional GaN high electron mobility transistor and the NMOSFET from the off state to the on state.
  • both the enhanced bidirectional GaN high electron mobility transistor and the NMOSFET are turned off, and the voltage at the internal node V BUS is the charging voltage 20V.
  • the external node V USB is 0V
  • the first gate G1 and the second gate G2 are also 0V
  • the voltages of the reference point R2, the second pole D2 and the source S are the same as the external node V USB .
  • the driving module 22 In the turn-on phase, in order to turn on the enhanced bidirectional GaN high electron mobility transistor and the NMOSFET, the driving module 22 will collect the voltage of the reference point R2 (same as the voltage at the second pole D2) in real time and perform the operation based on the reference point R2. operation, so that the voltage difference between the output third driving voltage and the collected reference point R2 is kept at the third difference 5V in real time. Therefore, when the voltage at the reference point R2 collected by the driving module 22 is 0V, the driving module 22 gradually applies the first driving voltage to 5V to the first grid G1 and the second grid G2 in 100.4 ⁇ s.
  • the first driving voltage applied by the first gate G1 and the second gate G2 starts to be greater than 2V, that is, the first gate G1 (greater than 2V) and the first electrode D1 (0V)
  • the voltage difference is greater than the threshold voltage 1.5V of the enhanced bidirectional GaN high electron mobility transistor, the enhanced bidirectional GaN high electron mobility transistor is turned on, the second gate G1 (greater than 2V) and the source S (0V ) is greater than the threshold voltage 2V of the NMOSFET, the NMOSFET is turned on, and the external node V USB begins to have a voltage.
  • the voltage at the external node V USB rises from 0V to 20V.
  • the voltage at the reference point R2 will also rise from 0V to 20V. Since the driving module 22 collects the voltage of the reference point R2 in real time and performs calculations based on the reference point R2, the difference between the output third driving voltage and the collected voltage of the reference point R2 is maintained at a third difference of 5V in real time. Therefore, when the voltage at the reference point R2 also rises from 0V to 20V, the third driving voltage output from the control terminal C of the driving module 22 will gradually rise to 25V accordingly.
  • the voltages at the external node V USB and the reference point R2 are both maintained at 20V, and the third driving voltage output by the control terminal C of the driving module 22 is also maintained at 25V.
  • the first gate The voltage difference between G1 (25V) and the first pole D1 (20V) is kept at 5V, which is greater than the threshold voltage 1.5V of the enhanced bidirectional gallium nitride high electron mobility transistor.
  • the second gate G2 (25V) and the source S (20V) The voltage difference is kept at 5V, which is greater than the threshold voltage of the NMOSFET 2V.
  • the enhanced bidirectional GaN high electron mobility transistor and the NMOSFET are always in the on state to ensure the normal charging.
  • Fig. 19 is a simulation diagram of the reverse shutdown provided by this embodiment. Combining Fig. 3, Fig. 15 and Fig. 19, when the charging is completed; or, encountering an excessive voltage during the charging process will cause damage to the mobile phone 100, etc. In abnormal situations, it is necessary to change the enhanced bidirectional GaN high electron mobility transistor and NMOSFET from the on state to the off state.
  • both the enhanced bidirectional GaN high electron mobility transistor and the NMOSFET are turned on, the external node V USB , the internal node V BUS , the second pole D2, the source S, and the drain
  • the voltages of D3 and the reference point R2 are both charging voltages of 20V.
  • the voltages of the first grid G1 and the second grid G2 are 25V.
  • the drive module 22 In the turn-off phase, in order to turn off the enhanced bi-directional GaN high electron mobility transistor and the NMOSFET, the drive module 22 will collect the voltage of the reference point R2 in real time (same as the voltage at the second pole D2) and perform the operation based on the reference point R2. operation, so that the voltage difference between the output fourth driving voltage and the collected reference point R2 is kept at the fourth difference 0V in real time. Therefore, at 301 ⁇ s, the fourth driving voltage output by the driving module 22 gradually drops to 20V. It can be seen from Figure 12 that at 302 ⁇ s, the fourth driving voltage applied by the first gate G1 and the second gate G2 drops to 21.5V, that is, the voltage of the first gate G1 (21.5V) and the first pole D1 (20V).
  • the voltage difference is equal to the threshold voltage 1.5V of the enhanced bidirectional gallium nitride high electron mobility transistor, and the voltage difference between the second gate G1 (21.5V) and the source S (20V) is less than the threshold voltage 2V of the NMOSFET.
  • the bidirectional GaN high electron mobility transistor and the NMOSFET start to turn off, and the voltage at the internal node V BUS starts to drop from 20V.
  • the voltage at the internal node V BUS gradually drops from 20V to 0V.
  • the voltage at the reference point R2 gradually drops from 20V to 0V. Since the driving module 22 collects the voltage of the reference point R2 in real time and performs calculations based on the reference point R2, the difference between the output fourth driving voltage and the collected voltage of the reference point R2 remains at the fourth difference 0V in real time. Therefore, when the voltage at the reference point R2 gradually drops from 20V to 0V, the fourth driving voltage output from the control terminal C of the driving module 22 will gradually drop to 0V accordingly.
  • the voltages at the internal node V BUS and the reference point R2 are both kept at 0V, and the fourth driving voltage output by the control terminal C of the driving module 22 is also kept at 0V, so that the first gate
  • the voltage difference between the pole G1 (0V) and the first pole D1 (0V) is kept at 0V, which is less than the threshold voltage 1.5V of the enhanced bidirectional gallium nitride high electron mobility transistor, the second gate G2 (0V) and the source
  • the voltage difference of S(0V) is kept at 0V, which is less than the threshold voltage of the NMOSFET 2V, so that the enhanced bidirectional GaN high electron mobility transistor and the NMOSFET are always in the off state.
  • a complete turn-off of the enhancement-mode bidirectional GaN high electron mobility transistor is achieved.

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Abstract

本申请实施例提供了一种开关电路及电子设备，涉及电路技术领域，可以减小导通阻抗。该开关电路包括外部节点、内部节点、增强型氮化镓高电子迁移率晶体管和驱动模块；增强型氮化镓高电子迁移率晶体管包括第一栅极、第一极和第二极；驱动模块包括控制端；第一栅极与控制端耦合，第一极和外部节点耦合，内部节点和第二极耦合；外部节点接收充电电压，驱动模块控制增强型氮化镓高电子迁移率晶体管导通或关断，当增强型氮化镓高电子迁移率晶体管导通时将充电电压传输至内部节点；或者，内部节点接收充电电压，驱动模块控制增强型氮化镓高电子迁移率晶体管导通或关断，当增强型氮化镓高电子迁移率晶体管导通时充电电压传输至外部节点。

Description

开关电路及电子设备
本申请要求于2021年11月26日提交中国国家知识产权局、申请号为202111425594.7、申请名称为“开关电路及电子设备”的中国专利申请的优先权，其全部内容通过引用结合在本申请中。
技术领域
本申请涉及电路技术领域，尤其涉及一种开关电路及电子设备。
背景技术
双向充电为既可以通过充电器、移动电源等给手机、平板、笔记本电脑等终端进行充电，即进行正向充电；也可以把手机、平板、笔记本电脑等电子设备变为移动电源，直接给其他待充电电子设备充电，例如，可以给手机、平板、笔记本电脑、智能穿戴式设备等电子设备充电，即进行反向充电。双向充电功能的电子设备由于能够有效、及时的为其他电子设备提供应急电能，因此，受到越来越多用户的关注。
为了实现双向充电的双向导通与关断功能，一般在电子设备中设置两个背靠背的金属氧化物半导体场效应晶体管(Metal Oxide Semiconductor field effect transistor，MOSFET)，避免使用一个MOSFET时，由于MOSFET本身具有体二极管，即便MOSFET关断，还是会有电流通过MOSFET的体二极管泄露，即一个MOSFET无法实现关断功能。然而，MOSFET导通阻抗大，开关频率低，导致器件发热严重，输入输出功率受限。
发明内容
为了解决上述技术问题，本申请提供一种开关电路及电子设备。可以减小导通阻抗。
第一方面，本申请实施例提供一种开关电路，该开关电路包括：外部节点、内部节点、第一开关模块和驱动模块；第一开关模块包括增强型氮化镓高电子迁移率晶体管，增强型氮化镓高电子迁移率晶体管包括第一栅极、第一极和第二极；驱动模块包括控制端；第一栅极与控制端耦合，第一极和外部节点耦合，内部节点和第二极耦合；外部节点用于接收充电电压，并将充电电压传输至增强型氮化镓高电子迁移率晶体管；驱动模块用于控制增强型氮化镓高电子迁移率晶体管导通或关断，当增强型氮化镓高电子迁移率晶体管导通时，充电电压传输至内部节点，以通过内部节点传输至充电转换芯片；或者，内部节点用于接收充电电压，并将充电电压传输至增强型氮化镓高电子迁移率晶体管；驱动模块用于控制增强型氮化镓高电子迁移率晶体管导通或关断，当增强型氮化镓高电子迁移率晶体管导通时，充电电压传输至外部节点，以通过外部节点传输至待充电电子设备。
通过驱动模块控制增强型氮化镓高电子迁移率晶体管导通或关断，实现了开关电路的双向导通与关断功能。且由于增强型氮化镓高电子迁移率晶体管自身特性不具有体二极管，所以当增强型氮化镓高电子迁移率晶体管关断时也不会存在电流泄露的问题，这样一来，通过一个增强型氮化镓高电子迁移率晶体管即可实现双向的导通以及关断功能，相比于设置两个背靠背的金属氧化物半导体场效应晶体管，减小了元器件的布局尺寸。此外，由于增强型氮化镓高电子迁移率晶体管本身具有导通阻抗低的特性，因此，增强型氮化镓高电子迁移率晶体管的设置，可以减少开关电路的损耗，降低开关电路的功耗，进而可以缓解器件发热的问题。
示例性的，当外部节点接收充电电压时，外部节点例如可以与充电器耦合，内部节点例如可以与充电转换芯片耦合；当内部节点接收充电电压时，内部节点例如可以与充电转换芯片耦合，外部节点例如可以与待充电电子设备耦合。
示例性的，待充电电子设备例如为待充电手机等。
在一些可能实现的方式中，驱动模块还包括参考端，增强型氮化镓高电子迁移率晶体管的第二极和控制模块的参考端耦合于参考点；当外部节点接收充电电压时，驱动模块基于参考点的电压向第一栅极发送第一驱动电压，增强型氮化镓高电子迁移率晶体管响应于第一驱动电压导通，并将充电电压传输至内部节点；驱动模块还基于参考点的电压向第一栅极发送第二驱动电压，增强型氮化镓高电子迁移率晶体管响应于第二驱动电压关断；或者，当内部节点接收充电电压时，驱动模块基于参考点的电压向第一栅极发送第三驱动电压，增强型氮化镓高电子迁移率晶体管响应于第三驱动电压导通，并将充电电压传输至外部节点；驱动模块还向第一栅极发送第四驱动电压，增强型氮化镓高电子迁移率晶体管响应于第四驱动电压关断。
通过设置增强型氮化镓高电子迁移率晶体管，当正向充电时，驱动模块基于参考点的电压输出第一驱动电压，以使第一栅极的电压与第二极的电压的差值大于增强型氮化镓高电子迁移率晶体管的第一栅极与第二极的阈值电压，进而使第一开关模块保持导通的状态，保证充电的正常进行；当正向关断(完成充电或遇到异常情况)时，驱动模块基于参考点的电压输出第二驱动电压，以使第一栅极的电压与第二极的电压的差值小于增强型氮化镓高电子迁移率晶体管的第一栅极与第二极的阈值电压，进而使第一开关模块由导通变为关断；当反向充电时，驱动模块基于参考点的电压输出第三驱动电压，以使第一栅极的电压与第一极的电压的差值大于增强型氮化镓高电子迁移率晶体管的第一栅极与第一极的阈值电压，进而使第一开关模块保持导通的状态，保证充电的正常进行；当反向截止(完成充电或遇到异常情况)时，为了防止第一开关模块在第一极处的电压下降时打开，驱动模块通过控制端输出第四驱动电压，以使第一栅极的电压与第一极的电压的差值一直小于增强型氮化镓高电子迁移率晶体管的第一栅极与第一极的阈值电压，进而使第一开关模块一直处于关断状态，即实现双向导通与关断功能。此外，在正向充电、正向关断以及反向充电时，驱动模块基于参考点的电压输出驱动电压，简化驱动模块的运算过程。
在一些可能实现的方式中，在上述增强型氮化镓高电子迁移率晶体管的第二极和控制模块的参考端耦合于参考点的基础上，开关电路还包括第二开关模块，第二开关模块用于当驱动模块向第一栅极发送第四驱动电压时阻止内部节点将接收到的充电电压传输至参考点；驱动模块基于参考点的电压向第一栅极发送第四驱动电压，增强型氮化镓高电子迁移率晶体管响应于第四驱动电压关断。即驱动模块输出驱动电压(第一驱动电压、第二驱动电压、第三驱动电压和第四驱动电压)时都是基于参考点的变化而变化的，这样一来，可以进一步简化驱动模块的运算。
在一些可能实现的方式中，在开关电路包括第二开关模块的基础上，第二开关模块包括N型金属氧化物半导体场效应晶体管；N型金属氧化物半导体场效应晶体管包括第二栅极、源极和漏极；源极与第二极耦合，漏极与内部节点耦合，第二栅极获取的信号可以控制N型金属氧化物半导体场效应晶体管的导通与否。当N型金属氧化物半导体场效应晶体设置于内部节点和增强型氮化镓高电子迁移率晶体管的第二极之间时，可以使参考点与内部节点隔开，这样一来，不仅在正向充电、正向关断以及反向充电时，驱动模块基于参考点的电压输出驱动电压，以使第一开关模块在正向充电和反向充电时导通以及在正向关断时关断；且在反向关断时，驱动模块可以基于参考点的电压输出驱动电压，以使第一开关模块在反向关断时彻底关断，即在不同的阶段，驱动模块都是基于参考点的电压输出驱动电压，如此，进一步简化驱动模块的运算过程。此外，虽然本实施例中开关电路包括增强型双向氮化镓高电子迁移率晶体管以及N型金属氧化物半导体场效应晶体管，但是由于增强型双向氮化镓高电子迁移率晶体管本身具有导通阻抗低的特性，因此，相比于两个金属氧化物半导体场效应晶体管，本实施例的开关电路的导通阻抗降低，经验证，包括增强型双向氮化镓高电子迁移率晶体管以及N型金属氧化物半导体场效应晶体管的开关电路相比于两个金属氧化物半导体场效应晶体管的开关电路，导通阻抗可下降25％，这样一来，可以减少开关损耗，降低功耗，缓解发热。
在一些可能实现的方式中，在开关电路包括第二开关模块的基础上，第二开关模块包括N型金属氧化物半导体场效应晶体；N型金属氧化物半导体场效应晶体管包括第二栅极、源极和漏极；漏极分别与第二极以及内部节点耦合，源极与参考点耦合，第二栅极获取的信号可以控制N型金属氧化物半导体场效应晶体管的导通与否。当N型金属氧化物半导体场效应晶体管设置于参考点和增强型氮化镓高电子迁移率晶体管的第二极之间时，可以使参考点与内部节点隔开，这样一来，不仅在正向充电、正向关断以及反向充电时，驱动模块基于参考点的电压输出驱动电压，以使第一开关模块在正向充电和反向充电时导通以及在正向关断时关断；且在反向关断时，驱动模块可以基于参考点的电压输出驱动电压，以使第一开关模块在反向关断时彻底关断，即在不同的阶段，驱动模块都是基于参考点的电压输出驱动电压，如此，进一步简化驱动模块的运算过程。此外，虽然本实施例中开关电路包括增强型双向氮化镓高电子迁移率晶体管以及N型金属氧化物半导体场效应晶体管，但是由于增强型双向氮化镓高电子迁移率晶体管本身具有导通阻抗低的特性，因此，相比于两个金属氧化物半导体场效应晶体管，本实施例的开 关电路的导通阻抗降低，且由于第二开关模块并未位于充电通路上，因此第二开关模块可以为小功率器件，经验证，包括增强型双向氮化镓高电子迁移率晶体管以及N型金属氧化物半导体场效应晶体管的开关电路(未位于充电通路)相比于两个金属氧化物半导体场效应晶体的开关电路，本申请实施例的开关电路可使导通阻抗下降40％以上，这样一来，可以进一步减少开关损耗，降低功耗，缓解发热。
在一些可能实现的方式中，在漏极分别与第二极以及内部节点耦合，源极与参考点耦合的基础上，N型金属氧化物半导体场效应晶体管的导通阻抗的范围是大于或等于1欧姆，且小于或等于100欧姆；N型金属氧化物半导体场效应晶体管的第二栅极到源极的耐压范围是大于或等于5V，且小于或等于60V；N型金属氧化物半导体场效应晶体管的漏源击穿电压的范围大于或等于10V，且小于或等于120V。即第二开关模块为小功率金属氧化物半导体场效应晶体管，第二开关模块损耗可以较低，可以进一步降低开关电路的功耗。
在一些可能实现的方式中，在第二开关模块包括N型金属氧化物半导体场效应晶体管的基础上，第一栅极与第二栅极耦合。第一栅极可以单独获取信号，也可以与第二栅极耦合，通过驱动模块控制端输出的信号同时控制第一开关模块和第二开关模块的导通或关断，这样一来，简化开关电路的结构。
在一些可能实现的方式中，开关电路还包括电阻；电阻的第一端与参考点耦合，电阻的第二端接地设置，通过设置电阻，在正向关断和反向关断时，可以使参考点的电压快速的下降至0V，驱动模块基于参考点的电压也快速的下降至0V，即使得增强型双向氮化镓高电子迁移率晶体管和N型金属氧化物半导体场效应晶体管里的电荷快速的放掉，进而使得增强型双向氮化镓高电子迁移率晶体管和N型金属氧化物半导体场效应晶体管快速的关断。
在一些可能实现的方式中，在开关电路包括电阻的基础上，电阻的电阻值大于100千欧姆，即不会使得参考点在对地进行放电时，电流过大，同时还可以使得参考点的电压较为快速的下降至0V。
在一些可能实现的方式中，增强型氮化镓高电子迁移率晶体管可以是普通的增强型双向氮化镓高电子迁移晶体管，即增强型氮化镓高电子迁移率晶体管的栅源阈值电压和栅漏阈值电压不同；也可以增强型双向氮化镓高电子迁移晶体管，即栅极和第二极的阈值电压和栅极和第一极的阈值电压相等。
第二方面，本申请实施例提供一种电子设备，该电子设备包括上述任一项所述的开关电路，能够实现上述开关电路的所有效果。
附图说明
图1为本申请实施例提供的一种电子设备的结构示意图；
图2为本申请实施例提供的电子设备的一种应用场景示意图之一；
图3为本申请实施例提供的电子设备的一种应用场景示意图之一；
图4为本申请实施例提供的一种开关电路的结构示意图；
图5为本申请实施例提供的又一种电子设备的结构示意图；
图6为本申请实施例提供的又一种开关电路的结构示意图；
图7为本申请实施例提供的又一种开关电路的结构示意图；
图8为本申请实施例提供的又一种电子设备的结构示意图；
图9为本申请实施例提供的正向充电时的一种仿真图；
图10为本申请实施例提供的正向关断时的一种仿真图；
图11为本申请实施例提供的反向充电时的一种仿真图；
图12为本申请实施例提供的反向关断时的一种仿真图；
图13为本申请实施例提供的又一种开关电路的结构示意图；
图14为本申请实施例提供的又一种开关电路的结构示意图；
图15为本申请实施例提供的又一种电子设备的结构示意图；
图16为本申请实施例提供的正向充电时的又一种仿真图；
图17为本申请实施例提供的正向关断时的又一种仿真图；
图18为本申请实施例提供的反向充电时的又一种仿真图；
图19为本申请实施例提供的反向关断时的又一种仿真图。
具体实施方式
下面将结合本申请实施例中的附图，对本申请实施例中的技术方案进行清楚、完整地描述，显然，所描述的实施例是本申请一部分实施例，而不是全部的实施例。基于本申请中的实施例，本领域普通技术人员在没有作出创造性劳动前提下所获得的所有其他实施例，都属于本申请保护的范围。
本文中术语“和/或”，仅仅是一种描述关联对象的关联关系，表示可以存在三种关系，例如，A和/或B，可以表示：单独存在A，同时存在A和B，单独存在B这三种情况。
本申请实施例的说明书和权利要求书中的术语“第一”和“第二”等是用于区别不同的对象，而不是用于描述对象的特定顺序。例如，第一目标对象和第二目标对象等是用于区别不同的目标对象，而不是用于描述目标对象的特定顺序。
在本申请实施例中，“示例性的”或者“例如”等词用于表示作例子、例证或说明。本申请实施例中被描述为“示例性的”或者“例如”的任何实施例或设计方案不应被解释为比其它实施例或设计方案更优选或更具优势。确切而言，使用“示例性的”或者“例如”等词旨在以具体方式呈现相关概念。
在本申请实施例的描述中，除非另有说明，“多个”的含义是指两个或两个以上。例如，多个处理单元是指两个或两个以上的处理单元；多个系统是指两个或两个以上的系 统。
本申请实施例提供一种电子设备，本申请实施例提供的电子设备可以是手机、电脑、平板电脑、个人数字助理(personal digital assistant，简称PDA)、车载电脑、电视、智能穿戴式设备、智能家居设备等，本申请实施例对上述电子设备的具体形式不作特殊限定。以下为了方便说明，以电子设备是手机为例进行说明。
以下对本申请实施例提供的电子设备的具体结构和用途进行说明。
如图1所示，图1示出了本申请实施例提供的一种电子设备的结构示意图，手机100包括对外接口10、开关电路20、充电转换芯片30、电池40和片上系统(System on Chip，SoC)50等结构。其中，片上系统(SoC芯片)50内部例如集成有中央处理器(Central Processing Unit，CPU)、图像处理器(Graphic Processing Unit，GPU)和调制解调器(Modem)等。对外接口10的设置使得手机100可以与外接设备耦合。其中，对外接口10例如为USB接口等。外接设备例如可以包括充电器、移动电源、耳机、待充电手机、平板电脑、笔记本电脑等电子设备。片上系统(SoC芯片)50例如可以和对外接口10耦合，该片上系统(SoC芯片)50可以接收对外接口10的电压，并基于充电协议对接入对外接口10的外接设备的类型进行识别，以确定对外接口10接入的是待充电手机，还是充电器等。
示例性的，结合图1和图2，当对外接口10接入的外接设备为充电器200，即充电器200需要为手机100进行充电时，开关电路20导通。充电器200输出的充电电压通过对外接口10、开关电路20传输至充电转换芯片30。充电转换芯片30对充电电压进行转换，并将转换后的电压传输至电池40，以实现对电池40的充电。此外，还将转换后的电压传输至片上系统(SoC芯片)50，以为片上系统(SoC芯片)50内部集成的CPU、GPU和Modem等供电，保证CPU、GPU和Modem的正常工作。充电完成后，开关电路20关断，充电器200输出的充电电压无法传输至充电转换芯片30。充电转换芯片30不能为电池40供电以及为片上系统(SoC芯片)50内部集成的CPU、GPU和Modem等供电。此时，电池40为片上系统(SoC芯片)50内部集成的CPU、GPU和Modem等提供工作时的电压，保证片上系统(SoC芯片)50内部集成的CPU、GPU和Modem等的正常工作。
示例性的，结合图1和图3，当对外接口10接入的外接设备为待充电手机300时，即手机100需要为待充电手机300进行充电时，开关电路20导通。电池40输出的充电电压通过充电转换芯片30、开关电路20以及对外接口10传输至待充电手机300，以为待充电手机300进行充电。
为了解决背景技术中的问题，本申请实施例提供了一种开关电路，该开关电路包括第一开关模块和驱动模块。该第一开关模块为增强型氮化镓高电子迁移率晶体管。增强型氮化镓高电子迁移率晶体管包括第一栅极、第一极和第二极。驱动模块与第一栅极耦合，用于向第一栅极输出驱动电压。
可以理解的是，增强型氮化镓高电子迁移率晶体管的导通或关断与其阈值电压有关。
对于增强型氮化镓高电子迁移率晶体管而言，在正向充电(第一极处的电压需要传 输至第二极处)的情况下，当第一栅极的电压和第二极的电压的差值大于阈值电压时，增强型氮化镓高电子迁移率晶体管导通；当第一栅极的电压和第二极的电压的差值小于阈值电压时，增强型氮化镓高电子迁移率晶体管关断。在反向充电(第二极处的电压需要传输至第一极处)的情况下，当第一栅极的电压和第一极的电压的差值大于阈值电压时，增强型氮化镓高电子迁移率晶体管导通；当第一栅极的电压和第一极的电压的差值小于阈值电压时，增强型氮化镓高电子迁移率晶体管关断。
由于第一栅极的电压即为驱动模块输出的驱动电压，因此，可以通过驱动模块输出的驱动电压控制增强型氮化镓高电子迁移率晶体管导通或关断，也就是说，为了实现双向充电的双向导通与关断功能，驱动电压是驱动模块向第一栅极输出的一个电压，该电压可以控制增强型氮化镓高电子迁移率晶体管的打开或关断。且由于增强型氮化镓高电子迁移率晶体管自身特性不具有体二极管，所以当增强型氮化镓高电子迁移率晶体管关断时也不会存在电流泄露的问题。这样一来，通过增强型氮化镓高电子迁移率晶体管即可实现双向的导通以及关断功能。相比于设置两个背靠背的MOS管，减小了元器件的布局尺寸。此外，由于增强型氮化镓高电子迁移率晶体管本身具有导通阻抗低的特性，因此，增强型氮化镓高电子迁移率晶体管的设置，可以减少开关电路的损耗，降低开关电路的功耗，以及缓解发热。
对于增强型氮化镓高电子迁移率晶体管的类型，本申请实施例不作具体限定。例如可以是普通的增强型氮化镓高电子迁移率晶体管，还可以是增强型双向氮化镓高电子迁移率晶体管。
当第一开关模块为普通的增强型氮化镓高电子迁移率晶体管时，第一极例如为增强型氮化镓高电子迁移率晶体管的漏极，第二极例如为增强型氮化镓高电子迁移率晶体管的源极。在正向充电(漏极处的电压需要传输至源极处)的情况下，增强型氮化镓高电子迁移率晶体管的第一栅极电压与源极电压的差值大于栅源阈值电压时，增强型氮化镓高电子迁移率晶体管导通；增强型氮化镓高电子迁移率晶体管的第一栅极电压与源极电压的差值小于栅源阈值电压时，增强型氮化镓高电子迁移率晶体管关断。在反向充电(源极处的电压需要传输至漏极处)的情况下，增强型氮化镓高电子迁移率晶体管的第一栅极电压与漏极电压的差值大于栅漏阈值电压时，增强型氮化镓高电子迁移率晶体管导通；增强型氮化镓高电子迁移率晶体管的第一栅极电压与漏极电压的差值小于栅漏阈值电压时，增强型氮化镓高电子迁移率晶体管导通。其中，栅源阈值电压和栅漏阈值电压不同。
所谓增强型双向氮化镓高电子迁移率晶体管，即为增强型氮化镓高电子迁移率晶体管的第一极和第二极等效。亦即第二极所在的区域到沟道区的距离等于第一极所在的区域到沟道区的距离，且第二极所在的区域的材料与第一极所在的区域的材料相同以及器件结构相同。在正向充电(第一极处的电压需要传输至第二极处)的情况下，增强型双向氮化镓高电子迁移率晶体管的第一栅极电压与第二极电压的差值大于栅极和第二极的阈值电压时，增强型双向氮化镓高电子迁移率晶体管导通；增强型双向氮化镓高电子迁移率晶体管的第一栅极电压与第二极电压的差值小于栅极和第二极的阈值电压时，增强型双向氮化镓高电子迁移率晶体管关断。在反向充电(第二极处的电压需要传输至第一 极处)的情况下，增强型双向氮化镓高电子迁移率晶体管的第一栅极电压与第一极电压的差值大于栅极和第一极的阈值电压时，增强型双向氮化镓高电子迁移率晶体管导通；增强型双向氮化镓高电子迁移率晶体管的第一栅极电压与第一极电压的差值小于栅极和第一极的阈值电压时，增强型双向氮化镓高电子迁移率晶体管关断。其中，栅极和第二极的阈值电压和栅极和第一极的阈值电压相等。
需要说明的是，对于驱动模块的结构，本申请实施例不对驱动模块的具体结构进行限定。例如驱动模块包括驱动芯片和微控制单元(Microcontroller Unit，MCU)，其中，MCU和驱动芯片可以单独设置，也可以是驱动芯片上集成有MUC。
通过设置增强型氮化镓高电子迁移率晶体管以降低功耗的开关电路有多种，以下以典型示例进行详细说明。为了更加方便、清楚的对后续方案进行描述，下述示例以第一开关模块为增强型双向氮化镓高电子迁移率晶体管为例进行说明。可以理解的是，由于增强型双向氮化镓高电子迁移率晶体管的第一栅极和第二极的阈值电压和第一栅极和第一极的阈值电压相等，因此下述实施例不论是第一栅极和第二极的阈值电压，还是第一栅极和第一极的阈值电压，均称为增强型双向氮化镓高电子迁移率晶体管的阈值电压。此外，为了区分不同阶段时增强型双向氮化镓高电子迁移率晶体管第一栅极接收的驱动电压，下述示例以正向充电时增强型双向氮化镓高电子迁移率晶体管第一栅极接收的驱动电压为第一驱动电压，其中，第一驱动电压可以控制增强型双向氮化镓高电子迁移率晶体管在正向充电时导通；正向关断时增强型双向氮化镓高电子迁移率晶体管第一栅极接收的驱动电压为第二驱动电压，其中，第二驱动电压可以控制增强型双向氮化镓高电子迁移率晶体管正向关断；反向充电时增强型双向氮化镓高电子迁移率晶体管第一栅极接收的驱动电压为第三驱动电压，其中，第三驱动电压可以控制增强型双向氮化镓高电子迁移率晶体管在反向充电时导通；反向关断时增强型双向氮化镓高电子迁移率晶体管第一栅极接收的驱动电压为第四驱动电压，其中，第四驱动电压可以控制增强型双向氮化镓高电子迁移率晶体管反向关断。下述内容不构成对本申请的限定。
一个示例中，如图4所示，开关电路20包括内部节点V BUS、外部节点V USB、第一开关模块21和驱动模块22。第一开关模块21包括增强型双向氮化镓高电子迁移率晶体管。该增强型双向氮化镓高电子迁移率晶体管包括第一栅极G1、第一极D1和第二极D2。驱动模块22包括参考端R1和控制端C。第一极D1和外部节点V USB耦合，第二极D2和参考点R2耦合，参考端R1和参考点R2耦合，内部节点V BUS和第二极D2耦合。驱动模块22的控制端C与第一栅极G1耦合，驱动模块22通过控制端C向第一栅极G1输出驱动电压。结合图5，开关电路20通过外部节点V USB与对外接口10耦合，开关电路20通过内部节点V BUS与充电转换芯片30耦合。
驱动模块22会实时采集参考点R2(第二极D2)处的电压。在正向充电(外部节点V USB接收的充电电压需要传输至内部节点V BUS)的情况下，即需要增强型双向氮化镓高电子迁移率晶体管导通以及保持导通状态时，驱动模块22基于参考点R2(第二极D2)处的电压输出第一驱动电压，其中，第一驱动电压与采集的参考点R2(第二极D2)处的电压的差值实时保持在第一差值，第一差值大于增强型双向氮化镓高电子迁移率晶体管的阈值电压，即第一栅极G1的电压与第二极D2的电压的压差大于增强型双 向氮化镓高电子迁移率晶体管的阈值电压，这样一来，正向充电时，可以保证增强型双向氮化镓高电子迁移率晶体管一直处于导通状态，进而保证正向充电的正常进行。
当充电完成等情况下，需要使增强型双向氮化镓高电子迁移率晶体管由导通状态变为关断状态时，驱动模块22基于参考点R2(D2)处的电压输出第二驱动电压，其中，第二驱动电压与采集的参考点R2(D2)处的电压的差值实时保持在第二差值，第二差值小于增强型双向氮化镓高电子迁移率晶体管的阈值电压，即第一栅极G1的电压与第二极D2的电压的压差小于增强型双向氮化镓高电子迁移率晶体管的阈值电压，这样一来，使得增强型双向氮化镓高电子迁移率晶体管关断。
在反向充电(内部节点V BUS接收的充电电压需要传输至外部节点V USB)的情况下，即需要增强型双向氮化镓高电子迁移率晶体管导通以及保持导通状态时，驱动模块22基于参考点R2(D2)处的电压输出第三驱动电压，其中，第三驱动电压与采集的参考点R2(D2)处的电压的差值为第三差值，第三差值大于增强型双向氮化镓高电子迁移率晶体管的阈值电压。又因为第一极D1处的电压的最大值等于参考点R2(D2)处的电压，所以第一栅极G1的电压与第一极D1的电压的差值必然大于增强型双向氮化镓高电子迁移率晶体管的阈值电压，这样一来，反向充电时，可以保证增强型双向氮化镓高电子迁移率晶体管一直处于导通状态，进而保证反向充电的正常进行。
当充电完成等情况下，需要使增强型双向氮化镓高电子迁移率晶体管由导通状态变为关断状态时，驱动模块22向第一栅极G1输出第四驱动电压，第四驱动电压例如为0。由于第四驱动电压为0V，而即便由于增强型双向氮化镓高电子迁移率晶体管关断使得第一极D1处的电压下降，第一极D1处的电压也会大于或等于0V，即第四驱动电压(0V)和第一极D1(大于或等于0V)的差值必然会小于或等于0，亦即第一栅极G1的电压与第一极D1的电压的差值必然会小于或等于0，小于增强型双向氮化镓高电子迁移率晶体管的阈值电压。因此，实现了增强型双向氮化镓高电子迁移率晶体管的彻底关断。
通过上述介绍可知，设置一个增强型双向氮化镓高电子迁移率晶体管即可实现双向导通和关断功能，结构简单。此外，由于增强型氮化镓高电子迁移率晶体管本身具有导通阻抗低的特性，因此，增强型氮化镓高电子迁移率晶体管的设置，可以减少开关电路的损耗，降低开关电路的功耗，以及缓解发热。此外，在正向充电、正向关断以及反向充电时，驱动模块基于参考点的电压输出驱动电压，简化驱动模块的运算过程。
下面结合电子设备对图4所示的开关电路20实现双向导通与关断功能的具体原理进行介绍。其中，以充电电压为20V和增强型双向氮化镓高电子迁移率晶体管的阈值电压为1.5V为例进行的说明。
示例性的，结合图2、图4和图5，当对外接口10接入的外接设备为充电器200，即充电器200需要为手机100进行充电时，亦即外部节点V USB接收的充电电压需要传输至内部节点V BUS时，需要使增强型双向氮化镓高电子迁移率晶体管由关断状态变为导通状态。当增强型双向氮化镓高电子迁移率晶体管为关断状态时，外部节点V USB处的电压为充电电压20V，此时，内部节点V BUS为0V，第一极D1和外部节点V USB处的电压相同，第一栅极G1、第二极D2以及参考点R2与内部节点V BUS的电压相同。为了 使得增强型双向氮化镓高电子迁移率晶体管导通，驱动模块22会实时采集参考点R2的电压(与第二极D2处的电压相同)并基于参考点R2进行运算，使得输出的第一驱动电压与采集到的参考点R2的电压差值实时保持在第一差值5V。因此当驱动模块22采集参考点R2的电压为0V，驱动模块22向第一栅极G1发送5V的第一驱动电压。此时第一栅极G1的电压(5V)和第二极D2的电压(0V)的差值为5V，大于增强型双向氮化镓高电子迁移率晶体管的阈值电压1.5V，增强型双向氮化镓高电子迁移率晶体管导通。第二极D2的电压由0V逐渐上升为20V。由于驱动模块22会实时采集参考点R2(第二极D2)的电压，并基于采集的参考点R2(第二极D2)的电压实时调节输出至第一栅极G1的第一驱动电压，其中，第一驱动电压和参考点R2(第二极D2)的电压的差值一直保持在第一差值5V。所以当参考点R2(第二极D2)的电压逐渐上升为20V时，驱动模块22也会基于参考点R2(第二极D2)的电压实时调节其输出的第一驱动电压由5V逐渐上升为25V。这样一来，第一栅极G1和第二极D2的压差保持在5V，即大于增强型双向氮化镓高电子迁移率晶体管的阈值电压1.5V，在充电过程中，使得增强型双向氮化镓高电子迁移率晶体管一直处于导通的状态。外部节点V USB的充电电压20V通过导通增强型双向氮化镓高电子迁移率晶体管传输至内部节点V BUS，以通过内部节点V BUS传输至充电转换芯片30。充电转换芯片30对充电电压进行转换，并将转换后的电压传输至电池40，以实现对电池40的充电。此外，还将转换后的电压传输至片上系统(SoC芯片)50，以为片上系统(SoC芯片)50内部集成的CPU、GPU和Modem等供电，保证CPU、GPU和Modem的正常工作。
当充电完成；或者，在充电过程中遇到过大的电压会对手机100造成损伤等异常情况时，需要增强型双向氮化镓高电子迁移率晶体管由导通状态变为关断状态。当增强型双向氮化镓高电子迁移率晶体管为导通状态时，外部节点V USB、内部节点V BUS、第一极D1、第二极D2以及参考点R2的电压均为充电电压20V，第一栅极G1的电压为25V。为了使得增强型双向氮化镓高电子迁移率晶体管关断，驱动模块22会实时采集参考点R2的电压(与第二极D2处的电压相同)并基于参考点R2进行运算，使得输出的第一驱动电压与采集到的参考点R2的电压差值实时保持在第二差值0V。因此当驱动模块22采集参考点R2的电压为20V，驱动模块22向第一栅极G1发送20V的第二驱动电压。此时，第一栅极G1的电压和第二极D2的电压的差值为0V，小于增强型双向氮化镓高电子迁移率晶体管的阈值电压1.5V，增强型双向氮化镓高电子迁移率晶体管关断。且由于驱动模块22会实时采集参考点R2(第二极D2)的电压，并基于采集的参考点R2(第二极D2)的电压实时调节输出至第一栅极G1的第二驱动电压，其中，第二驱动电压和参考点R2(第二极D2)的电压的差值一直保持在第二差值0V。即便增强型双向氮化镓高电子迁移率晶体管关断后，参考点R2(第二极D2)的电压会下降至0V，驱动模块22输出的第二驱动电压也会随之下降，使得第一开关模块21的第一栅极G1和第二极D2的电压的差值一直保持在0V，保证增强型双向氮化镓高电子迁移率晶体管一直处于关断状态。这样一来，充电器200输出的充电电压无法传输至充电转换芯片30。充电转换芯片30不能为电池40供电以及为片上系统(SoC芯片)50内部集成的CPU、GPU和Modem等供电。
示例性的，结合图3、图4和图5，当对外接口10接入的外接设备为待充电手机300，即手机100需要为待充电手机300进行充电时，亦即内部节点V BUS接收的充电电压需要传输至外部节点V USB时，需要使增强型双向氮化镓高电子迁移率晶体管由关断状态变为导通状态。当增强型双向氮化镓高电子迁移率晶体管为关断状态时，内部节点V BUS处的电压为充电电压20V，此时，外部节点V USB为0V，第一极D1和外部节点V USB的电压相同，参考点R2、第二极D2与内部节点V BUS处的电压相等。为了使得增强型双向氮化镓高电子迁移率晶体管导通，驱动模块22会实时采集参考点R2(第二极D2)的电压，并基于采集的参考点R2(第二极D2)进行运算，使得输出的第三驱动电压与采集到的参考点R2的电压差值实时保持在第三差值5V。因此当驱动模块22采集参考点R2的电压为20V，驱动模块22通过控制端C向第一栅极G1发送25V的第三驱动电压。此时第一栅极G1的电压(25V)和第一极D1的电压(0V)的差值为25V，大于增强型双向氮化镓高电子迁移率晶体管的阈值电压1.5V，增强型双向氮化镓高电子迁移率晶体管导通，第一极D1的电压由0V逐渐上升为20V。由于参考点R2(第二极D2)的电压一直保持在20V，所以驱动模块22一直向第一栅极G1发送25V的第三驱动电压，第三驱动电压(25V)与第一极D1(最大为20V)的电压的差值大于或等于5V，大于增强型双向氮化镓高电子迁移率晶体管的阈值电压1.5V，在充电过程中，使得增强型双向氮化镓高电子迁移率晶体管一直处于导通的状态，保证充电的正常进行。
当充电完成；或者，在充电过程中遇到过大的电压会对待充电手机300造成损伤等异常情况时，需要增强型双向氮化镓高电子迁移率晶体管由导通状态变为关断状态。当增强型双向氮化镓高电子迁移率晶体管为导通状态时，内部节点V BUS、外部节点VUSB、第一极D1、第二极D2以及参考点R2的电压均为充电电压20V。第一栅极G1的电压例如为25V。为了使得增强型双向氮化镓高电子迁移率晶体管关断。此时驱动模块22不能基于参考点R2进行运算，使得输出的第四驱动电压与参考点R2的电压的差值保持在第四差值0V，即第四驱动电压与参考点R2的电压相等。这是因为如果驱动模块22基于参考点R2的电压输出第四驱动电压，由于参考点R2的电压与内部节点V BUS相同都是20V，那么驱动模块22输出的第四驱动电压为20V。而又因为第一极D1的电压会由于增强型双向氮化镓高电子迁移率晶体管关断而逐渐降低，例如当降低到15V时，第一栅极G1(20V)与第一极D1(15V)的差值为5V，增强型双向氮化镓高电子迁移率晶体管导通，不能实现增强型双向氮化镓高电子迁移率晶体管的关断。因此为了防止增强型双向氮化镓高电子迁移率晶体管在关断过程中出现导通的问题，本实施例中，驱动模块22直接输出0V的第四驱动电压。当驱动模块22输出0V的第四驱动电压至第一栅极G1时，由于即便第一极D1处的电压下降，第一极D1处的电压也会大于或等于0V，亦即第一栅极G1的电压(0V)与第一极D1的电压(大于或等于0V)的差值必然会小于或等于0，小于增强型双向氮化镓高电子迁移率晶体管的阈值电压1.5V。因此，实现了增强型双向氮化镓高电子迁移率晶体管的彻底关断。
又一个示例中，参见图6，与上一示例不同的是，开关电路20还包括第二开关模块23。第二开关模块23耦合于第二极D2和内部节点V BUS之间。第二开关模块23包括MOSFET，例如可以包括NMOSFET。NMOSFET包括第二栅极G2、源极S和漏极 D3。第二开关模块23的源极S与第一开关模块21的第二极D2耦合，第二开关模块23的漏极D3与内部节点V BUS耦合，第二开关模块23的第二栅极G2与第一开关模块21的第一栅极G1耦合。
设置第二开关模块23，可以使参考点R2与内部节点V BUS隔开，这样一来，不仅在正向充电、正向关断以及反向充电时，驱动模块22基于参考点R2的电压输出驱动电压，以使第一开关模块21在正向充电和反向充电时导通以及在正向关断时关断；且在反向关断时，驱动模块22可以基于参考点R2的电压输出驱动电压，以使第一开关模块21在反向关断时彻底关断，即在不同的阶段，驱动模块22都是基于参考点R2的电压输出驱动电压，如此，进一步简化驱动模块的运算过程。此外，虽然本实施例中开关电路20包括增强型双向氮化镓高电子迁移率晶体管以及NMOSFET，但是由于增强型双向氮化镓高电子迁移率晶体管本身具有导通阻抗低的特性，因此，相比于两个MOSFET，本实施例的开关电路20的导通阻抗降低，经验证，包括增强型双向氮化镓高电子迁移率晶体管以及NMOSFET的开关电路20相比于两个MOSFET的开关电路，导通阻抗可下降25％，这样一来，可以减少开关损耗，降低功耗，缓解发热。
此外，继续参见图6，开关电路20还包括电阻24，电阻24的一端与参考点R2耦合，电阻24的另一端接地设置。通过设置电阻24，在正向关断和反向关断时，可以使参考点R2的电压快速的下降至0V，驱动模块22基于参考点R2的电压也快速的下降至0V，即使得增强型双向氮化镓高电子迁移率晶体管和NMOSFET里的电荷快速的放掉，进而使得增强型双向氮化镓高电子迁移率晶体管和NMOSFET快速的关断。
示例性的，电阻24的电阻值例如大于100千欧。当设置电阻24的电阻值大于100千欧时，既不会使得参考点R2在对地进行放电时，电流过大，同时还可以使得参考点R2的电压较为快速的下降至0V。
需要说明的是，上述示例中以开关电路20中设置电阻24，且电阻24的一端接地设置为例进行的说明，但不构成对本申请的限定。可选的，为了实现增强型双向氮化镓高电子迁移率晶体管和NMOSFET快速的关断，参见图7，还可以将驱动模块22的参考端R1接地设置。通过参考端R1接地设置使得参考点R2的电压快速的下降至0V，以及驱动模块22基于参考点R2的电压也快速的下降至0V。
以下结合电子设备对图6所示的开关电路20实现双向导通与关断功能的具体原理进行介绍。其中，以第二开关模块23为NMOSFET，且NMOSFET的阈值电压为2V为例进行的说明。
图9是本实施例提供的正向充电时的仿真图，结合图2、图8和图9，当对外接口10接入的外接设备为充电器200，即充电器200需要为手机100进行充电时，亦即外部节点V USB接收的充电电压需要传输至内部节点V BUS时，需要使增强型双向氮化镓高电子迁移率晶体管和NMOSFET由关断状态变为导通状态。
关断阶段，即图9中在100μs之前，增强型双向氮化镓高电子迁移率晶体管和NMOSFET均关断，外部节点V USB处的电压为充电电压20V，此时，内部节点V BUS为0V，第一栅极G1和第二栅极G2也为0V，参考点R2、第二极D2和源极S的电压与内部节点V BUS相同。
导通阶段，为了使得增强型双向氮化镓高电子迁移率晶体管和NMOSFET导通，驱动模块22会实时采集参考点R2的电压(与第二极D2处的电压相同)并基于参考点R2进行运算，使得输出的第一驱动电压与采集到的参考点R2的电压差值实时保持在第一差值5V。因此当驱动模块22采集参考点R2的电压为0V，驱动模块22在100μs时逐渐向第一栅极G1和第二栅极G2施加第一驱动电压至5V。通过图9可知，在101.8μs时，第一栅极G1和第二栅极G2施加的第一驱动电压开始大于2V，即第一栅极G1(大于2V)和第二极D2(0V)的电压差值大于增强型双向氮化镓高电子迁移率晶体管的阈值电压1.5V，增强型双向氮化镓高电子迁移率晶体管导通，第二栅极G1(大于2V)和源极S(0V)的电压差值大于NMOSFET的阈值电压2V，NMOSFET导通，内部节点V BUS处开始有电压。
上升阶段，101.8μs到105μs之间，增强型双向氮化镓高电子迁移率晶体管和NMOSFET均导通后，内部节点V BUS处的电压从0V上升至20V。当然参考点R2处的电压也会从0V上升至20V。由于驱动模块22会实时采集参考点R2的电压(与第二极D2处的电压相同)并基于参考点R2进行运算，使得输出的第一驱动电压与采集到的参考点R2的电压差值实时保持在第一差值5V。因此，当参考点R2处的电压也会从0V上升至20V，驱动模块22的控制端C输出的第一驱动电压会逐渐上升为25V。
导通保持阶段，105μs之后，内部节点V BUS和参考点R2处的电压均保持在20V，驱动模块22的控制端C输出的第一驱动电压也保持在25V，这样一来，第一栅极G1(25V)和第二极D2(20V)的压差保持在5V，即大于增强型双向氮化镓高电子迁移率晶体管的阈值电压1.5V，第二栅极G2(25V)和源极S(20V)的压差保持在5V，即大于NMOSFET的阈值电压2V，在充电过程中，使得增强型双向氮化镓高电子迁移率晶体管和NMOSFET一直处于导通的状态。外部节点V USB的充电电压20V通过导通增强型双向氮化镓高电子迁移率晶体管传输至内部节点V BUS，以通过内部节点V BUS传输至充电转换芯片30。充电转换芯片30对充电电压进行转换，并将转换后的电压传输至电池40，以实现对电池40的充电。此外，还将转换后的电压传输至片上系统(SoC芯片)50，以为片上系统(SoC芯片)50内部集成的CPU、GPU和Modem等供电，保证CPU、GPU和Modem的正常工作。
图10是本实施例提供的正向关断时的仿真图，结合图2、图8和图10，当充电完成；或者，在充电过程中遇到过大的电压会对手机100造成损伤等异常情况时，需要增强型双向氮化镓高电子迁移率晶体管和NMOSFET由导通状态变为关断状态。
导通阶段，即图10中在305μs之前，增强型双向氮化镓高电子迁移率晶体管和NMOSFET均导通，外部节点V USB、内部节点V BUS、第二极D2、源极S、漏极D3以及参考点R2的电压均为充电电压20V。第一栅极G1和第二栅极G2的电压为25V。
关断阶段，为了使得增强型双向氮化镓高电子迁移率晶体管和NMOSFET关断，驱动模块22会实时采集参考点R2的电压(与第二极D2处的电压相同)并基于参考点R2进行运算，使得输出的第二驱动电压与采集到的参考点R2的电压差值实时保持在第二差值0V。因此，在305μs时，驱动模块22输出的第二驱动电压逐渐下降至20V。通过图10可知，在307.4μs时，第一栅极G1和第二栅极G2施加的第二驱动电压下降至 21.5V，即第一栅极G1(21.5V)和第二极D2(20V)的电压差值等于增强型双向氮化镓高电子迁移率晶体管的阈值电压1.5V，第二栅极G1(21.5V)和源极S(20V)的电压差值小于NMOSFET的阈值电压2V，增强型双向氮化镓高电子迁移率晶体管和NMOSFET开始关断，内部节点V BUS处的电压开始下降。
下降阶段，307.4μs到312.3μs之间，由于增强型双向氮化镓高电子迁移率晶体管和NMOSFET在关断时，并不能一下彻底关断。因此，内部节点V BUS处的电压从20V逐渐下降至0V。当然，参考点R2处的电压从20V逐渐下降至0V。由于驱动模块22会实时采集参考点R2的电压并基于参考点R2进行运算，使得输出的第二驱动电压与采集到的参考点R2的电压差值实时保持在第二差值0V。因此，当参考点R2处的电压从20V逐渐下降至0V，驱动模块22的控制端C输出的第二驱动电压随之会逐渐下降至0V。
关断保持阶段，312.3μs之后，内部节点V BUS和参考点R2处的电压均保持在0V，驱动模块22的控制端C输出的第二驱动电压也保持在0V，这样一来，第一栅极G1(0V)和第二极D2(0V)的压差保持在0V，即小于增强型双向氮化镓高电子迁移率晶体管的阈值电压1.5V，第二栅极G2(0V)和源极S(0V)的压差保持在0V，即小于NMOSFET的阈值电压2V，使得增强型双向氮化镓高电子迁移率晶体管和NMOSFET一直处于关断的状态。充电器200输出的充电电压无法传输至充电转换芯片30。充电转换芯片30不能为电池40供电以及为片上系统(SoC芯片)50内部集成的CPU、GPU和Modem等供电。
图11是本实施例提供的反向充电时的仿真图，结合图3、图8和图11，当对外接口10接入的外接设备为待充电手机300时，即手机100需要为待充电手机300进行充电时，亦即内部节点V BUS接收的充电电压需要传输至外部节点V USB时，需要使增强型双向氮化镓高电子迁移率晶体管和NMOSFET由关断状态变为导通状态。
关断阶段，即图11中在101μs之前，增强型双向氮化镓高电子迁移率晶体管和NMOSFET均关断，内部节点V BUS处的电压为充电电压20V，此时，外部节点V USB为0V，第一栅极G1和第二栅极G2也为0V，参考点R2、第二极D2和源极S的电压与外部节点V USB相同。
导通阶段，为了使得增强型双向氮化镓高电子迁移率晶体管和NMOSFET导通，驱动模块22会实时采集参考点R2的电压(与第二极D2处的电压相同)并基于参考点R2进行运算，使得输出的第三驱动电压与采集到的参考点R2的电压差值实时保持在第三差值5V。因此当驱动模块22采集参考点R2的电压为0V，驱动模块22在101μs时逐渐向第一栅极G1和第二栅极G2施加第一驱动电压至5V。通过图11可知，在104.5μs时，第一栅极G1和第二栅极G2施加的第一驱动电压开始大于2V，即第一栅极G1(大于2V)和第一极D1(0V)的电压差值大于增强型双向氮化镓高电子迁移率晶体管的阈值电压1.5V，增强型双向氮化镓高电子迁移率晶体管导通，第二栅极G1(大于2V)和源极S(0V)的电压差值大于NMOSFET的阈值电压2V，NMOSFET导通，外部节点V USB处开始有电压。
上升阶段，104.5μs到105μs之间，增强型双向氮化镓高电子迁移率晶体管和 NMOSFET均导通后，外部节点V USB处的电压从0V上升至20V。当然参考点R2处的电压也会从0V上升至20V。由于驱动模块22会实时采集参考点R2的电压并基于参考点R2进行运算，使得输出的第三驱动电压与采集到的参考点R2的电压差值实时保持在第三差值5V。因此，当参考点R2处的电压也会从0V上升至20V，驱动模块22的控制端C输出的第三驱动电压也会随之逐渐上升为25V。
导通保持阶段，110μs之后，外部节点V USB和参考点R2处的电压均保持在20V，驱动模块22的控制端C输出的第三驱动电压也保持在25V，这样一来，第一栅极G1(25V)和第一极D1(20V)的压差保持在5V，即大于增强型双向氮化镓高电子迁移率晶体管的阈值电压1.5V，第二栅极G2(25V)和源极S(20V)的压差保持在5V，即大于NMOSFET的阈值电压2V，在充电过程中，使得增强型双向氮化镓高电子迁移率晶体管和NMOSFET一直处于导通的状态，保证充电的正常进行。
图12是本实施例提供的反向关断时的仿真图，结合图3、图8和图12，当充电完成；或者，在充电过程中遇到过大的电压会对手机100造成损伤等异常情况时，需要增强型双向氮化镓高电子迁移率晶体管和NMOSFET由导通状态变为关断状态。
导通阶段，即图12中在310μs之前，增强型双向氮化镓高电子迁移率晶体管和NMOSFET均导通，外部节点V USB、内部节点V BUS、第二极D2、源极S、漏极D3以及参考点R2的电压均为充电电压20V。第一栅极G1和第二栅极G2的电压为25V。
关断阶段，为了使得增强型双向氮化镓高电子迁移率晶体管和NMOSFET关断，驱动模块22会实时采集参考点R2的电压(与第二极D2处的电压相同)并基于参考点R2进行运算，使得输出的第四驱动电压与采集到的参考点R2的电压差值实时保持在第四差值0V。因此，在310μs时，驱动模块22输出的第四驱动电压逐渐下降至20V。通过图12可知，在310.7μs时，第一栅极G1和第二栅极G2施加的第四驱动电压下降至21.5V，即第一栅极G1(21.5V)和第一极D1(20V)的电压差值等于增强型双向氮化镓高电子迁移率晶体管的阈值电压1.5V，第二栅极G1(21.5V)和源极S(20V)的电压差值小于NMOSFET的阈值电压2V，增强型双向氮化镓高电子迁移率晶体管和NMOSFET开始关断，内部节点V BUS处的电压开始从20V下降。
下降阶段，310.7μs到350μs之间，由于增强型双向氮化镓高电子迁移率晶体管和NMOSFET在关断时，并不能一下彻底关断。因此，内部节点V BUS处的电压从20V逐渐下降至0V。当然，参考点R2处的电压从20V逐渐下降至0V。由于驱动模块22会实时采集参考点R2的电压并基于参考点R2进行运算，使得输出的第四驱动电压与采集到的参考点R2的电压差值实时保持在第四差值0V。因此，当参考点R2处的电压从20V逐渐下降至0V，驱动模块22的控制端C输出的第四驱动电压随之会逐渐下降至0V。
关断保持阶段，350μs之后，内部节点V BUS和参考点R2处的电压均保持在0V，驱动模块22的控制端C输出的第四驱动电压也保持在0V，这样一来，第一栅极G1(0V)和第一极D1(0V)的压差保持在0V，即小于增强型双向氮化镓高电子迁移率晶体管的阈值电压1.5V，第二栅极G2(0V)和源极S(0V)的压差保持在0V，即小于NMOSFET的阈值电压2V，使得增强型双向氮化镓高电子迁移率晶体管和NMOSFET一直处于关断的状态。实现了增强型双向氮化镓高电子迁移率晶体管的彻底 关断。
再一个示例中，参见图13，与上一示例不同的是，第二开关模块23的设置位置。具体的，第二开关模块23的漏极D3分别与第一开关模块21的第二极D2以及内部节点V BUS耦合，第二开关模块23的源极S与参考点R2耦合，第二开关模块23的第二栅极G2与第一开关模块21的第一栅极G1耦合。
由于第二开关模块23并未位于充电通路上，因此第二开关模块23例如可以为小功率MOSFET，例如可以为小功率Si NMOSFET。示例性的，第二开关模块22的导通阻抗的范围是大于或等于1欧姆，且小于或等于100欧姆；第二开关模块22栅极到源极的耐压范围是大于或等于5V，且小于或等于60V；漏源击穿电压的范围大于或等于10V，且小于或等于120V。这样一来，本示例除了具有上述示例所具有的有益效果外，由于第二开关模块23可以为小功率MOSFET，所以本实施例的开关电路20可以进一步降低导通阻抗。经验证，本实施例的开关电路20可使导通阻抗下降40％以上，这样一来，可以进一步减少开关损耗，降低功耗，缓解发热。
此外，与上述示例相同，参见图13，开关电路20可以包括电阻24。其中，电阻24的一端与参考点R2耦合，电阻24的另一端接地设置。在其他可选实施例中，参见图14，还可以将驱动模块22的参考端R1接地设置。
以下结合电子设备对图13所示的开关电路20实现双向导通与关断功能的具体原理进行介绍。其中，以第二开关模块23为NMOSFET，且NMOSFET的阈值电压为2V为例进行的说明。
图16是本实施例提供的正向充电时的仿真图，结合图2、图15和图16，当对外接口10接入的外接设备为充电器200，即充电器200需要为手机100进行充电时，亦即外部节点V USB接收的充电电压需要传输至内部节点V BUS时，需要使增强型双向氮化镓高电子迁移率晶体管和NMOSFET由关断状态变为导通状态。
关断阶段，即图16中在100μs之前，增强型双向氮化镓高电子迁移率晶体管和NMOSFET均关断，外部节点V USB处的电压为充电电压20V，此时，内部节点V BUS为0V，第一栅极G1和第二栅极G2也为0V，参考点R2、第二极D2和源极S的电压与内部节点V BUS相同。
导通阶段，为了使得增强型双向氮化镓高电子迁移率晶体管和NMOSFET导通，驱动模块22会实时采集参考点R2的电压(与第二极D2处的电压相同)并基于参考点R2进行运算，使得输出的第一驱动电压与采集到的参考点R2的电压差值实时保持在第一差值5V。因此，当驱动模块22采集参考点R2的电压为0V时，驱动模块22在100μs时逐渐向第一栅极G1和第二栅极G2施加第一驱动电压至5V。通过图16可知，在100.8μs时，第一栅极G1和第二栅极G2施加的第一驱动电压开始大于2V，即第一栅极G1(大于2V)和第二极D2(0V)的电压差值大于增强型双向氮化镓高电子迁移率晶体管的阈值电压1.5V，增强型双向氮化镓高电子迁移率晶体管导通，第二栅极G1(大于2V)和源极S(0V)的电压差值大于NMOSFET的阈值电压2V，NMOSFET导通，内部节点V BUS处开始有电压。
上升阶段，100.8μs到102μs之间，增强型双向氮化镓高电子迁移率晶体管和 NMOSFET均导通后，内部节点V BUS处的电压从0V上升至20V。当然参考点R2处的电压也会从0V上升至20V。由于驱动模块22会实时采集参考点R2的电压(与第二极D2处的电压相同)并基于参考点R2进行运算，使得输出的第一驱动电压与采集到的参考点R2的电压差值实时保持在第一差值5V。因此，当参考点R2处的电压也会从0V上升至20V，驱动模块22的控制端C输出的第一驱动电压也会随之逐渐上升为25V。
导通保持阶段，102μs之后，内部节点V BUS和参考点R2处的电压均保持在20V，驱动模块22的控制端C输出的第一驱动电压也保持在25V，这样一来，第一栅极G1(25V)和第二极D2(20V)的压差保持在5V，即大于增强型双向氮化镓高电子迁移率晶体管的阈值电压1.5V，第二栅极G2(25V)和源极S(20V)的压差保持在5V，即大于NMOSFET的阈值电压2V，在充电过程中，使得增强型双向氮化镓高电子迁移率晶体管和NMOSFET一直处于导通的状态。外部节点V USB的充电电压20V通过导通增强型双向氮化镓高电子迁移率晶体管传输至内部节点V BUS，以通过内部节点V BUS传输至充电转换芯片30。充电转换芯片30对充电电压进行转换，并将转换后的电压传输至电池40，以实现对电池40的充电。此外，还将转换后的电压传输至片上系统(SoC芯片)50，以为片上系统(SoC芯片)50内部集成的CPU、GPU和Modem等供电，保证CPU、GPU和Modem的正常工作。
图17是本实施例提供的正向关断时的仿真图，结合图2、图15和图17，当充电完成；或者，在充电过程中遇到过大的电压会对手机100造成损伤等异常情况时，需要增强型双向氮化镓高电子迁移率晶体管和NMOSFET由导通状态变为关断状态。
导通阶段，即图17中在302μs之前，增强型双向氮化镓高电子迁移率晶体管和NMOSFET均导通，外部节点V USB、内部节点V BUS、第二极D2、源极S、漏极D3以及参考点R2的电压均为充电电压20V。第一栅极G1和第二栅极G2的电压为25V。
关断阶段，为了使得增强型双向氮化镓高电子迁移率晶体管和NMOSFET关断，驱动模块22会实时采集参考点R2的电压(与第二极D2处的电压相同)并基于参考点R2进行运算，使得输出的第二驱动电压与采集到的参考点R2的电压差值实时保持在第二差值0V。因此，在302μs时，驱动模块22输出的第二驱动电压逐渐下降至20V。通过图17可知，在303μs时，第一栅极G1和第二栅极G2施加的第二驱动电压下降至21.5V，即第一栅极G1(21.5V)和第二极D2(20V)的电压差值等于增强型双向氮化镓高电子迁移率晶体管的阈值电压1.5V，第二栅极G1(21.5V)和源极S(20V)的电压差值小于NMOSFET的阈值电压2V，增强型双向氮化镓高电子迁移率晶体管和NMOSFET开始关断，内部节点V BUS处的电压开始下降。
下降阶段，303μs到305μs之间，由于增强型双向氮化镓高电子迁移率晶体管和NMOSFET在关断时，并不能一下彻底关断。因此，内部节点V BUS处的电压从20V逐渐下降至0V。当然，参考点R2处的电压从20V逐渐下降至0V。由于驱动模块22会实时采集参考点R2的电压并基于参考点R2进行运算，使得输出的第二驱动电压与采集到的参考点R2的电压差值实时保持在第二差值0V。因此，当参考点R2处的电压从20V逐渐下降至0V，驱动模块22的控制端C输出的第二驱动电压随之会逐渐下降至0V。
关断保持阶段，305μs之后，内部节点V BUS和参考点R2处的电压均保持在0V， 驱动模块22的控制端C输出的第二驱动电压也保持在0V，这样一来，第一栅极G1(0V)和第二极D2(0V)的压差保持在0V，即小于增强型双向氮化镓高电子迁移率晶体管的阈值电压1.5V，第二栅极G2(0V)和源极S(0V)的压差保持在0V，即小于NMOSFET的阈值电压2V，使得增强型双向氮化镓高电子迁移率晶体管和NMOSFET一直处于关断的状态。充电器200输出的充电电压无法传输至充电转换芯片30。充电转换芯片30不能为电池40供电以及为片上系统(SoC芯片)50内部集成的CPU、GPU和Modem等供电。
图18是本实施例提供的反向充电时的仿真图，结合图3、图15和图18，当对外接口10接入的外接设备为待充电手机300时，即手机100需要为待充电手机300进行充电时，亦即内部节点V BUS接收的充电电压需要传输至外部节点V USB时，需要使增强型双向氮化镓高电子迁移率晶体管和NMOSFET由关断状态变为导通状态。
关断阶段，即图18中在100.4μs之前，增强型双向氮化镓高电子迁移率晶体管和NMOSFET均关断，内部节点V BUS处的电压为充电电压20V，此时，外部节点V USB为0V，第一栅极G1和第二栅极G2也为0V，参考点R2、第二极D2和源极S的电压与外部节点V USB相同。
导通阶段，为了使得增强型双向氮化镓高电子迁移率晶体管和NMOSFET导通，驱动模块22会实时采集参考点R2的电压(与第二极D2处的电压相同)并基于参考点R2进行运算，使得输出的第三驱动电压与采集到的参考点R2的电压差值实时保持在第三差值5V。因此当驱动模块22采集参考点R2的电压为0V，驱动模块22在100.4μs时逐渐向第一栅极G1和第二栅极G2施加第一驱动电压至5V。通过图18可知，在100.48μs时，第一栅极G1和第二栅极G2施加的第一驱动电压开始大于2V，即第一栅极G1(大于2V)和第一极D1(0V)的电压差值大于增强型双向氮化镓高电子迁移率晶体管的阈值电压1.5V，增强型双向氮化镓高电子迁移率晶体管导通，第二栅极G1(大于2V)和源极S(0V)的电压差值大于NMOSFET的阈值电压2V，NMOSFET导通，外部节点V USB处开始有电压。
上升阶段，100.48μs到101μs之间，增强型双向氮化镓高电子迁移率晶体管和NMOSFET均导通后，外部节点V USB处的电压从0V上升至20V。当然参考点R2处的电压也会从0V上升至20V。由于驱动模块22会实时采集参考点R2的电压并基于参考点R2进行运算，使得输出的第三驱动电压与采集到的参考点R2的电压差值实时保持在第三差值5V。因此，当参考点R2处的电压也会从0V上升至20V，驱动模块22的控制端C输出的第三驱动电压也会随之逐渐上升为25V。
导通保持阶段，101μs之后，外部节点V USB和参考点R2处的电压均保持在20V，驱动模块22的控制端C输出的第三驱动电压也保持在25V，这样一来，第一栅极G1(25V)和第一极D1(20V)的压差保持在5V，即大于增强型双向氮化镓高电子迁移率晶体管的阈值电压1.5V，第二栅极G2(25V)和源极S(20V)的压差保持在5V，即大于NMOSFET的阈值电压2V，在充电过程中，使得增强型双向氮化镓高电子迁移率晶体管和NMOSFET一直处于导通的状态，保证充电的正常进行。
图19是本实施例提供的反向关断时的仿真图，结合图3、图15和图19，当充电完 成；或者，在充电过程中遇到过大的电压会对手机100造成损伤等异常情况时，需要增强型双向氮化镓高电子迁移率晶体管和NMOSFET由导通状态变为关断状态。
导通阶段，即图12中在301μs之前，增强型双向氮化镓高电子迁移率晶体管和NMOSFET均导通，外部节点V USB、内部节点V BUS、第二极D2、源极S、漏极D3以及参考点R2的电压均为充电电压20V。第一栅极G1和第二栅极G2的电压为25V。
关断阶段，为了使得增强型双向氮化镓高电子迁移率晶体管和NMOSFET关断，驱动模块22会实时采集参考点R2的电压(与第二极D2处的电压相同)并基于参考点R2进行运算，使得输出的第四驱动电压与采集到的参考点R2的电压差值实时保持在第四差值0V。因此，在301μs时，驱动模块22输出的第四驱动电压逐渐下降至20V。通过图12可知，在302μs时，第一栅极G1和第二栅极G2施加的第四驱动电压下降至21.5V，即第一栅极G1(21.5V)和第一极D1(20V)的电压差值等于增强型双向氮化镓高电子迁移率晶体管的阈值电压1.5V，第二栅极G1(21.5V)和源极S(20V)的电压差值小于NMOSFET的阈值电压2V，增强型双向氮化镓高电子迁移率晶体管和NMOSFET开始关断，内部节点V BUS处的电压开始从20V下降。
下降阶段，302μs到309.6μs之间，由于增强型双向氮化镓高电子迁移率晶体管和NMOSFET在关断时，并不能一下彻底关断。因此，内部节点V BUS处的电压从20V逐渐下降至0V。当然，参考点R2处的电压从20V逐渐下降至0V。由于驱动模块22会实时采集参考点R2的电压并基于参考点R2进行运算，使得输出的第四驱动电压与采集到的参考点R2的电压差值实时保持在第四差值0V。因此，当参考点R2处的电压从20V逐渐下降至0V，驱动模块22的控制端C输出的第四驱动电压随之会逐渐下降至0V。
关断保持阶段，309.6μs之后，内部节点V BUS和参考点R2处的电压均保持在0V，驱动模块22的控制端C输出的第四驱动电压也保持在0V，这样一来，第一栅极G1(0V)和第一极D1(0V)的压差保持在0V，即小于增强型双向氮化镓高电子迁移率晶体管的阈值电压1.5V，第二栅极G2(0V)和源极S(0V)的压差保持在0V，即小于NMOSFET的阈值电压2V，使得增强型双向氮化镓高电子迁移率晶体管和NMOSFET一直处于关断的状态。实现了增强型双向氮化镓高电子迁移率晶体管的彻底关断。
以上所述，以上实施例仅用以说明本申请的技术方案，而非对其限制；尽管参照前述实施例对本申请进行了详细的说明，本领域的普通技术人员应当理解：其依然可以对前述各实施例所记载的技术方案进行修改，或者对其中部分技术特征进行等同替换；而这些修改或者替换，并不使相应技术方案的本质脱离本申请各实施例技术方案的范围。

Claims (11)

  1. 一种开关电路，其特征在于，包括：外部节点、内部节点、第一开关模块和驱动模块；
    所述第一开关模块包括增强型氮化镓高电子迁移率晶体管，所述增强型氮化镓高电子迁移率晶体管包括第一栅极、第一极和第二极；所述驱动模块包括控制端；
    所述第一栅极与所述控制端耦合，所述第一极和所述外部节点耦合，所述内部节点和所述第二极耦合；
    所述外部节点用于接收充电电压，并将所述充电电压传输至所述增强型氮化镓高电子迁移率晶体管；所述驱动模块用于控制所述增强型氮化镓高电子迁移率晶体管导通或关断，当所述增强型氮化镓高电子迁移率晶体管导通时，所述充电电压传输至所述内部节点，以通过所述内部节点传输至充电转换芯片；
    或者，
    所述内部节点用于接收充电电压，并将所述充电电压传输至所述增强型氮化镓高电子迁移率晶体管；所述驱动模块用于控制所述增强型氮化镓高电子迁移率晶体管导通或关断，当所述增强型氮化镓高电子迁移率晶体管导通时，所述充电电压传输至所述外部节点，以通过所述外部节点传输至待充电电子设备。
  2. 根据权利要求1所述的开关电路，其特征在于，所述驱动模块还包括参考端，所述第二极和所述参考端耦合于参考点；
    当所述外部节点接收所述充电电压时，所述驱动模块基于所述参考点的电压向所述第一栅极发送第一驱动电压，所述增强型氮化镓高电子迁移率晶体管响应于所述第一驱动电压导通，并将所述充电电压传输至所述内部节点；所述驱动模块还基于所述参考点的电压向所述第一栅极发送第二驱动电压，所述增强型氮化镓高电子迁移率晶体管响应于所述第二驱动电压关断；
    或者，
    当所述内部节点接收所述充电电压时，所述驱动模块基于所述参考点的电压向所述第一栅极发送第三驱动电压，所述增强型氮化镓高电子迁移率晶体管响应于所述第三驱动电压导通，并将所述充电电压传输至所述外部节点；所述驱动模块还向所述第一栅极发送第四驱动电压，所述增强型氮化镓高电子迁移率晶体管响应于所述第四驱动电压关断。
  3. 根据权利要求2所述的开关电路，其特征在于，还包括第二开关模块，所述第二开关模块用于当所述驱动模块向所述第一栅极发送第四驱动电压时阻止所述内部节点将接收到的充电电压传输至所述参考点；
    所述驱动模块基于所述参考点的电压向所述第一栅极发送第四驱动电压，所述增强型氮化镓高电子迁移率晶体管响应于所述第四驱动电压关断。
  4. 根据权利要求3所述的开关电路，其特征在于，所述第二开关模块包括N型金属氧化物半导体场效应晶体管；所述N型金属氧化物半导体场效应晶体包括第二栅极、源极和漏极；所述源极与所述第二极耦合，所述漏极与所述内部节点耦合。
  5. 根据权利要求3所述的开关电路，其特征在于，所述第二开关模块包括N型金属氧化物半导体场效应晶体管；所述N型金属氧化物半导体场效应晶体包括第二栅极、源极和漏极；所述漏极分别与所述第二极以及所述内部节点耦合，所述源极与所述参考点耦合。
  6. 根据权利要求5所述的开关电路，其特征在于，所述N型金属氧化物半导体场效应晶体管的导通阻抗的范围是大于或等于1欧姆，且小于或等于100欧姆；所述N型金属氧化物半导体场效应晶体管的第二栅极到源极的耐压范围是大于或等于5V，且小于或等于60V；所述N型金属氧化物半导体场效应晶体管的漏源击穿电压的范围大于或等于10V，且小于或等于120V。
  7. 根据权利要求4或5所述的开关电路，其特征在于，所述第一栅极与所述第二栅极耦合。
  8. 根据权利要求2所述的开关电路，其特征在于，所述开关电路还包括电阻；所述电阻的第一端与所述参考点耦合，所述电阻的第二端接地设置。
  9. 根据权利要求8所述的开关电路，其特征在于，所述电阻的电阻值大于100千欧姆。
  10. 根据权利要求1所述的开关电路，其特征在于，所述增强型氮化镓高电子迁移率晶体管包括增强型双向氮化镓高电子迁移晶体管。
  11. 一种电子设备，其特征在于，包括权利要求1-10任一项所述的开关电路。
PCT/CN2022/117882 2021-11-26 2022-09-08 开关电路及电子设备 Ceased WO2023093216A1 (zh)

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