Classifications

• • H—ELECTRICITY
  • H01—ELECTRIC ELEMENTS
  • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
  • H01P3/00—Waveguides; Transmission lines of the waveguide type
  • H01P3/02—Waveguides; Transmission lines of the waveguide type with two longitudinal conductors
  • H01P3/08—Microstrips; Strip lines
  • H01P3/081—Microstriplines
  • H01P3/082—Multilayer dielectric
• • H—ELECTRICITY
  • H01—ELECTRIC ELEMENTS
  • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
  • H01P1/00—Auxiliary devices
  • H01P1/10—Auxiliary devices for switching or interrupting
• • H—ELECTRICITY
  • H01—ELECTRIC ELEMENTS
  • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
  • H01P1/00—Auxiliary devices
  • H01P1/32—Non-reciprocal transmission devices
• • H—ELECTRICITY
  • H01—ELECTRIC ELEMENTS
  • H01P—WAVEGUIDES; RESONATORS, LINES, OR OTHER DEVICES OF THE WAVEGUIDE TYPE
  • H01P3/00—Waveguides; Transmission lines of the waveguide type
  • H01P3/16—Dielectric waveguides, i.e. without a longitudinal conductor
• • H—ELECTRICITY
  • H01—ELECTRIC ELEMENTS
  • H01Q—ANTENNAS, i.e. RADIO AERIALS
  • H01Q13/00—Waveguide horns or mouths; Slot antennas; Leaky-waveguide antennas; Equivalent structures causing radiation along the transmission path of a guided wave
  • H01Q13/20—Non-resonant leaky-waveguide or transmission-line antennas; Equivalent structures causing radiation along the transmission path of a guided wave

Definitions

• This invention relates generally to high frequency transmission lines and more particularly to planar type waveguide structures for millimeter wave applications.
• planar transmission lines are widely used to channel the flow of high-frequency electrical energy. Common examples of these transmission lines are the coaxial line, the hollow metallic waveguide and the optical fiber. All these waveguiding structures are useful in long link applications. However, in situations where the distance between the transmitting and receiving points is below a few inches, as in an integrated circuit, planar transmission lines offer an attractive alternative to these types of transmission lines. A variety of planar transmission line configurations are possible. In one type of planar transmission line, metallic conductors play a primary role in the waveguiding process. This includes the well known microstrip transmission line, the slotline, the coplanar waveguide, and coplanar strip line. In another type of planar transmission line, a dielectric strip plays a primary role in the waveguiding process.
• This invention is directed to a multi-layer dielectric slab structure including a substrate layer and a guiding layer, with the substrate layer being bounded on the bottom side by a metal ground plane.
• a dielectric loading strip metallized on the top face comprises the top layer and forms the remainder of the structure.
• the layers of dielectric are primarily chosen with respect to their permittivities to keep the propagating energy away from the conductor surfaces, and thereby reduce conductor losses.
• the conductive coating on the top surface of the dielectric loading strip permits single mode propagation over a relatively wide band and radiation losses due to coupling of the desired mode to the substrate modes are reduced and the polarization of the dominant mode is such as to render the structure of the transmission line relatively insensitive to small deviations from parallelism among the different interfaces of the structure.
• a whole family of devices evolves from such a structure. These include antennas, power couplers, switches, isolators and circulators.
• Figure 1A shows a perspective view of a known prior art microstrip line
• Figure 1B shows a perspective view of a known prior art slotline
• Figure 1C shows a perspective view of a known prior art coplanar waveguide
• Figure 1D shows a perspective view of a known prior art coplanar strip
• Figure 2A shows a perspective view of a known prior art dielectric strip guide
• Figure 2B shows a perspective view of a known prior art inverted dielectric strip guide
• Figure 3 is schematically illustrative of a guided wave mode in a slab type waveguide in accordance with the subject invention
• Figure 4 shows a perspective view of one embodiment of a compound dielectric multi-conductor transmission line in accordance with this invention
• Figure 5 shows a perspective view of an alternative embodiment of the transmission line shown in Figure 4;
• Figure 6 is a perspective view illustrative of a first embodiment of an antenna constructed in accordance with the waveguide structure shown in Figure 4;
• Figure 7 is a cross-sectional view of the antenna shown in Figure 6 taken along the lines 7-7 thereof and further illustrates the energy distribution across the interface of the dielectric layers to cause radiation;
• Figure 8 is a perspective view illustrative of a second embodiment of the antenna configured from the transmission line shown in Figure 5;
• Figure 9 is a perspective view illustrative of a first embodiment of an isolator device configured from the transmission line shown in Figure 4;
• Figure 10 is a perspective view illustrative of a second embodiment of an isolator device constructed from the embodiment of the transmission line shown in Figure 4;
• FIGS 11A and 11B are schematic diagrams helpful in understanding the operation of the isolator device as shown in Figures 9 and 10;
• Figure 12 is a perspective view illustrative of a circulator type of coupling device configured from the transmission line shown in Figure 4;
• Figure 13 is a schematic diagram illustrative of the pattern of the top loading strips formed on the device shown in Figure 12;
• Figure 14 is a partial cross sectional view illustrative of a modification of the embodiment shown in Figure 12;
• Figure 15 is a perspective view illustrative of an optically controlled switch device formed in accordance with the transmission line shown in Figure 4;
• Figure 16 is a perspective view illustrative of a device for providing a transition from rectangular waveguide to the subject transmission line as shown in Figure 4;
• Figures 17 and 18 are schematic diagrams helpful in understanding the operation of the transition device shown in Figure 16;
• Figure 19 is a perspective view illustrative of an embodiment for providing efficient coupling of power generated in an external device into the waveguide structure as shown in Figure 4;
• Figure 20 is a perspective view illustrative of another embodiment of a configuration of coupling power generated in an external device into the transmission line structure shown in Figure 4. Detailed Description of the Invention
• a mode is a spatial distribution of energy across the cross-section of the guiding structure.
• a waveguiding structure can propagate several modes. Each of the modes has a characteristic cut-off frequency below which the waveguiding structure will not support it. It is customary to choose the cross sectional dimensions of the waveguiding structure such that, over the frequency range of interest, only one mode will be supported. This mode is often the one with the lowest cut-off frequency and is called the dominant mode and the cut-off frequency for the next higher order mode represents the useful bandwidth for the waveguiding structure.
• the strip dielectric waveguide structure of Itoh disclosed in the above referenced U.S. Pat. 4,028,643, also has limitations. For one, its dominant mode is actually an admixture of two orthogonal polarization states. Consequently, some leakage via coupling to the parasitic substrate modes is unavoidable. Further, this coupling grows stronger as one progresses to lower frequencies since then, the difference between the guiding structure and the region external to it cannot be distinguished by the propagating energy. Additionally, the struc-ture has no inherent mechanism for propagating direct current energy. Considering now the drawings, in the structures shown in Figure 1A through 1D, which are illustrative of known prior art, metallic strips are of primary importance in the waveguiding process.
• a conducting strip 10 is mounted on a dielectric 11 which is coated on its bottom surface with a metallic ground plane 12.
• two parallel conductors 13 are placed upon dielectric 14.
• a coplanar waveguide configuration is shown in which three parallel conductors 15 are placed on dielectric 16, the two outer conductive strips acting as a ground plane.
• coplanar conductive strips 17 are mounted on dielectric 18, but the edges of the strips are not coextensive with the edges of the dielectric slab, as they were in the slotline of Figure 1B.
• microstrip structure shown in Figure 1A has proven the most versatile and successful among the prior art configurations using metallic strips.
• the microstrip type of transmission line has been successfully used in applications up to 60GHz, but even at those frequencies, some of the problems associated with its use are evident.
• the substrate modes are suppressed by choosing a dielectric substrate that is thin enough. At 60 GHz, a typical substrate thickness must not exceed 8 mils. At higher frequencies, even thinner substrates must be used.
• the impedance of a transmission line in microstrip is primarily determined by the ratio of the conductor strip width W, to the dielectric thickness h, i.e. w/h.
• the value of W is bounded at the upper end by the requirement that it be small compared to the wavelength of the propagating energy at the frequency in question.
• the lower bound on W is determined by the accuracy and reproducibility with which a narrow line can be fabricated.
• a fourth problem concerns the thermal properties of the structure. Ironically, this consideration leads to the conclusion that the substrate is not thin enough. If the dielectric substrate is a semiconductor on which truly planar transmitting sources are inte ⁇ grated, then the heat generated within these sources would have to be removed if the device is to survive operation. Unfortunately, most electrical insulators are also thermal insulators, diamond and beryllium oxide being exceptions, and consequently, the heat generating device would be thermally isolated from a heat sink, unless the substrate was made very thin.
• Planar dielectric waveguides offer more convenient substrate and guide dimensions, and also have low loss.
• a dielectric strip guide is shown in which a dielectric strip 19 is mounted on a dielectric slab 20 which is coated on its bottom surface with metallic ground plane 21.
• An inverted strip guide is shown in Figure 2B in which a dielectric strip 22 is sandwiched between dielectric slab 23 and metallic ground plane 24.
• a key feature of planar dielectric structures is that they have very low loss at frequencies where the structures of Figure 1A- Figure 1D cannot be used at all.
• planar dielectric waveguides have been used at optical frequencies spanning the infrared to visible range. This is in part due to the outright absence of conductors or the relative remoteness of the conductor surfaces from the propagating energy.
• planar dielectric structures of Figures 2A and 2B suffer from the inherent limitation of being multi-modal in that they all support at least two modes. Any attempt to realize single mode operation usually results in a mode that is too weakly bound to the structure to be of any practical use.
• the coupling between these two modes can result in high radiation loss at discontinuities and bends, as well as increased coupling to the spurious substrate modes mentioned in connection with microstrip and dielectric strip waveguide.
• planar dielectric structures are their extreme sensitivity to the condition of the interface.
• any roughness in the surfaces of the guiding or cladding media or any bubbles trapped between them during the bonding process can have a profound influence on the losses due to random scattering from these centers at the boundaries.
• a substrate 30 whose permittivity is ⁇ s and whose thickness is d s is clad on one side, i.e. the bottom face, by a metal ground layer 31.
• the other side or top face of the substrate 30 is bonded to a dielectric guiding slab layer 32 whose permittivity is ⁇ g and whose thickness is h.
• a relatively narrower dielectric loading strip 33 of width W, thickness d 1 and permittivity ⁇ 1 is bonded to the other side or top face of guiding slab 32.
• the propagating direction for the electrical energy is along its longitudinal axis.
• the upper face of the loading strip 33 is clad with a relatively thin metal layer or coating 34, which covers at least one third of the width W of the loading strip and extends uniformly, periodically or aperiodically along its length depending on the needs of the user.
• the permittivity ⁇ g of the guiding layer 32 is made to be greater than both the permittivity ⁇ S of the substrate 30 and the permittivity ⁇ 1 of strip 33 due to the fact that energy being propagated seeks the medium with the highest permittivity.
• the nature and thickness of metal ground layer 31 and metal cladding layer 34 are not critical.
• a waveguide structure in accordance with the configuration shown in Figure 4 was constructed with a guiding slab layer 32 of RT "Duroid" 6010 and both the substrate dielectric 30 and strip 33 of alumina. "Duroid” is a trademark of Rogers Corp. for filled tetrafluoroeythylene material.
• ⁇ 1 ⁇ s the preferred ratio between each of the thicknesses d s and d 1 , of the substrate and strip and that of the guiding slab layer h is approximately 0.75. In the example described above, for practical reasons the thickness ratio was altered to 0.8.
• the line loss in this waveguide was measured at 94GHz and was found to be only 0.4db/inch, compared with a loss of 2.5db/inch at that frequency for microstrip, an improvement of almost six to one.
• the transmission line of Figure 4 combines the wideband feature of microstrip and the low loss characteristic of planar dielectric waveguides. Like the planar dielectric waveguide, it has no "sidewalls" so that scattering losses are reduced. Ohmic conductor losses are reduced substantially below those in an equivalent microstrip structure, permitting operation at higher frequencies. Further, the amount by which they are reduced increases as the frequency is increased. In the example presented, the conductor losses are 56% of their microstrip contributions at 75GHz. At 100GHz, they are 33% of their microstrip contributions. This is a significant result since conductor losses are known to increase with increasing frequency.
• the dominant TM 0 mode is not widely separated from the TE 0 mode.
• the addition of metal conductors 31 and 34 in the manner shown is particularly significant in the instant invention in that the separation between these modes is widened, thus permitting single-mode operation over a wider band.
• the condition for mode separation is somewhat more complicated but nevertheless calculable when ⁇ s ⁇ ⁇ 1 .
• the dominant mode is the TM 0 mode whose polarization is such that, the electric field is largely at right angles to the metal conductors as required. This entire phenomenon was not previously anticipated.
• the top conducting strip 34 serves as a reflector to confine the propagating energy to region 1 of the dielectric layers even at low frequencies.
• the substrate modes in the "wings" of the structure (region 2) have no such confinement means. Consequently, the effective dielectric constant of the wings will decrease with frequency whereas it will remain substantially constant in the region under the strip. This difference in the effective dielectric constants is what produces the confinement of the propagating energy which will therefore be well guided even at low frequencies and dc.
• the thickness of the microstrip substrate 11 of the known prior structure shown in Figure 1 is limited because of the need to suppress the spurious substrate modes, as indicated previously.
• the most troublesome among these modes is the TE 0 mode for a grounded dielectric slab.
• the structure shown in Figure 4 suppresses the propagation of this TE 0 mode, thus permitting the use of thicker substrates 30 at a given operating frequency than would be possible with a comparable microstrip guide. As a result, losses at waveguide bends and waveguide discontinuities will be greatly reduced.
• the dimensions of the waveguide ( Figure 4) at the frequency range of 75-100GHz will be larger than those of a microstrip structure ( Figure 1) designed for that range.
• the choices of the guiding layer 32 thickness h, the substrate dielectric 30 thickness d s , and the loading strip 33 thickness d 1 are determined by the desired frequency of operation and by the dielectric permittivities ⁇ g , ⁇ s and ⁇ 1 .
• ⁇ 1 6.6 ⁇ 0 which corresponds to BeO
• ⁇ g 12.9 ⁇ 0 which corresponds to GaAs
• the advantage of thicker substrate material is surrendered somewhat if a large dielectric discontinuity exists at the relevant interfaces.
• Any or all of the dielectric elements 30, 32 and 33 of the structure shown in Figure 4 may be semiconductors.
• Another unexpected result that emerges from a consideration of the new structure, particularly when, semiconductors are used, is a method for exciting the dominant mode.
• the conventional shunt and series excitation in microstrip line are well known.
• the excitation source may be located at the interface 29 between guiding layer 32 and substrate dielectric 30 or at the interface 31 between guiding layer 32 and strip 33.
• the excitation source would be oriented with its current transport direction parallel to the desired direction of propagation of the energy i.e. parallel to the longitudinal axis of the strip.
• the transmission line of Figure 4 resembles microstrip with the closeness of the resemblance under the designer's control. At low frequencies, its behavior is identical to microstrip.
• the structure of the present invention may be viewed as a means to extend the frequency of operation of microstrip circuits without having to change the substrate thickness.
• a 70 mil thick conventional microstrip configuration is only usable from dc to 14GHz while the 70 mil compound dielectric slab in accordance with the invention and as presented in the design example above is usable from dc to 100GHz.
• the characteristic impedance of the transmission line shown in Figure 4 is determined primarily by the ratio of the width W of the loading strip 33 to the effective guiding layer thickness, (which is always somewhat larger than the actual thickness h), when the width is small compared to the wavelength.
• width W can also be used to provide impedance matching and frequency filtering. For widths comparable to or larger than the wavelength, no complicated field analysis is required to define the impedance level, which is dependent on the operating frequency. However, this change need not be very large. In the design example presented, a 50 ⁇ line at 75GHz becomes a 64 ⁇ line at 100GHz. Smaller variations in impedance are possible with alternative designs at the cost of higher line loss. In the structure of Figure 4 , an appreciable amount of energy is propagated in the guiding and the substrate layers 32 and 30 where no "sidewall" losses are manifested. A smaller proportion of energy is present in the loading strip 33. This energy, however, is subjected to sidewall scattering loss. Besides, the fields at the edge of the conducting strip 34 are relatively higher, so that the scattering losses could be greater.
• FIG. 5 shows the special case of a symmetric linear taper so that loading strip 36 has the cross section of an isoceles trapezoid.
• Strip 36 rests on guiding slab layer 37 which is in turn mounted on substrate dielectric layer 38.
• Ground plane 39 is coated on the bottom of layer 38 while a metal cladding layer 40 is formed on the top surface of the tapered loading strip 36.
• tapers may also be used for strip 36 such as concave and convex circular, concave and convex hyperbolic, exponential, etc.
• the technique of tapering the strip 36 has some additional latent advantages: (i) It permits a wider range of conductor linewidths and can be realized without running into the mechanical difficulty of having to mount very thin strips edge-on on the guiding layer; (ii) The tapered sides "soften" the discontinuity at the strip's edge. This has the effect of focusing the energy towards the center of the strip. This focusing effect is increased if the taper is such that the slope at any point on it relative to the vertical is greater than the critical angle for that interface. The tapered sides also increase the separation between the TE 0 and TM 0 modes beyond the effect previously described. Thus, a wider operating bandwidth is permitted. Alternatively, for a given operating bandwidth, the conductor losses may be reduced even further.
• One of the anticipated disadvantages is the sensitivity of the propagating energy to imperfections of the tapered sides. This is expected to be more critical for high impedance (narrow conductor width) lines. However, such sensitivity will have a smaller impact than any corresponding effect in a competitive planar dielectric waveguide.
• the wide bandwidth afforded by the waveguide of this invention makes the medium ideally suited for digital transmission.
• one or more of the dielectric layers 30, 32, 33 of Figure 4 may be replaced by a non-reciprocal medium, including, for example, a ferroelectric or ferrimagnetic material such as Barium titanate or a ferrite.
• a ferroelectric or ferrimagnetic material such as Barium titanate or a ferrite.
• the relatively small volumes in which the propagating waves are confined would enable one to use smaller amounts of control energy, and yet maintain the control energy density (energy/volume) at high enough levels to manipulate the guided energy. In practice, this means that one may use smaller magnetic field strengths to manipulate the high-frequency energy in devices such as ferrite phase shifters and modulators as well as in circulators and isolators.
• the heat dissipation problem outlined earlier can be more effectively overcome by using materials, such as BeO, that are electrical insulators but thermal conductors for the substrate dielectric and/or dielectrie strip. Since these materials can be brought in direct contact with the power-generating device, they can serve as a low thermal resistance path between the device and a heat sink.
• materials such as BeO, that are electrical insulators but thermal conductors for the substrate dielectric and/or dielectrie strip. Since these materials can be brought in direct contact with the power-generating device, they can serve as a low thermal resistance path between the device and a heat sink.
• FIGS. 6 and 8 are illustrative of two embodiments of antennas which can be configured from the basic transmission line structure shown in Figure 4.
• the dielectric loading strip 33 on top of the guiding layer 32 includes an abrupt transition to a region 42 of reduced thickness a which includes a plurality of transverse metal lines or strips 44 formed on the top surface thereof and which operate as a diffraction grating.
• the reduced region 42 can be made of a different dielectric which leads to consideration of the embodiment shown in Figure 8.
• a linear tapered transition of both the dielectric loading strip 33 as well as the top layer of metallization 34 is provided as shown by reference numerals 48 and 50, respectively, with a pointed radiator element comprised of dielectric having a permittivity ⁇ rad being provided as an extension of the layer 33.
• the permittivity ⁇ rad of the radiating element 52 is made to be greater than the oermittivities of both the guiding layer 32 as well as the permittivities ⁇ s and ⁇ 1 of the layers 30 and 33. Since energy propagating in the transmission line seeks the highest dielectric medium, the radiating element 52 in effect lifts the guided mode off of the guiding layer 32.
• the radiating element 52 as shown also includes a tapered nose section 54 so as to efficiently radiate energy to free space or alternatively, to couple energy efficiently from free space into the transmission line.
• the element 52 can be configured to include a suitable corrugation grid pattern or a combination thereof in a manner shown in Figure 6. It may also be possible to fabricate the radiating element 52 from materials which are alterable electronically so as to alter the direction of radiation and thereby provide a scanning mode.
• FIGS 9 and 10 shown thereat are two configurations of a device known as a signal isolator which permits the flow of a signal on one direction but not in the other direction.
• a section of an anisotropic material such as a ferrite or electro-optic medium into the transmission line structure of Figure 4.
• a section 56 of magnetically biased ferrite which is tapered at both ends is inserted in the length of dielectric loading strip 33 beneath the outer layer of metallization 34.
• the taper is for providing an impedance match between ferrite material of the section 56 and the dielectric material of the loading strip 33.
• a generally rectangular slab 58 of lossy absorber material is affixed to one side 60 of the ferrite section 56, the purpose of which will be considered subsequently.
• a second embodiment of an isolator device is shown in Figure 10 and differs from the embodiment shown in Figure 9 in the shape of the ferrite section included along the length of the dielectric strip 33. As shown, a section 56' of ferrite material having a sloped type of transition is included. These devices operate in the following manner.
• the electromagnetic field is distributed symmetrically with respect to the cross section of any transmission line.
• H dc a magnetic field
• the field crowds towards one edge of the line for a wave propagating in the guiding layer 32 in one direction, but crowds towards the other edge when the direction of propagation is reversed, as shown in Figures 11A and 11B.
• means for generating a magnetic biasing dc field H dc applies the magnetic field upwardly through the dielectric layers 30, 32 and 33 as well as the ferrite section 56 and 56' and the energy propagating along the line will be crowded towards the slab 58 of absorber material in one direction as shown in Figure 11 while being crowded toward the opposite edge for energy propagation in the opposite direction.
• energy will be absorbed strongly in one direction of propagation and not so for the reverse direction, thus providing a device which is commonly known as an isolator.
• Additional impedance matching and bandwidth requirements may be met by including a stepped or gradual discontinuity in the height of the ferrite sections 56 and 56' relative to that of the dielectric layer 33.
• this asymmetric field crowding can also be used to realize a device which is known as a circulator.
• a four port circulator configured in accordance with the subject invention is shown in
• FIG. 12 There a pair of ferrite strips 60 and 62 shaped in the form of back to back U-shaped elements and having outer layers of metallization 42 formed thereon are bonded to the upper surface of the dielectric layer 32 in relatively close proximity to each other.
• the ferrite strips 60 and 62 have a spacing P over a length L so that energy being guided beneath the strips 60 and 62 in the guiding layer 32 can be cross coupled.
• the ferrite strip 60 terminate in upper and lower port signals 2 and 3. It is a well known property that waves traveling in one direction along a transmission line excite waves which then travel in the opposite direction in another line which is electromagnetically coupled thereto. If a biased anisotropic material is used, e.g. ferrite, the asymmetric field distribution produces strong coupling between the lines for signal waves traveling in only one direction.
• a biased anisotropic material e.g. ferrite
• a circulator which will act as a clockwise "turnstile" for energy.
• H dc By reversing the polarity of H dc , the direction of circulation is reversed.
• a modulator results while an abrupt reversal, on the other hand, results in a switch.
• the isolator embodiment ( Figures 9 and 10) can with reversal of the biasing magnetic field become a single-pole, single-throw switch while the circulator ( Figure 12) becomes a double-pole, double-throw switch.
• FIG. 15 where there is disclosed an embodiment of an optically triggered switch device including a semiconductor member as one of the elements between the outer metallized layers 31 and 42.
• a loading strip 66 of semiconductor material having dimensions similar to the dielectric loading strip 33 ( Figure 4), is affixed to the top surface of the dielectric guiding layer 32; however, a gap 68 is formed in the top metallization 42 where light from a triggered light source, not shown, is directed to and impinges on the semiconductor strip at 69.
• Adjacent the semiconductor loading strip 66 is a relatively short arm member 70 of semiconductor material also coated with a top layer of metallization 42 and which is terminated at one end in a lossy load member 72.
• the arm is placed adjacent the semiconductor member 66 on the input side of the gap 68 and the dielectric or the semiconductor strip 66 is selected such that without the metallization 42, the waveguide is effectively open, i.e..
• the gap 68 when the gap 68 is illuminated with light of suitable wavelength, a hole-electron plasma is generated at the upper surface 69 which bridges the gap in the metallization 42 so that the waveguide is no longer cut off and transmits millimeter wave energy in the underlying guiding layer 32 past the gap 68.
• input millimeter wave energy is reflected back from the gap 68 where it is coupled to the side arm region including member 70 into the load 72 while the waveguide is in a cut-off state.
• the semiconductor material of the strip 66 can be chosen so that there is no cut-off condition. In this instance, the intensity and/or position of the optical beam directed to the gap 68 is used to modulate the amplitude and/or the phase of the millimeter wave propagating beneath the strip 66.
• semiconductor materials can be used as one or more of the layers. This even includes the guiding layer 32. Furthermore, active and passive sources can be integrated into the semiconductor.
• the waveguide structure of the subject invention thus has the potential for realizing complete circuit and system functions on a single semiconductor wafer; in other words, it is compatible with monolithic integration.
• transitions and/or coupling devices are required for the type of transmission line structure shown in Figure 4.
• One such transition is shown in Figure 16 and comprises a transition for use in connection with rectangular waveguide.
• the flange 74 of a piece of rectangular waveguide 76 includes a rectangular aperture 78 which is adapted to receive an outwardly projecting tapered wedge section 80 of the guiding layer 32 which is designed to fit into the interior of the waveguide 76 via the aperture 78.
• the principle of energy coupling is furthermore illustrated in Figure 17.
• FIG 18 shown schematically is the propagation of energy in the rectangular waveguide 76 for the dominant mode.
• the positioning of the wedge 80 of the dielectric layer 32 is shown inserted therein such that its plane is parallel to the broadwalls of the rectangular waveguide 76.
• the energy propagating in the waveguide 17 will be launched onto the guiding layer 32 through the tapered section 80 since the dominant mode in rectangular waveguide may be considered as a superposition of two criss-crossing rays 82.
• the angles between these two rays change with the frequency and accordingly, the useful operating bandwidth defines a range of angles ⁇ i .
• ⁇ For a given dielectric, one can therefore define ⁇ such that coupling will occur for the smallest value of ⁇ i .
• FIGS. 19 and 20 shown thereat are two embodiments for coupling power between external power source devices and the waveguide structure of the present invention.
• an active device 90 is shown mounted on one side surface of a heat sink 92.
• Coupling to the dielectric guiding layer 32 is provided by a right-angled strip of metallization 94 which is formed over the face 96 of the active device 90 and onto the top surface of the guiding layer 32 where it extends under the bottom face of the loading strip 33 and terminating in a taper 98 as shown to provide an impedance match.
• a bias filter comprised of a pattern 100 of metallization consisting of wide and narrow sections of metal are formed on the top surface of the guiding layer 32 which are coupled by conductor line segments 101 between the tip 98 of the metallization strip 94 and a bias voltage source, not shown, coupled to terminal 102.
• the tapered strip of metallization 94 in effect acts like a metallized antenna which is inserted into the interface 31 between the guiding layer 32 and the loading strip 33 to radiate energy into the guiding layer 32 of the transmission line in the desired mode.
• the metallization strip 94 can be configured into a grating to maximize coupling to the device.
• the embodiment shown in Figure 20, is illustrative of a configuration wherein the active device 90 is mounted on the top surface of the heat sink 92, in which case coupling between the active device 90 and the guiding dielectric layer 32 is made simply by a planar section of metallization 94' extending from the top surface 104 of the active device to the interface 31 between the dielectric layers 32 and 33.
• the metallization 94' additionally includes a tapered end section 106 over the active device 90.
• the bias filter 100 is again coupled between bias voltage terminal 102 and the tapered tip 98 of the metallisation strip 94' by segments of a line conductor 101.

Landscapes

• Waveguides (AREA)
• Waveguide Aerials (AREA)

Applications Claiming Priority (12)

Publications (2)

Family

ID=27560230

Family Applications (1)

Country Status (3)

Cited By (1)

Families Citing this family (9)

Family Cites Families (16)

• 1985
  • 1985-12-12 WO PCT/US1985/002423 patent/WO1986003891A2/en not_active Ceased
  • 1985-12-12 DE DE19853587607 patent/DE3587607T2/de not_active Expired - Fee Related
  • 1985-12-12 EP EP86900457A patent/EP0205570B1/de not_active Expired - Lifetime

Non-Patent Citations (1)

Also Published As

Similar Documents

Legal Events