EP0897544A1 - Concept de systeme radar/sonar pour couverture distance/doppler etendue - Google Patents

Concept de systeme radar/sonar pour couverture distance/doppler etendue

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Publication number
EP0897544A1
EP0897544A1 EP97927606A EP97927606A EP0897544A1 EP 0897544 A1 EP0897544 A1 EP 0897544A1 EP 97927606 A EP97927606 A EP 97927606A EP 97927606 A EP97927606 A EP 97927606A EP 0897544 A1 EP0897544 A1 EP 0897544A1
Authority
EP
European Patent Office
Prior art keywords
pulses
accordance
subpulses
pulse
doppler
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Withdrawn
Application number
EP97927606A
Other languages
German (de)
English (en)
Other versions
EP0897544A4 (fr
Inventor
Grealie A. Andrews, Jr.
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Individual
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Individual
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Filing date
Publication date
Priority claimed from US08/796,749 external-priority patent/US5808580A/en
Application filed by Individual filed Critical Individual
Publication of EP0897544A1 publication Critical patent/EP0897544A1/fr
Publication of EP0897544A4 publication Critical patent/EP0897544A4/fr
Withdrawn legal-status Critical Current

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Classifications

    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S13/00—Systems using the reflection or reradiation of radio waves, e.g. radar systems; Analogous systems using reflection or reradiation of waves whose nature or wavelength is irrelevant or unspecified
    • G01S13/02—Systems using reflection of radio waves, e.g. primary radar systems; Analogous systems
    • G01S13/50—Systems of measurement based on relative movement of target
    • G01S13/58—Velocity or trajectory determination systems; Sense-of-movement determination systems
    • G01S13/581—Velocity or trajectory determination systems; Sense-of-movement determination systems using transmission of interrupted pulse modulated waves and based upon the Doppler effect resulting from movement of targets
    • G01S13/582—Velocity or trajectory determination systems; Sense-of-movement determination systems using transmission of interrupted pulse modulated waves and based upon the Doppler effect resulting from movement of targets adapted for simultaneous range and velocity measurements
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S13/00—Systems using the reflection or reradiation of radio waves, e.g. radar systems; Analogous systems using reflection or reradiation of waves whose nature or wavelength is irrelevant or unspecified
    • G01S13/88—Radar or analogous systems specially adapted for specific applications
    • G01S13/89—Radar or analogous systems specially adapted for specific applications for mapping or imaging
    • G01S13/90—Radar or analogous systems specially adapted for specific applications for mapping or imaging using synthetic aperture techniques, e.g. synthetic aperture radar [SAR] techniques
    • G01S13/9021—SAR image post-processing techniques
    • G01S13/9029—SAR image post-processing techniques specially adapted for moving target detection within a single SAR image or within multiple SAR images taken at the same time
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S13/00—Systems using the reflection or reradiation of radio waves, e.g. radar systems; Analogous systems using reflection or reradiation of waves whose nature or wavelength is irrelevant or unspecified
    • G01S13/02—Systems using reflection of radio waves, e.g. primary radar systems; Analogous systems
    • G01S13/06—Systems determining position data of a target
    • G01S13/08—Systems for measuring distance only
    • G01S13/10—Systems for measuring distance only using transmission of interrupted, pulse modulated waves
    • G01S13/20—Systems for measuring distance only using transmission of interrupted, pulse modulated waves whereby multiple time-around echoes are used or eliminated
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S13/00—Systems using the reflection or reradiation of radio waves, e.g. radar systems; Analogous systems using reflection or reradiation of waves whose nature or wavelength is irrelevant or unspecified
    • G01S13/02—Systems using reflection of radio waves, e.g. primary radar systems; Analogous systems
    • G01S13/06—Systems determining position data of a target
    • G01S13/08—Systems for measuring distance only
    • G01S13/10—Systems for measuring distance only using transmission of interrupted, pulse modulated waves
    • G01S13/22—Systems for measuring distance only using transmission of interrupted, pulse modulated waves using irregular pulse repetition frequency
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S13/00—Systems using the reflection or reradiation of radio waves, e.g. radar systems; Analogous systems using reflection or reradiation of waves whose nature or wavelength is irrelevant or unspecified
    • G01S13/02—Systems using reflection of radio waves, e.g. primary radar systems; Analogous systems
    • G01S13/06—Systems determining position data of a target
    • G01S13/08—Systems for measuring distance only
    • G01S13/10—Systems for measuring distance only using transmission of interrupted, pulse modulated waves
    • G01S13/24—Systems for measuring distance only using transmission of interrupted, pulse modulated waves using frequency agility of carrier wave
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S15/00—Systems using the reflection or reradiation of acoustic waves, e.g. sonar systems
    • G01S15/02—Systems using the reflection or reradiation of acoustic waves, e.g. sonar systems using reflection of acoustic waves
    • G01S15/50—Systems of measurement, based on relative movement of the target
    • G01S15/58—Velocity or trajectory determination systems; Sense-of-movement determination systems
    • G01S15/582—Velocity or trajectory determination systems; Sense-of-movement determination systems using transmission of interrupted pulse-modulated waves and based upon the Doppler effect resulting from movement of targets
    • G01S15/584—Velocity or trajectory determination systems; Sense-of-movement determination systems using transmission of interrupted pulse-modulated waves and based upon the Doppler effect resulting from movement of targets with measures taken for suppressing velocity ambiguities, i.e. anti-aliasing
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S15/00—Systems using the reflection or reradiation of acoustic waves, e.g. sonar systems
    • G01S15/88—Sonar systems specially adapted for specific applications
    • G01S15/89—Sonar systems specially adapted for specific applications for mapping or imaging
    • G01S15/8902—Side-looking sonar
    • G01S15/8904—Side-looking sonar using synthetic aperture techniques

Definitions

  • This invention is a waveform/signal-processing concept that eliminates the range and Doppler ambiguities of a radar or sonar system by a factor related to the time-bandwidth product of the transmitted waveform.
  • the measurement of the range to a target is accomplished by measuring the time delay between each transmitted pulse and the arrival of echoes from the target.
  • Velocity measurement is computed from the change in phase of the echoes from pulse to pulse, i.e. the target's Doppler shift.
  • Ambiguities result in one or both of these measurements depending on the pulse repetition frequency (PRF) of the radar or sonar.
  • PRF pulse repetition frequency
  • the present invention can be applied to both radar and sonar tracking systems, for ease of explanation, the present invention will be described with respect to a radar system.
  • a fundamental decision in the design of radar systems is the selection of the PRF or its reciprocal, the pulse repetition interval (PRI) .
  • This decision will affect range and/or Doppler ambiguities which in turn will affect such capabilities as (a) the radar location and tracking of targets, (b) the necessary clutter rejection for search and tracking radars, and (c) the cross-range resolution and/or swath width of synthetic aperture radars (SAR) .
  • the selection of the PRF would classify the system as being a low PRF radar system, a medium PRF system or a high PRF system.
  • a low PRF radar is defined as a radar with the PRF low enough that the first range ambiguity is greater than the maximum anticipated target detection range, thereby resulting in no range ambiguities.
  • a low PRF is generally selected for long range search applications which are most concerned with maximum volume surveillance.
  • Tracking is a secondary capability that is usually accomplished with scan-to-scan measurements of the target location. Velocity is not measured directly but may be computed from a change in the target location from scan to scan. Target tracking is limited in a dense target scenario or highly maneuvering targets because of the relatively slow measurements of velocity and poor angular resolution.
  • Moving target indicators (MTI) and coherent integration are usually employed for moving target detection and clutter rejection.
  • a high PRF radar is defined as a radar with the PRF high enough that the first Doppler ambiguity is greater than the Doppler shift of the maximum anticipated target velocity, thereby producing no Doppler/velocity ambiguities.
  • a high PRF is generally selected for applications such as airborne intercept radars and shorter range tracking and weapon control radars that are most concerned with target velocity and high speed maneuvering targets.
  • Target range is usually computed after detection using staggered PRFs and algorithms such as "the Chinese remainder theorem”.
  • Target tracking is limited for these radars in a dense target scenario or highly maneuvering targets because of the limitations of the algorithms for resolving target range ambiguities.
  • the velocity resolution, the target signal to noise ratio (SNR) , and clutter rejection are also indirectly affected by the need to resolve range ambiguities.
  • the algorithms for resolving range ambiguities require several target detections at different PRFs during an antenna dwell. This means that all the returns cannot be coherently integrated to give the maximum velocity resolution, maximum SNR, or maximum clutter rejection.
  • An additional concern of these radars is that the clutter rejection needs are increased. Because of range ambiguities, the clutter folds over in range, increasing the clutter level in each range cell ⁇ * and also causing close-in, high-level clutter to interfere with the detection of long-range, low-level targets.
  • a medium PRF radar is defined as a radar with the PRF not high enough for the first Doppler ambiguity to be greater than the Doppler shift of the maximum anticipated target velocity, thereby resulting in Doppler/velocity ambiguities.
  • the PRF is not low enough for the first range ambiguity to be greater than the maximum anticipated target detection range, thereby resulting in range ambiguities.
  • both range and Doppler ambiguities must be resolved. Since there are fewer range ambiguities than with a high PRF and fewer Doppler ambiguities than with a low PRF, the impact of each of these ambiguities and the complexity of resolving them are reduced.
  • An imaging synthetic aperture radar is a unique application that cannot tolerate either range or Doppler ambiguities.
  • the Doppler spectral spread across the antenna pattern due the platform velocity is determined by the values selected for those parameters.
  • the PRF must be at least two times the Doppler spread to prevent Doppler ambiguities in the image. Usually an even higher PRF is selected to prevent returns from the skirts of the antenna pattern from folding over into the image.
  • the resulting PRF determines the maximum range, or swath width, of the imaged area. This swath width is usually considerable less than desirable for the efficient and economical utilization of the radar and its platform.
  • U.S. Patent 4,746,922 issued to Prenat includes a transmitter circuit producing pulses at different repetition frequencies (PRFs) .
  • PRFs repetition frequencies
  • a receiver circuit would receive echo signals which are filtered so as to eliminate those signals due to fixed targets and then apply the remaining signals to a bank of frequency filters with the required phase corrections to compensate for the different PRFs. Therefore, an echo signal will only be produced from the frequency filter corresponding to its associated Doppler frequency. Because of the ambiguity in the measurement of the Doppler frequency, the tuned frequencies of these filters are all less than the minimum PRF of the transmitted pulses.
  • U.S. Patent 4,106,019 issued to Alexander et al describes a system for measuring unambiguous target range for high velocity targets.
  • Target range and Doppler frequency data from three sequential transmission dwells of radar returns are stored, each dwell having a different PRF.
  • a correlation unit is included which would insure that the velocity of the target satisfies a 17 path algorithm across three adjacent dwells in at least one of five range azimuth profiles or paths.
  • U.S. Patent 5,442,359 issued to Euhin illustrates a method of resolving Doppler frequency shift ambiguities modulated with a periodic waveform having a plurality of pulses with a period having an unequal interpulse interval (PRI) .
  • PRI interpulse interval
  • the present invention overcomes the deficiencies of the prior art by providing method of and operation for preventing range ambiguities and Doppler shift ambiguities from occurring in pulse Doppler radar systems within a selected maximum range and maximum Doppler shift.
  • a waveform is utilized containing a train of frequency coded pulses whenever each pulse is separated from its neighboring pulse by the PRI of the waveform.
  • Each of the pulses is made up of a burst of continuous wave (CW) (or alternatively phase coded)subpulses of varying frequency.
  • CW continuous wave
  • the fir ⁇ t embodiment described will be the simplest embodiment in which the sub-pulses are CW pulses. Later, an embodiment with the sub-pulses phase coded will be described.
  • Each of the pulses contains the same subpulse frequencies, but in varying order.
  • a number of delays as well as correlators are utilized with a Fast Fourier Transfer to provide the proper output.
  • FIG. 1 is a radar wave form with each pulse coded with different frequency codes
  • FIG. 2 is an illustration of one of the frequency coded pulses in the burst of FIG. 1;
  • FIG. 3 is a matched filter receiver for the burst waveform shown in FIGS, l and 2;
  • FIG. 4 is a functional diagram of an implementation of a frequency hopping code generator
  • FIG. 5 is a functional diagram of an implementation of a timing pulses generator
  • FIG. 6 is a functional diagram of an implementation of Doppler compensation for the receiver shown in FIG. 3 with the n tn correlation shown in detail;
  • FIG. 7 is a functional diagram of an implementation of potential Doppler processing for the Doppler Processor shown in Fig. 3.
  • FIG. 8 is a graph of a response of a filter matched to the 10 element Costas coded waveform specified by the sequence of Equation 17;
  • FIG. 9 is a graph of a response of a "mismatched" filter to the 10 element Costas coded waveform specified by the sequence of Equation 17, wherein the weighting function is Equation 19;
  • FIG. 10 is a functional diagram of an implementation of a phase-code sub-pulse modulator
  • FIG. 11 is a graph of a response of a filter matched to the 10 element Costas coded waveform specified by the sequence of Equation 22 with the sub-pulses coded with the phase codes specified by Equations 23 and 24; and
  • FIG. 12 is a graph of a response of a filter matched to a sequence of four 10 element Costas coded waveforms with the sub-pulses of each code coded with the phase codes specified by Equations 23 and 24.
  • the present invention would allow a practitioner to select the PRF in a manner to prevent Doppler ambiguities for the maximum velocity of interest.
  • the invention would also select the number of coded pulses to be used in the burst waveform so that the maximum range of interest is less than the first range ambiguity.
  • the present invention would utilize a wide bandwidth waveform to give good range resolution without prohibiting other similarly designed radars from using the same bandwidth as long as all the radars use different subsets of the available frequency hopping codes.
  • the present invention would allow the design of an SAR with an antenna aperture size, transmit frequency, PRF and platform velocity to obtain the desired image resolution and then select the number of coded pulses to be used in the burst waveform to achieve the desired swathwidth of coverage.
  • This waveform, a ⁇ shown in FIG. l. consists of a train of frequency coded pulses where T is the PRI.
  • the frequency codes, labeled C ⁇ to C N are different on each pulse up to the N th pulse.
  • Each of the codes C ⁇ to C N are separated from adjacent codes by a quiescent interval. They are then repeated in bursts of N pulses per burst.
  • Each of the N pulses are themselves made up of a burst of continuous wave (CW) subpulses as shown in FIG. 2 for M subpulses.
  • CW continuous wave
  • the pulsewidth of each subpulse is ⁇ p and the total pulse width of the pulse is M ⁇ p .
  • the frequencies are labeled f ⁇ n , f2n' f 3 n' etc. in the order of their appearance in the pulse. This labeling is not related to the particular value of that frequency component, i.e. the frequency, f ln , is not necessarily greater than f 2n , and the frequency, f2n' ⁇ s not necessarily greater than f 3n , etc.
  • the characteristics of this frequency coding are: (1) The subpulses are contiguous in time, i.e. no spaces between them. (2) The code of each pulse is a member of a set which contains the same frequency subpulses, only the order of appearance of each frequency is changed from pulse to pulse.
  • the codes have an ambiguity function (a two dimensional auto-correlation in time delay and Doppler shift) with a single peak at zero time delay and zero Doppler shift and with low sidelobes approaching 1/M for all other values outside the mainlobe.
  • the Costas frequency hopped codes as described in "A Study of a Class of Detection Waveforms Having Nearly Ideal Range-Doppler Ambiguity Properties", appearing in Proceedings of the IEEE, Vol. 72, No. 8, (August 1984) authored by John P. Costas, meet these requirements.
  • the codes have a cross-ambiguity function (a two dimensional cross-correlation in time delay and Doppler shift) with no large peaks at any time delay or Doppler shift and with the peak values approaching 2/M.
  • the receiver for the present invention may be colocated with the transmitter or located at a location remote from said transmitter.
  • a key component of this concept is that the receiver is implemented as a matched-filter for the entire N-pulse burst.
  • a functional diagram of such a receiver 10 is shown in FIG. 3.
  • the key components of this receiver are: (1) analog delay lines or digital storage devices 12, 14, 16 and 18, (2) correlators 20, 22, 24, 26, 28, and (3) a Doppler Processor 30 which in this figure is implemented with a properly weighted Discrete Fourier Transform (DFT) or an FFT.
  • DFT Discrete Fourier Transform
  • FFT Fast Fourier Transform
  • the correlations prior to the storage or delays. It is also important to note that one less delay line or digital storage device is employed than the number of pulses N. Additionally, the same number of correlators as N pulses are included.
  • the delays are analog delay lines whose delay time is equal to the PRI, T.
  • the delays are digital storage (memory or shift registers) with a memory location for each range cell and the number of range cells is also determined by the PRI, T.
  • the number of range cells is MT/ ⁇ p , where M is the code length in FIG. 2, T is the PRI, and ⁇ p is the sub-pulse width in FIG. 2.
  • the correlators of Figure 3 would be implemented digitally for example with FFTs or as a stretch processor. It may be matched to maximize signal-to-noise or signal-to— clutter depending on the application.
  • An implementation of a Doppler compensated correlator is described later.
  • the function of these correlators is to correlate the returns from targets, labeled C ⁇ to C' N , with time delayed and Doppler shifted replicas, labeled R ⁇ to R N , of the codes selected for the radar.
  • the number of these correlators is equal to the number of PRIs over which it is desired to have no range ambiguities.
  • processor 3 can be any processor that is designed to separate or reject target returns based on the doppler shift of the signals. Many of these processors can be implemented as FFTs with the input signal samples properly weighted. Specific radar Doppler processors currently in use include: (1) the airborne MTI
  • the number of pulses processed in the Doppler processor, N j may be more or less than the number of codes,
  • N used for preventing range ambiguities.
  • N j will be much greater than N.
  • outputs from the correlators will be accumulated until N j are gathered.
  • the key factor in this invention is that the phase shift from pulse to pulse is related to the Doppler shift of the target returns as it is with a radar waveform whose pulses remains unchanged from pulse to pulse. Therefore, the coding used in this invention does not destroy or obscure these phase shifts.
  • the Doppler Processor 30 would include weighting for clutter process or filter side lobe control, moving target indication, main beam clutter filter and zero Doppler filter, as well as Doppler compensation.
  • the matched filter receiver of FIG.3 is shown at the point in time when the entire burst waveform has been received from a target and a match occurs.
  • the operation of the matched filter receiver as the returns from each code is received can be described by noting that, when the waveform of FIG. 1 is transmitted and strikes a target, the signals reflected back to the radar are characterized by the same codes in the same order and separated by the same PRI, T.
  • ⁇ the round trip propagation time to the target
  • the code, C - is received first at the input of the matched filter receiver of FIG. 3.
  • C'- ⁇ is the received time delayed and Doppler shifted return from C 1 .
  • this receiver is implemented digitally, the received signals are first sent through a quadrature detector and an analog-to-digital converter (A/D) .
  • a functional description of a digital implementation is presented herein to facilitate understanding of the invention.
  • the digitized C ⁇ is put into the storage device 12 and saved for a time, T. It is also sent to the correlator 20 where a cross-correlation of C ⁇ and R N occurs. R N is a time delayed and Doppler shifted replica of C N . Since the cross- -correlation of these two codes is minimal, property (5) above, only a small signal with a level of about 2/M will be sent to the Doppler Processor 30.
  • T later C' j ⁇ is shifted to device 14 and also sent to the correlator 22 where a crosscorrelation of C'- ⁇ and R ⁇ - ! occurs.
  • the code, C' 2 / is received from the target.
  • the digitized C' 2 is put into the storage device labeled 12 and saved for a time, T. It is also sent to the correlator 20 where a cross-correlation of C' 2 and R N occurs. Since the cross-correlation of both these two sets of codes are minimal, property (5) above, only small signals with a level of about 2/M will again be sent to the Doppler Processor 30.
  • This process is repeated again after another time interval, T, later when the code, C' 3 , is received from the target.
  • C' 2 and C ⁇ are shifted to storage device 14 and storage device 16 respectively.
  • the digitized C' 3 is put into the storage device 12 and saved for a time, T. It is also sent to the correlator 20 where a cross-correlation of C' 3 and R N occurs.
  • C' 2 is sent to the correlator 22 where a cross-correlation of C' 2 and RN- J . occurs.
  • C ⁇ is sent to the correlator 24 where a cross-correlation of C f 1 and R ⁇ - 2 occurs. All these cross-correlations are minimal so that minimal signals with levels about 2/M are sent to the Doppler Processor 30.
  • C' N is received from the target.
  • C' N _ 1 , C' N _ 2 , to C ⁇ are shifted to the next storage devices 14 to 18 respectively.
  • the digitized C' N is put into the storage device 12 and saved for a time, T. It is also sent to the correlator 20 where an auto-correlation with R N occurs; C 'N-1 *- S sent to the correlator 22 where a auto-correlation with R N-1 occurs; etc.
  • C'- ⁇ is sent to the correlator 28 where a auto-correlation with R-L occurs. The peaks of all of these auto-correlations are maximum so that maximum signals are sent to the Doppler Processor 30.
  • the peaks occur at a time, t+(N-l)T, after the beginning of transmission of the burst of N codes.
  • c is the velocity of propagation.
  • the nucleus of the implementation of FIG.4 is two stable, coherent oscillators denoted as stable local oscillator 32 and stable code oscillator 34.
  • the frequencies of the output sinusoidal signals are f 0 and l/ ⁇ p .
  • f 0 is some convenient frequency such as the radar's local oscillator frequency.
  • l/ ⁇ p is the frequency separation of the subpulses of the code, property (3) above.
  • the specific code can then be generated by gating on each of these frequencies one at a time at the proper time. This is accomplished by gates 44, 46, 48, 50 and 52. Gate 44 is directly connected to the stable local oscillator 32. Gates 46, 48, 50 and 52 are connected to the outputs of respective mixers 36, 38, 40 and 42. Each of these gates are turned on one at a time as determined by the timing pulses labeled V ⁇ through P M . These timing pulses are generated by timing pulses generator 54 when the desired code is selected. The width of these timing pulses is ⁇ p . Each of the resulting subpulses are combined to form the radar pulse with the desired code. The subpulses are combined in a summing device 56.
  • FIG. 5 A functional diagram of an example of an implementation of the timing pulses generator is shown in FIG. 5, the components of which are currently available as digital hardware.
  • An economical design would be to compute all the existing codes of the desired code length and store them for use when a subset is selected for radar operation. This would allow the subset to be changed if desired, for example when multiple radars are operating in near proximity.
  • a subset, N, of the N c code sequences is selected for the radar operation and is stored in a separate memory location 60.
  • N the number of the N code sequences is selected using the same PRI trigger that is used to trigger a transmitter.
  • This code sequence is placed into register 62.
  • the PRI trigger generator 66 is a part of the usual radar timing circuitry.
  • the output timing pulses, P m result from reading the sequence in register 62 one element at a time in order at each time interval, ⁇ p , as shown in reference 64 of FIG. 5.
  • the signal for 64 to advance to the next element of the code sequence is generated by a counter 68.
  • the counter 68 advances each time interval, ⁇ p , using a trigger generated by a threshold detector 70 with the output from the stable code oscillator 34 as its input.
  • the threshold detector 70 generates the desired trigger to be counted each time interval, ⁇ p .
  • the counter 68 is reset and the next pulse with a different code is generated. After all N code sequences have been used, the first sequence is selected again and the cycle repeats.
  • the transmitted signal is a set of N orthogonal codes such as Costas codes transmitted sequentially with a pulse repetition interval, T, as shown in FIG.1.
  • an orthogonal code waveform is defined as a waveform (a) whose normalized cross-correlation with a time-delayed, Doppler-shifted replica has a single peak of M at zero time delay and zero Doppler shift, and in regions outside the vicinity of this peak has a nominal normalized level of one and (b) whose normalized cross-correlation with different codes have no peaks.
  • the n*-* 1 code of this sequence is illustrated in FIG. 2.
  • the transmitted sequence can be expressed mathematically,
  • Tf ( 2 ) which has shifted the local oscillator frequency, f 0 , to the transmit frequency, f ⁇ .
  • l mn is an integer between 1 and M that represents the difference frequency for the m th sub-pulse of the n ⁇ n code or pulse.
  • the parameter, l mn f defines the code.
  • the waveform described by Equation (1) is transmitted, it strikes a target at range, R, that is traveling with a velocity whose radial component is V R .
  • XRO X ⁇ ( ⁇ ⁇ * ⁇ T > f mn ⁇ fmn ⁇ dmn
  • d mn is the Doppler shift of the m tn frequency component of the n rn code, or pulse, i.e. mn c TM.
  • the received signal is N M / ⁇
  • Such a matched filter has an impulse response that is the complex conjugate of received waveform of Equation (3) .
  • the transmit frequency is removed in the front end of the usual radar receiver using quadrature (I,Q) detection and the resulting baseband signal is sampled and digitized using analog-to-digital (A/D) converters.
  • quadrature detection is represented by substituting Equation
  • the exponential in front of the summations is simply the phase shift of the transmit frequency due to the propagation time out to the target and back. It does not affect the structure of the receiver nor the magnitude of the receiver output. Therefore it can be dropped from further consideration.
  • FIG. 6 A functional diagram of a correlator that is matched to the resulting received waveform of Equation (4) is illustrated in FIG. 6. Shown is a method of compensation for the variation of Doppler shifts across the bandwidth of the waveform and compensation for the phase "noise" caused by shifting the frequency components back and forth from pulse to pulse.
  • the components labeled Delayl through DelayN-1 and Correlator1 through CorrelatorN are the same as in FIG. 3 and are repeated here only for clarity. The remainder of the figure is the details of Correlator n.
  • each of the N correlators are functionally identical.
  • Correlator n 80 is shown in detail and will described.
  • the M sub-pulses for the n* * " code are selected.
  • the M sub-pulses are labeled S Mn , . . . S mn , . . . S ln and are selected by selecting the locations in the memory (or storage devices labeled Delayl through DelayN-1) of the samples of the received signals that are received at a time, ⁇ p , apart.
  • ⁇ p is the sub-pulse width.
  • Each of the sub-pulses are fed to sub-pulse filters 94, 96, 98 and 100.
  • the function of each of these filters is to select the M samples of each of the M sub-pulses of the n th code and to perform the appropriate Doppler correction to each sample. Then by combining (summing) the M samples, a matched filter is formed that is matched to the time-delayed and Doppler-shifted return from each sub-pulse for Doppler shifts of 0, 1/NT, 2/NT, . . . (N-1)/NT.
  • the sub-pulse filter 102 for the ⁇ r"" sub-pulse of the n*-* 1 code is shown in detail.
  • the SS- set of sub-pulse filters which are the components labeled
  • Sub-pulse Filter 1 through Sub-pulse Filter M in FIG. 6 form a part of Correlator n 80 and such a set form a part of each of the N correlators.
  • the M samples are labeled X Mmn , X (M - 1)lnn , . . . X imn , . . . X lmn and are selected by selecting the locations in the memory (or storage devices 86, 88, 90, 92 of the samples of the received signals that are received at a time, ⁇ p /M, apart.
  • ⁇ p /M is the sampling period (the reciprocal of the sampling frequency) .
  • each of these "delay lines” represent a delay of ⁇ p /M and form a part of Sub-Pulse Filter m which forms a part of Correlator n 80, and such a set form a part of each of the M sub-pulse filters of each of the N correlators.
  • the required Doppler corrections are applied to each of these samples in the components 112, 114, 116 and 118.
  • These sets of components form a part of Sub-Pulse Filter m which forms a part of Correlator n 80 and such a set form a part of each of the M sub-pulse filters of each of the N correlators.
  • Doppler filter is formed for each of the potential Dopplers up to the maximum designed unambiguous Doppler shift, 1/T. The resolution of these filters and the separation between them is 1/NT. After this sub-pulse filtering, pulse filters and pulse-to-pulse Doppler filters along with any additional Doppler processing is accomplished in the Doppler Processor of FIG. 3.
  • the Doppler correction is accomplished by multiplying each sample (the i th sample being shown in detail) by N reference signals, one for each of the Doppler filters.
  • the N multipliers are labeled by 120, 122, 124, 126 and 128 in FIG. 6.
  • the underlined variables in FIG. 6 and the equations herein represent vectors, matrices, and tensors.
  • the ith reference vector for the i tn sample of the m tn sub-pulse of the n th code is
  • l mn is an integer from 1 to M that is specified by the particular code. It selects the frequency of the m*- h sub-pulse of the n*- n code.
  • the total reference tensor for the n*-* 1 code is
  • This vector makes up the output from the component labeled Doppler-Correction i,m,n in FIG.6 and is combined with similar vector outputs from the Doppler correction components labeled 112, 114, 116 and 118 to form a matrix of dimension MxN.
  • This matrix provides the input to the component labeled ⁇ i in FIG. 6.
  • the output from the component labeled ⁇ in FIG. 6 is again a vector of length N .
  • This vector makes up the output from the component labeled Sub-pulse Filter m in FIG. 6 and is combined with similar vector outputs from the sub-pulse filters labeled Sub-pulse Filter 1 through Sub-pulse Filter M to form a matrix of dimension MxN.
  • This matrix forms the output of the component labeled Correlator n in FIG. 6.
  • This output is the result of correlating the input signal with time-delayed and Doppler shifted replicas of the n th code. corrections have been made for variation of the Doppler shifts over the bandwidth and the W 9
  • phase "noise” caused by moving each frequency component forward and backwards in time from code to code i.e. pulse to pulse
  • N corrections were made for each sample of the input signal, one for each of the Doppler filters that will be formed in the Doppler Processor of FIG. 3. All the effects of the pulse to pulse changes in the code has been removed and remaining Doppler processing can be conventional except that weighting to control the shape of the filters is done in the frequency domain instead of the time domain as is more common. It should be emphasized again that these operations, after the Doppler corrections are made to each sample, may be performed in any order. Referring to Equation (10) , it is seen that, from the characteristics of the data, the order of the summations over i, m, and n is arbitrary.
  • This is the input to the Doppler Processor illustrated functionally in FIG. 7, where the matched filtering of each pulse or code is completed along with the Doppler filtering. This is analogous to the pulse compression function of the conventional matched filter for linear frequency modulated signals or other coded waveforms.
  • the matched filter or pulse compression of each pulse is completed by simply coherently summing the outputs from each of the M sub-pulse filters of each of the N Doppler compensated correlators. This is accomplished in the component labeled ⁇ m in FIG.7.
  • the tensor S" k ⁇ nn is reduced to a data matrix S 1 '' ⁇ at this point and each pulse has been decoded and compressed in range.
  • the range resolution is c ⁇ p /2M where ⁇ p is the sub-pulse width and M is the code length.
  • Doppler filters are formed which include all the N pulses. This provides N Doppler frequency outputs as shown in FIG. 7 and labeled O ⁇ , D 2 , D 3 , . . . D k , . . . D N .
  • the outputs from the Doppler Filters of FIG. 7 can be further processed for more Doppler resolution by simply collecting these outputs until the desired number, such as N j , is obtained.
  • the succeeding Doppler processing would be conventional Doppler processors such as MTI, FFT, etc. as shown in FIG. 7 since the effects of the pulse to pulse coding have been removed at this point.
  • the processing simply consists of coherently summing the data described by Equation (10) .
  • Equation (10) if d mn equals then the summation over the M samples of the M sub-pulses of the N pulses gives a normalized amplitude of M 2 N as a matched filter should.
  • the "window" function is applied to the pulse compression filter by applying weights representing the chosen “window” function to the sub-pulses of each of the codes which are the inputs to the component labled ⁇ jj , in FIG. 7.
  • the weights are applied in an order determined by the magnitude of the frequency of the sub-pulse and not in the order determined by the time of the sub-pulse. This is usually referred to as "frequency” weighting and is unique to this invention in that the sequence of weighting in time is determined by the specific code of that pulse which, in turn, determines the order of the appearance of the frequencies in time. For example, if the n th pulse is a 10 element Costas code described by the sequence
  • l mn [l 248 5 10973 6] (17) which results in the sub-pulses of the coded pulse having frequencies as determined by Equation (2) .
  • the matched filter (or autocorrelation) response to this waveform is shown in FIG. 8 from the time the code first begins to enter the filter, labeled -10 sub-pulses on the time axis, until the time that it is completely in the filter, labeled 0 sub-pulses on the time axis. Continuing along the time axis, which is not plotted, would be the mirror image of this curve as the waveform exits the matched filter. Therefore, there is only one peak and that peak occurs when the time delay between the waveform and the response of the matched filter is 0.
  • the sidelobe region is the region outside the single peak which is referred to as the mainlobe.
  • the width of the mainlobe is given by the total pulsewidth, M ⁇ p , divided by the square of the code length, M 2 .
  • This narrow mainlobe response results in the expression "pulse compression”. Therefore, the "compressed" pulsewidth is ⁇ p /M, or ⁇ p /10 in this example.
  • the sidelobe region in this example extends from -10 ⁇ p to -0.l ⁇ p .
  • the nominal sidelobe level for the codes is determined by the length of the code, M. Although only one code is represented in FIG. 8, a sequence of N codes are used with this invention resulting in the sidelobes being reduced by another factor of 1/N. It is seen from FIG.8 that the level of the sidelobes relative to the peak of the mainlobe is nominally 0.1 as determined by the code length of 10. However it is a characteristic of these codes that when the time delay between the waveform and the response of the filter is less than one sub-pulse width (-1 to 0 in FIG.
  • the filter response is a sinc(x) function which results in a peak sidelobe level of -13.7 dB regardless of the particular code of a set of Costas codes or regardless of the length of the code.
  • the control of the sidelobes in this region can be accomplished in two ways. The first requires the application of a "window" function to the sub-pulses and the second applies a modulation to the CW sub-pulses.
  • the window function reduces the sidelobes in this region, but a small signal-to-noise ratio (S/N) loss results since the filter becomes slightly mismatched.
  • S/N signal-to-noise ratio
  • a similar "window" function is applied to the input signals from pulse to pulse in a conventional way to control the sidelobes of the Doppler filters.
  • the Doppler filters are formed in the component labeled ⁇ n in FIG. 7 which include all the N pulses. This provides N Doppler frequency outputs as shown in FIG. 7 and labeled D l r D 2 , D 3 , ...D k , ...D N .
  • Equation (5) is the weights applied to the N pulses of the waveform to control the Doppler sidelobes. These weights are combined with the reference signals of Equation (5) and applied to Doppler Correction components of FIG. 6 as before. With the weighting function included, the reference signals of Equation (5) become
  • the second method of reducing the sidelobes occuring in the region of the auto-correlation function with less than one sub-pulse width relative delay is to apply a modulation to the sub-pulses.
  • the modulation itself must have an autocorrelation function with a single peak at zero delay and low peak sidelobes.
  • Properly selected binary phase codes meet this criterion.
  • Advantages of phase coding the sub-pulses include: (1) It increases the time-bandwidth by a facter equal to the phase code length, M p , and therefore increases the range resolution of the radar by the same factor. (2) It reduces the sidelobes near the mainlobe without the use of a window function and the associated loss in S/N.
  • FIG. 10 A functional diagram of an implementation of a phase-code sub-pulse modulator is shown in FIG. 10.
  • a set of N p selected phase codes of length M p is stored in memory 132 in FIG. 10.
  • the selected frequency code,C n for each pulse is sent through the modulator 130 where the selected phase code for each sub-pulse modulates each sub-pulse of the frequency code.
  • the phase code for each sub-pulse is selected by the component 134 of FIG. 10.
  • the signal for 134 to advance to the next phase code is a trigger generated by a threshold detector 136 with the output from the stable code oscillator 34 as its input.
  • the threshold detector 134 generates the desired trigger to change phase codes each time interval, ⁇ p .
  • the bandwidth of the sub-pulses becomes M p / ⁇ p for a phase code length with M p elements. This means that the frequency separation between elements (sub-pulses) of the frequency hopping codes will also be M p / ⁇ p and the total bandwidth will be increased by a factor of M p resulting in an increase in range resolution by the factor of M p .
  • Barker codes are defined by
  • the matched filter (or autocorrelation) response to this waveform is shown in FIG. 11 from the time the code first begins to enter the filter, labeled -10 sub-pulses on the time axis, until the time that it i ⁇ completely in the filter, labeled 0 sub-pulses on the time axis. Continuing along the time axis is the mirror image of this curve as the waveform exits the matched filter. Therefore, there is only one peak and that peak occurs when the time delay between the waveform and the response of the matched filter is 0. Comparing FIG. 11 with FIG. 8, the peak sidelobe near the single peak has been reduced from -13.7dB to -21.9dB and the average sidelobes are reduced from -31.6dB to -40.5dB. The narrower peak, higher range resolution is also illustrated.
  • FIG. 12 The matched filter (or autocorrelation) response to a sequence of four such Costas and Barker coded waveforms is shown in FIG. 12. Comparing FIG. 12 with FIG. 11, the peak sidelobe near the single peak has not been reduced significantly since it is controlled by the autocorrelation of the 13 element Barker code. However, the average sidelobes are reduced from -40.5dB to -46.7dB. This illustrates the coherent summation of the peaks of the autocorrelation function of the sequence of codes and the incoherent summation of the sidelobes.
  • the performance of this invention is fundamentally tied to the time-bandwidth product of the waveform in several ways. Since the frequency separation between the subpulses is M p / ⁇ p when the sub-pulses have a binary phase code of length M p applied and there are M subpulses (i.e. the code length is M) , then the waveform's bandwidth is M p M/ ⁇ p . The width (i.e. time duration) of the subpulses is ⁇ p ; therefore M subpulses results in a total pulsewidth of M ⁇ p . The time-bandwidth product, then, is M p M 2 . In several ways the larger the time-bandwidth product, M p M 2 (or the longer the code length, M) , the greater the benefits of this waveform/signal-processing concept.
  • the energy level in the sidelobes of the auto-correlation function is approximately the reciprocal of the time-bandwidth product, l/M p M 2 .
  • the lower these sidelobes the easier to locate the peak and therefore to detect and locate the target. Additionally, it improves the ability to detect small targets in the vicinity of a large target.
  • the energy level in the peaks of the cross-correlation function is approximately two times the reciprocal of the time-bandwidth product, 2/M p M 2 .
  • the number of frequency codes in a set meeting the five properties described above is determined from the length of the code, M, which as shown before is the square root of the time-bandwidth product when no phase coding of the sub-pulses is employed.
  • M the length of the code
  • the existence of a large number of codes allows a large sub-set of these codes to be used in a burst with no range ambiguities and equally important allows multiple radars to operate in the same vicinity and in the same frequency band by using different sub-sets of the set of codes. In this example, if it is desired to have the first range ambiguity to be 10 times the interpulse period, then 10 codes would have to be used which would conceivably allow 20 radars to operate nearby with minimal interference.
  • MpM' 1 there are advantages to achieving this with more bandwidth and less time. It results in less eclipsing of targets whose returns arrive at the radar when the transmitter is on, and it also leads to greater range resolution which is important for tracking in a dense target scenario and for target classification and raid count; and for SAR applications, this results in higher resolution images.
  • this invention can be realized in a number of embodiments of which the disclosed embodiments are only several of many equivalent alternatives
  • a radar system for determining the range and velocity of one or more targets, comprising: a code device for producing at least one series of N radio frequency pulses directed at the targets, each of said pulses separated from adjacent pulses by a time T, each of said pulses containing a plurality of M contiguous subpulses, each of said M subpulses in each of said pulses exhibiting a different frequency than the remaining subpulses in that particular pulse, and further wherein the order of appearance of said M subpulses in each of said pulses is unique with respect to the remaining pulses in said series of N pulses; a transmitter connected to said code device for transmitting said at least one series of N pulses directed at the targets; and a receiver for receiving said at least one series of N pulses reflected from the targets, said receiver including a plurality of delay devices, each of said delay devices storing one of said N pulses for a time T; a plurality of correlators, each of said correlators connected to either an input of said receiver or one of said delay devices, each of said

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Abstract

Procédé et dispositif permettant d'empêcher la survenue d'ambiguïtés distance et d'ambiguïtés Doppler dans un environnement aussi bien radar que sonar. Une série de N impulsions est produite, dont chacune contient un certain nombre de sous-impulsions contagieuses. Chaque sous-impulsion a une fréquence différente des autres sous-impulsions au sein de cette impulsion particulière. De plus, l'ordre d'apparition des sous-impulsions dans chaque impulsion est unique par rapport aux autres impulsions de la série. Un récepteur à filtre (10) et une unité de traitement Doppler (30) adaptés sont utilisés pour fournir des autocorrélations et des corrélations croisées (R1-RN), ce qui permet d'éviter les ambiguïtés distance et les ambiguïtés Doppler.
EP97927606A 1996-05-08 1997-04-30 Concept de systeme radar/sonar pour couverture distance/doppler etendue Withdrawn EP0897544A4 (fr)

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US796749 1991-11-25
US1677996P 1996-05-08 1996-05-08
US16779P 1996-05-08
US08/796,749 US5808580A (en) 1997-02-06 1997-02-06 Radar/sonar system concept for extended range-doppler coverage
PCT/US1997/006807 WO1997042520A1 (fr) 1996-05-08 1997-04-30 Concept de systeme radar/sonar pour couverture distance/doppler etendue

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AU712338B2 (en) 1999-11-04
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JP2000509811A (ja) 2000-08-02
AU3203497A (en) 1997-11-26

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