Classifications

• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/26—Systems using multi-frequency codes
  • H04L27/2601—Multicarrier modulation systems
  • H04L27/2647—Arrangements specific to the receiver only
  • H04L27/2655—Synchronisation arrangements
  • H04L27/2656—Frame synchronisation, e.g. packet synchronisation, time division duplex [TDD] switching point detection or subframe synchronisation
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/26—Systems using multi-frequency codes
  • H04L27/2601—Multicarrier modulation systems
  • H04L27/2626—Arrangements specific to the transmitter only
  • H04L27/2627—Modulators
  • H04L27/264—Pulse-shaped multi-carrier, i.e. not using rectangular window
  • H04L27/26416—Filtering per subcarrier, e.g. filterbank multicarrier [FBMC]
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/26—Systems using multi-frequency codes
  • H04L27/2601—Multicarrier modulation systems
  • H04L27/2647—Arrangements specific to the receiver only
  • H04L27/2649—Demodulators
  • H04L27/26534—Pulse-shaped multi-carrier, i.e. not using rectangular window
  • H04L27/2654—Filtering per subcarrier, e.g. filterbank multicarrier [FBMC]
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L25/00—Baseband systems
  • H04L25/02—Details ; arrangements for supplying electrical power along data transmission lines
  • H04L25/03—Shaping networks in transmitter or receiver, e.g. adaptive shaping networks
  • H04L25/03006—Arrangements for removing intersymbol interference
  • H04L2025/0335—Arrangements for removing intersymbol interference characterised by the type of transmission
  • H04L2025/03375—Passband transmission
  • H04L2025/03414—Multicarrier
• • H—ELECTRICITY
  • H04—ELECTRIC COMMUNICATION TECHNIQUE
  • H04L—TRANSMISSION OF DIGITAL INFORMATION, e.g. TELEGRAPHIC COMMUNICATION
  • H04L27/00—Modulated-carrier systems
  • H04L27/26—Systems using multi-frequency codes
  • H04L27/2601—Multicarrier modulation systems
  • H04L27/2697—Multicarrier modulation systems in combination with other modulation techniques

Definitions

• Multicarrier Digital Transmission System Using an OQAM Tran-.multiplexer
• the present invention relates to systems for the transmission of digital data over a communication channel using a multicarrier modulation and, more particularly, to an improved OQAM transmultiplexer method for use in such systems.
• a multicarrier transmission system as opposed to a single carrier system, uses a set of different frequencies distributed in the transmission channel frequency band to carry the data.
• the main advantage is that the bit rate can be adjusted for each carrier, according to the noise and distortion power in the vicinity of this carrier.
• a better approximation of the theoretical information capacity limit can be expected and, particularly, poor quality channels can be exploited, like some wireless communication channels, power lines or the telephone subscriber lines at high or very high frequencies.
• OFDM Orthogonal Frequency Division Multiplexing
• DMT Digital Multi-Tone
• FFT Fast Fourier Transform
• OFDM/DMT suffers from a number of weaknesses, which, overall, make it perform hardly better than single carrier transmission: a complex time equalizer has to be introduced in front of the receiver to reduce the channel impulse response length, very precise time synchronization is necessary, a long initialization phase is required, and the carriers, and subchannels, are poorly separated, which reduces the capacity of the system in the presence of jammers. In fact, a good quality channel is necessary for that scheme to work properly.
• Ample documentation can be found in the literature and a good list of references is given in the book by W.Y.Chen.
• a second approach aims at overcoming some of the OFDM/DMT limitations through the use of more sophisticated transforms than the FFT, namely, lapped transforms and wavelet transforms.
• the idea is to improve the separation between carriers, or subchannels.
• S.D.Sandberg and M.A.Tzannes « Overlapped Discrete Multitone Modulation for High Speed Copper Wire Communications » IEEE-JSAC, Vol.13, N°9, Dec. 1995.
• the lapped and wavelet approaches still retain some of the crucial OFDM/DMT limitations and particularly the time synchronization requirements.
• OQAM Orthogonally Multiplexed QAM System Using the Discrete Fourier Transform » IEEE Trans, on Communications, Vol.COM-29, July 1981.
• the subchannel sampling rate is twice the Nyquist subchannel frequency, or subchannel spacing, and the data are transmitted alternatively on the real and the imaginary part of the complex signal in any subchannel, with, again, an alternation between two adjacent subchannels.
• intersymbol interference is eliminated in a subchannel and between adjacent subchannels.
• the distortions introduced by the transmission channel can be eliminated by a multibranch equalizer in each subchannel.
• a single branch equalizer can be used, see the paper by L.Qin and M.Bellanger, « Equalization issues in Multicarrier Transmission Using Filter Banks » Annals of Telecommunications, Vol.52, N°l-2, January 1997.
• a signal containing synchronization patterns which define a timing structure consisting of frames, superframes and hyperframes is fed, in the emitter, to the input of one or several subchannels reserved for synchronization, the relevant information being carried by the magnitude or envelop of the complex OQAM signal.
• the same signal also contains service data giving the number of bits allocated to each subchannel.
• a short fixed pattern is introduced periodically to serve as a reference signal for the subchannel equalizers in the receiver.
• the signal corresponding to the synchronization subchannel (s) is coupled to a first cascade containing an amplitude equalizer, an envelop detector and a filter that delivers the control signal for the phase lock loop associated with the receiver clock generator and to a second cascade containing an amplitude and phase equalizer, a data detector and a block for the identifica- tion of the superframe and hyperframe synchronization patterns and the service data extraction.
• the other outputs of the analysis filter bank are coupled to cascade subchannel equalizers consisting of three elements each, namely an amplitude equalizer, a phase equalizer and a fine equalizer.
• Every subchannel equalizer is followed by a data extractor and both use the information provided by the synchronization subchannel to complete their functions.
• the output error signals are used to determine the number of bits assigned to each subchannel and the information is transmitted to the distant terminal via the synchronization subchannel (s) every hyperframe.
• FIG. 1 is a simplified schematic block diagram of a multicarrier transmission system, in accordance with the present invention
• FIG. 2 is a schematic block diagram of the digital multicarrier emitter
• FIG. 3 is a drawing showing the waveform of the envelop of the signal in the synchronization subchannel
• FIG. 4 is a schematic block diagram of the digital multicarrier receiver
• FIG. 5 is a schematic block diagram of the subchannel equalizer
• FIG. 6 is a schematic block diagram showing the functions involved in the processing of the signals received in the synchronization channel.
• the block diagram of a multicarrier transmission system is shown in FIG. 1, for the case of a digital subscriber telephone line application.
• the input data stream d(n) is fed to a multicarrier emitter 100, which forms the signal Se(n) and is itself coupled to a module 10 that performs the digital-to-analog (D/A) conversion and the emitter analog front end functions.
• the emitter analog front end consists of an amplifier and a low-pass or pass-band filter to limit the spectrum sent to the hybrid circuit 11.
• the hybrid is connected to the twisted pair line 14 and its receiving port is connected to a module 12 that performs the receiver analog front end functions and the analog-to-digital (A/D) conversion.
• the receiver analog front end consists of a low-pass or band-pass filter to prevent aliasing and a variable gain amplifier. If symmetric, or full duplex transmission is contemplated, the A/D converter is coupled to an echo canceller 13 that ensures an adequate level of separation between the two directions of transmission. The echo canceller produces the signal Sr(n) and it is coupled to a multicarrier receiver 200 that delivers the output data stream d* (n) .
• a detailed description of the analog front ends, hybrid circuits, A/D and D/A converters as well as echo cancellers is given in W.Y.Chen's book.
• the present invention is concerned with the multicarrier emitter block 100 and receiver block 200 shown in greater details in FIGs. 2 and 4 respectively.
• the input data d(n) are processed by a cascade of three blocks, namely a serial/parallel converter 110, an OQAM modulator 120 and a synthesis filter bank (SFB) 130, to produce the emitted digital multicarrier signal Se(n) .
• the received multicarrier digital signal Sr(n) is processed by a cascade of 4 blocks, namely an analysis filter bank (AFB) 210, a subchannel equalizer 220, a data extractor 230 and a parallel/serial converter 240, to produce the output data sequence d'(n).
• d 1 (n) and d(n) are identical, except for a delay.
• the specificity of the filter banks, SFB 130 and AFB 210 resides in the values of their coefficients, that are the same for both, or very close.
• the filter bank coefficients are computed from a prototype filter frequency response H(f) that is half-Nyquist in the pass-band and provides the maximum attenuation in the stop-band. Therefore, the cascade of the filter banks SFB and AFB exhibits a frequency response H 2 (f) that satisfies the first Nyquist criterion. It is advantageous to have H 2 (f) satisfy also the second Nyquist criterion, because intermediate signal samples take on well defined values.
• the intermediate signal samples at the output of the SFB-AFB cascade are ⁇ +1; 0; -1 ⁇ .
• Ht(f) cos( ⁇ N f / 2 fs) ; 0 ⁇
• the specificity here is that the signal samples take on real and imaginary values alternatively to obey the OQAM principle and the number of levels is determined by an external control signal denoted « scdatar » in FIG. 2.
• the control signal scdatar adjusts the number of bits transmitted by a subchannel to its estimated capacity, as will be explained below. For example, if 1 bit can be carried by subchannel i, the sample xi may take on the following values: ⁇ 1. But, if 2 bits can be carried, then the sample xi may take on the values: -1.5; -0.5; +0.5; +1.5.
• the serial-to-parallel converter 110 splits the input bit stream d(n) into as many substreams as used subchannels and, for each substream, constitutes groups of bits di, under the control of the external signal scdatar, to feed the OQAM modulator 120.
• An additional external signal denoted « timing 1/64 » in FIG. 2, is used to insert a reference pattern in the signal sequence as will be explained below and it is fed to both blocks serial/parallel converter and OQAM modulator.
• At least one subchannel is used to carry a synchronization signal described hereafter and service data.
• the corresponding signal xis is generated by the « synchro+data » unit 150, the service data, denoted « scdatae » in FIG. 2, being supplied by the receiver 200 shown in FIG. 4.
• the specific synchronization signal is designed to provide an efficient and robust control of the sampling times in the receiver and perform frame, superframe and hyperframe alignment. It contains the following superframe synchronization pattern.
• the signal used to control the phase lock loop associated with the oscillator that delivers the received signal sampling frequency, or receiver clock generator, is obtained by filtering the 2 kHz component in v(n).
• This function is advantageously realized in two steps, as follows.
• the signal ca(4n) is used to control the phase lock loop of the clock generator.
• the sign ⁇ in P0 and Pi is used as shown in FIG. 3 to ensure that rising and falling edges of the 2 kHz signal keep a fixed relative position in time, regardless of the transmitted service data.
• an additional feature of the OQAM modulation block 120 in the emitter is that it imposes fixed values to the first two samples of the superframe in each subchannel, for example: ⁇ [ 1; 1 ] .
• a specific sign may be attributed to each subchannel, in order to avoid producing a large peak in the emitted multicarrier signal Se(n) at the beginning of each superframe.
• the multicarrier received signal Sr(n) is processed by a cascade of 4 blocks, namely the analysis filter bank 210, a subchannel equalizer 220, a data extraction module 230 and a parallel-to-serial converter 240.
• the subchannel equalizer 220 which consists of a cascade of 3 distinct equalizers, is shown in more details in FIG. 5.
• the amplitude equalizer 221 receives the input sequence xir(n) and a reference amplitude ra supplied by the bit assignment module 250.
• the amplitude equalizer 221 computes a variable gain gl (n) , multiplies the input signal by that gain and transfers the result yi (n) to the phase equalizer 222.
• gl(n+l) ra/sgm(n+l) (6)
• the gain is updated at the beginning of the superframe, using the first two samples, denoted yi (n) and yi(n+l) .
• the following matrix system is solved in the least squares sense.
• the error signal is used for noise level estimation as described below.
• the fine equalizer 223 computes the following output
• the function of the fine equalizer is to complete the task of the two previous modules, in particular to remove the residual distortion. Its coefficients h k (n) generally take very small values and they can be updated at the superframe rate, using the same reference signal as the phase equalizer 222. In addition, they can be updated during regular transmission, according to the data directed equalizer principle, using the error signal ei (n) provided by the data extractor 230 and the least mean squares
• the capacity of a subchannel is determined by the « capacity+ bit assign » block 250.
• This block receives the error signal eip from the phase equalizer and the error signal ei from the data extraction block 230. It computes the following two variables .
• the quantity El(p) is computed every superframe and it is representative of the total distortion plus noise power present in the subchannel, before fine equalization.
• the quantity E2 (n) is computed at the rate 8 kHz and it is representative of the noise power in the subchannel. In normal operation, with the above equations (13) and (14), E2 (n) is smaller than El(p) and the difference depends on the improvement brought by the fine equalizer.
• Nb is calculated, for example through successive comparisons to thresholds, as
• Nb Int [ Vi Log 2 ( 1/ El(p» - I ] ; El(p) ⁇ 0.25 (15)
• E2 (n) is used to confirm the decision or improve it. For example, if E2 (n) is smaller than El (p) /4, the number of bits may be increased by one.
• the number of bits assigned to the subchannels is limited by the service data capacity. As pointed out earlier and shown in FIG. 3, the synchronization signal can transmit 14 bits of service data in a superframe with the numerical values given. If 3 bits are allotted to each subchannel, 4 subchannels can be handled per superframe and, if 240 subchannels are actually used, then, 60 superframes are sufficient to transmit the whole capacity information. The number of bits assigned to any subchannel is included in the range [0, 7] .
• the data extraction module 230 receives the signal vi (n) from the subchannel equalizer 220 and performs a quantization operation on the real and imaginary parts alternatively, using the quantization scale associated with the number of bits assigned to the subchannel.
• the binary representation of the quantized value dir is fed to the parallel/serial converter 240 and the quantization error ei (n) is sent back to the subchannel equalizer 220 to be used as per equation (12) .
• the parallel/serial converter 240 produces the output data stream d' (n) .
• the «synchro processing » block 270 is shown in more details in FIG. 6. It receives the synchronization subchannel signal xisr and performs amplitude equalization through block 271, as described previously for the other subchannels.
• the control signal ca(4n) is generated by block 273 as explained above and according to equations (3) and (4) .
• the subchannel signal xisr is also fed to an amplitude/phase equalizer 274 that produces a signal uis (n) , from which the binary data at the rate 8 kHz are recovered, with the help of a data detector 275.
• the data detector just takes the sign of the real and imaginary parts of uis (n) alternatively.
• the binary sequence bs (n) so obtained is fed to the « synchronization pattern and data extraction » block 276, that recognizes the superframe and hyperframe synchronization patterns and delivers the corresponding timing information denoted « timing 1/64 » in the figures.
• the block also separates the bit assign- ment data, denoted « scdatar » and delivered to the OQAM modulator 120 of each subchannel and to the serial/parallel converter 110 in the emitter of the system.

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• Engineering & Computer Science (AREA)
• Computer Networks & Wireless Communication (AREA)
• Signal Processing (AREA)
• Digital Transmission Methods That Use Modulated Carrier Waves (AREA)

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• 1999
  • 1999-11-09 FR FR9914036A patent/FR2800954B1/fr not_active Expired - Fee Related
• 2000
  • 2000-11-09 EP EP00989764A patent/EP1232593A4/de not_active Withdrawn
  • 2000-11-09 WO PCT/US2000/042048 patent/WO2001035561A1/en not_active Ceased

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