EP1497929A2 - Dispositif emetteur-recepteur - Google Patents

Dispositif emetteur-recepteur

Info

Publication number
EP1497929A2
EP1497929A2 EP03747095A EP03747095A EP1497929A2 EP 1497929 A2 EP1497929 A2 EP 1497929A2 EP 03747095 A EP03747095 A EP 03747095A EP 03747095 A EP03747095 A EP 03747095A EP 1497929 A2 EP1497929 A2 EP 1497929A2
Authority
EP
European Patent Office
Prior art keywords
chirp
signals
transceiver according
signal
pulses
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Withdrawn
Application number
EP03747095A
Other languages
German (de)
English (en)
Inventor
Manfred Koslar
Zbigniew Ianelli
Rainer Holz
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Inpixon GmbH
Original Assignee
Nanotron Technologies GmbH
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Priority claimed from DE2002115798 external-priority patent/DE10215798A1/de
Application filed by Nanotron Technologies GmbH filed Critical Nanotron Technologies GmbH
Publication of EP1497929A2 publication Critical patent/EP1497929A2/fr
Withdrawn legal-status Critical Current

Links

Classifications

    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04J—MULTIPLEX COMMUNICATION
    • H04J13/00—Code division multiplex systems
    • H04J13/10—Code generation
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/69—Spread spectrum techniques
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/69—Spread spectrum techniques
    • H04B1/707—Spread spectrum techniques using direct sequence modulation
    • H04B1/7073—Synchronisation aspects
    • H04B1/7075—Synchronisation aspects with code phase acquisition
    • H04B1/70755—Setting of lock conditions, e.g. threshold
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/69—Spread spectrum techniques
    • H04B2001/6912—Spread spectrum techniques using chirp

Definitions

  • the invention relates to a transceiver, a so-called. Transceiver, which is suitable within a transmission system for generating, transmitting as well as receiving and processing chirp signals.
  • chirp signals or combinations of chirp signals of different types are generated and transmitted, and different chirp signals or combinations of chirp signals are also received and processed.
  • dispersive delay lines are designed as surface wave filters (SAW) in such a way that they generate a corresponding chirp signal, that is to say a downchirp or upchirp signal, after excitation with a signal pulse.
  • SAW surface wave filters
  • the transceivers regularly also contain corresponding receiving devices which receive the up- or down-chirp signals and in turn process them further in circuits, where a received up-chirp can be, for example, a logic zero and a received down-chirp can be a logic one in the sense of digital technology.
  • Appropriate SAW filters are used to receive the chirp signals.
  • dispersive SAW filters cannot be produced for any high frequency ranges.
  • the chirp signals must therefore generally be generated in the IF position and then converted into the transmission frequency band using modulation devices. Before transmission, complex measures of image frequency suppression must also be taken.
  • the currently available dispersive SAW filters also have a high insertion loss (for example -24 dB), the compensation of which with suitable broadband amplifiers always results in an increased power consumption of the overall system.
  • chirp signal generation is the tuning of a voltage-controlled oscillator (VCO) with a ramp-shaped signal.
  • VCO voltage-controlled oscillator
  • a ramp-rising voltage at the control input can, for example, generate an upchirp, a ramp-falling voltage a downchirp.
  • this method is very simple and allows chirp signals to be generated directly in the transmission frequency position.
  • the control signal has a discontinuity during the transition from one chirp pulse to the other, as a result of which the output signal has a Switching function is superimposed, with the result that the spectrum widens undesirably. This means that the chirp signal in the transmission frequency position has to be bandpass-filtered before it is transmitted.
  • the ramp-shaped voltage signal at the VCO control input cannot be reset as quickly as desired, so that a sawtooth-shaped control signal results with a long ramp for chirp generation and a short retracting ramp.
  • This in turn undesirably generates a further very short chirp pulse with its own frequency-time characteristic, which is perceived as a disturbance on the receiver side. Blanking the short ramp again produces a switching function with the consequence of the spectral broadening of the transmission signal.
  • Another technically known and easily integrable method is the synthetic generation of any signals in the intermediate frequency position or in the baseband.
  • Sampled and higher-level quantized signals are stored in a memory and, if necessary, converted to digital / analogue and converted into the transmission frequency band.
  • This method is advantageous primarily because of the possible flexibility. It can also be used for the synthesis of chirp signals.
  • the disadvantage of this method is that a comparatively high amount of digital technology and storage space is required, especially when a large number of chirp signals of different characteristics have to be kept available with high quantization.
  • this memory requirement and the need for higher-level D / A converters are always associated with an increased power requirement in the transmitting part of the transceivers and, of course, a larger chip area if the transmitter functions have to be integrated.
  • the generation of chirp signals of different characteristics is associated with high circuit complexity, (for example by providing a large number of different dispersive SAW filters and the associated analog switches in the transmitter), with high power consumption in the transmitter (For example, to compensate for the insertion loss in the dispersive SAW filters), with complex measures for image frequency suppression and for spectral shaping in the transmission frequency band or with an increased need for chip area if complex digital circuits such as higher-level D / A converters have to be implemented.
  • the object of the invention is to provide a transceiver, i.e. transmitter and receiver, for the generation, transmission and reception of chirp signals of different characteristics, which is more simply constructed with respect to the different chirp signals generated than previously known transceivers, which have the greatest possible flexibility in offers the choice of the chirp characteristic that generates chirp signals or combinations of chirp signals in the transmission frequency band without going through an intermediate frequency position and that does without any spectral shaping and filter measures in the transmission band.
  • a transceiver i.e. transmitter and receiver
  • the transceiver according to the invention is used to generate, transmit and receive chirp pulses.
  • chirp pulses are linear frequency-modulated pulses of constant amplitude of duration T, within which the frequency between a lower and an upper frequency changes continuously linearly increasing (upchirp) or falling (downchirp).
  • the difference between the upper and lower frequency represents the bandwidth B of the chirp pulse.
  • the total duration T of the pulse, multiplied by the bandwidth B of the pulse, is referred to as the expansion or spreading factor ⁇ .
  • a chirp pulse passes through a dispersive filter with a suitable frequency-transit time characteristic, then a carrier frequency pulse with a sin (x) / x-shaped envelope is created at the output of this filter - a so-called compressed pulse.
  • the peak power of the compressed pulse is then increased by a factor B «T compared to the peak power of the input chirp pulse.
  • the compression of a chirp pulse is reversible. If a carrier frequency pulse with a sin (x) Ix-shaped envelope of bandwidth B passes through a dispersive filter with a suitable frequency group delay characteristic, then an energy-equal chirp pulse of length T is produced.
  • a message transmission with chirp pulses can be organized so that the symbol alphabet consists of the two elements "upchirp” and "downchirp". For example, an upchirp pulse would be transmitted for a logic zero, and a downchirp pulse for a logic 1.
  • a special form of chirp signals, or the combination of chirp signals, is the folding signal. It arises from the simultaneous generation and superimposition of an upchirp and a downchirp pulse. By choosing a suitable phase offset between the up and down chirp pulse, folding signals can be generated so that they have a positive or a negative deflection after the demodulation at the receiver, so that even with folding pulses, an active transmission of the two logic states (zero and one) is possible.
  • the aim of the invention is to provide a transceiver which generates and emits chirp signals on the transmitter side and which is capable of receiving and demodulating chirp signals on the receiver side.
  • Chirp signals were selected for message transmission because they have a number of advantages over other modulation signals:
  • a short pulse of high peak power can be transformed into an energy-equal but much longer chirp pulse, the transmission power being reduced accordingly, for example to the permitted peak power of a power-limited transmission channel.
  • This pulse is transmitted to the receiver via the transmission channel and compressed there. This again creates a short impulse that is excessive in power above the receive pulse. As a result, a signal with a much higher peak power and thus with a much greater distance to interference signals has been transmitted via the power-limited channel.
  • a chirp transmission system can stand out from other transmission systems that transmit with full signal power via power-limited channels in that its own signals are chirped, i.e. are transmitted with greatly reduced performance without the performance falling compared to the comparison systems.
  • Chirp transceivers are therefore suitable for use in environments where it is important to reduce the radiation exposure from transmitters (low human exposure).
  • Chirp signals are broadband signals, they can be generated in such a way that their spectrum completely fills an available transmission channel of bandwidth B.
  • a carrier frequency pulse with a sin (x) / x-shaped envelope is generated for a symbol to be transmitted and then transformed into a chirp pulse.
  • This carrier frequency pulse has an average width 0 which is determined as the reciprocal of the bandwidth B.
  • the available channel bandwidth B thus determines the possible temporal resolution of a chirp transmission system.
  • a matched filter receiver is used to receive chirp signals. This gain in spread can therefore be interpreted in such a way that the transmitted chirp signal in the receiver is compressed (i.e. despread) using a specially adapted matched filter (the dispersive delay line), while non-chirped signal components, e.g. superimposed interference signals, in the same matched filter of the receiver.
  • the possible spread gain reaches a maximum if the symbol duration 1 / R is equal to the chirp duration T. It becomes minimal when the symbol rate R is equal to the chirp bandwidth B.
  • each individual symbol experiences a time spread beyond its symbol boundaries.
  • a chirp pulse is generated for each symbol, which is longer than the symbol itself.
  • a sequence of temporally overlapping and superimposed chirp pulses then occurs at the output of the dispersive filter.
  • the temporal spread of the symbols can be determined by the quotient of the chirp duration T and symbol duration 1 / R. It reaches its maximum when symbol rate R and chirp bandwidth B match.
  • time-spread transmission can be used to suppress interference. It is assumed that time-spread symbols (in the example chirp pulses) are transmitted by the transmitter, over which broadband interference pulses (for example quasi-direct pulses) are superimposed on the transmission path.
  • the signal mixture of chirp pulses and interference pulses passes through a dispersive filter (chirp filter) at the receiver input, which compresses the chirp pulses into sin (x) / x-shaped pulses. All uncorrelated signal components, ie those that are not in the form of chirp pulses, are expanded in time. Your interference energy is distributed over a longer period of time, i.e. over several neighboring symbols. The likelihood that a single symbol will be destroyed by such a glitch decreases. This also reduces the bit error rate of the transmission.
  • a dispersive filter chirp filter
  • chirp signals for data transmission over broadband and interference-prone message channels offer a number of advantages which predestine them for use in the transceiver according to the invention.
  • a technically well-known variant of the synthesis of transmission signals which is common, for example, in software radio systems, is the digital generation of signals in the intermediate frequency position. This method is also suitable for the representation of chirp signals.
  • a memory for. B. ROM
  • the sampled and quantized chirp signal stored in the IF position.
  • the stored chirp sequence is fed to a digital / analog converter, from whose output the analog chirp signal can be tapped. Due to the high sampling rates required, this method can only be used for the lower frequency ranges (Low IF).
  • Suitable conversion mixers and associated filter measures for image frequency suppression are still required for the conversion into common transmission frequency positions, for example into the ISM band.
  • spectral filtering, image frequency suppression and band limitation in the transmission frequency band should be dispensed with in the sense of the inventive task. About- This strives for the simplest possible structure of the transmission device and the greatest possible flexibility in the selection of the transmission signals.
  • the real part and imaginary part of the chirp baseband signal provided are sampled, quantized and stored in the memory (for example RAM or ROM) as independent bit sequences.
  • the stored baseband sequences can be read out on demand and converted into a chirp signal in the transmission frequency position.
  • FIG. 1 A transmission device is shown by way of example in FIG. 1.
  • Fig. 2 illustrates the signals occurring at the various points in the arrangement.
  • Different chirp baseband signals are stored in a memory (see FIG. 1) as bit sequences (sequencel, sequence2, ...) separated by real part and imaginary part.
  • the selected chirp sequence pair in question is addressed via the “Addressing” block, which is connected, for example, to a digital data source.
  • FIG. 2a shows an example of three information symbols (LOW; HIGH; LOW) of a digital data source that are to be transmitted.
  • two bit sequences are read out via the readout device, for example a parallel / series converter, via the block "Addressing" (FIG. 1).
  • the two bit sequences g2 and g3 are located at the output of the parallel / series converter (see 2b, 2c), which are fed to the inputs of digital / analog converters (DAC).
  • the D / A-converted signals are filtered in the baseband with the two low-pass filters (TP).
  • the signals g4 arising at the output of the low-pass filters and g5 are converted directly into the desired transmission band with the aid of a suitable modulation device (for example an I / Q modulator).
  • a particular advantage of this method is that chirp signals of any characteristic (for example upchirps, downchirps or chirp signals with different BT products and different characteristics) can be stored in the memory; if there is sufficient storage space, they can be called up optionally, so that depending on the requirements the transmission, one or the other of the stored chirp signals can be accessed. It is also conceivable that the chirp sequences required during the commissioning or initialization process can be downloaded to the memory, but can also be replaced by reprogramming if necessary.
  • the transceiver thus has a programmable transmission part which allows the transmission signals to be selected with the greatest possible flexibility and to be transmitted without changes to the hardware (see FIG. 1).
  • Some parameters are necessary for the digital storage of a chirp signal, not least to estimate the memory requirement. This includes the chirp sample rate. It depends on the bandwidth of the chirp signal, its minimum value is determined by the sampling theorem.
  • the proposed method allows the bit quantization to be freely selected in the range from 1, 2 ... n bits. This means that in the simplest case of 1-bit quantization, sequences of the digital symbols "0" and "1" are sufficient to represent a chirp signal in the baseband. In this special case, the connected circuit is further simplified by the fact that the digital / analog converter is no longer necessary. In contrast to known methods of signal synthesis in the baseband, the transceiver according to the invention (according to FIG. 1) can synthesize the transmit signal from two stored binary sequences without an additional digital / analog converter.
  • folding signals are used for transmission.
  • up-chirp and down-chirp signals are superimposed in a certain way so that the resulting signal is purely real. Only one real part has to be stored in the baseband.
  • a simple modulation device for example a mixer or a modulator
  • This halves the effort involved in storing the signals and modulating them into the transmission frequency band.
  • both D / A converters are followed by suitable low-pass filters (LP), which have the task of limiting the spectrum in the baseband to the desired bandwidth.
  • LP low-pass filters
  • the spectral limitation only has to be carried out using these low-pass filters; if necessary, filters of a higher degree must be used.
  • the sampled and quantized baseband signals can be weighted with selectable filter functions (for example with a cosine roll-off characteristic) before being stored in the memory, so that the chirp sequences called up in the transmission case already have simple requirements for the spectral purity of the baseband signals suffice. This reduces the requirements for the downstream low-pass filters. It is also conceivable that this baseband pre-filtering already completely fulfills the spectral requirements for the chirp signal, so that further filter stages are no longer necessary.
  • selectable filter functions for example with a cosine roll-off characteristic
  • the carrier frequency TX 2441.75 MHz is first divided down to 244.175 MHz by a factor of 10: 1 using a two-stage frequency divider.
  • the frequency generated in this way corresponds to the sampling rate with which the chirp signals are to be synthesized in the baseband. Accordingly, 244 samples must be encoded within the symbol duration of 1 ⁇ s.
  • the multiplexer “(MUX) following the memory module serializes the data words placed next to one another in the memory.
  • the data bus from the memory chip is multiplexed into a data bus that is half as wide.
  • the data rate of the bit sequences read from the memory is doubled.
  • the incoming IROM and QROM data streams are logically linked in the subsequent BLOCK MAP (see table), see above that the desired symbols result.
  • the symbols are selected using a 4-bit data word MD. With just two pre-stored bit sequences, all of the symbols listed for the various chirp operating modes can be synthesized.
  • the two bit sequences for I and Q are converted into analog signals using two D / A converters and band-limited with the connected low-pass filters (Leapfrog filter in the example).
  • the output signals of the low-pass filter are then converted into the transmission frequency band using an I / Q modulator.
  • the chirp transmission system which is the subject of this invention, basically allows direct compression and demodulation of the incoming chirp signals into the baseband on the receiver side.
  • each of the receiver variants shown here is provided with an input stage for converting the received signal into the intermediate frequency position in the present invention. If dispersive filters can also be implemented in the desired higher carrier frequency positions in the foreseeable future, the IF stage can accordingly be omitted without affecting the other receiver structures according to the invention.
  • the transceiver For processing incoming chirp signals, the transceiver according to the invention initially has a conversion device (mixer, downconverter) on the receiver side, which converts the incoming signals into the intermediate frequency position.
  • the intermediate frequency signal is then passed to the inputs of two complementary dispersive delay lines, the frequency group delay characteristics of which must be matched to the chirp signal characteristics of the transmitter.
  • the compressed pulses generated at the outputs of the dispersive filters are demodulated with suitable detector circuits into the baseband, where they are converted into data pulses with threshold value comparators, which can be processed in the subsequent digital evaluation circuits of the receiver.
  • the receiving device is dependent on the use of dispersive filters (for example SAW filters), that is to say on the provision of various hardware components.
  • dispersive filters for example SAW filters
  • the entire receiver hardware remains unchanged, the receiver can also be easily used, for example when comparing or in service Inserts can be matched to a newly selected transmit chirp signal. If, for example, the dispersive filters are pluggably connected to the receiving device and can be easily replaced, one can speak for good reason of hardware programming of the receiver part.
  • the transmitting device and receiving device of the transceiver according to the invention can thus be conveniently programmed for the transmission of chirp signals of selectable chirp characteristics.
  • One of the operating modes of the described transceiver is data transmission with the aid of folding pulses.
  • the particular advantage of this operating mode is the low memory requirement for storing the chirp sequences and the simple hardware structure of the transmitter.
  • a folding pulse results from the superimposition of an upchirp and a downchirp pulse generated at the same time.
  • folding signals can be generated in such a way that the carrier frequency pulses arising in complementary dispersive filters after the compression on the receiver side always have the same envelope, but in the case of the “positive” folding pulses they have the same carrier phase and in the case of the "negative” folding pulses have a phase shift of 180 °.
  • Folding pulses are particularly easy to demodulate in the receiver.
  • Up-chirp and down-chirp components can be separated again using complementary dispersive filters with suitable frequency / delay characteristics.
  • a compressed upchirp pulse arises at the output of one delay line, and a compressed downchirp pulse arises at the output of the complementary delay line.
  • coherent demodulation into baseband is achieved.
  • the pulse shape corresponds to a squared sin (x) / x pulse, with a positive deflection in the case of a transmitted positive folding pulse and with a negative deflection in the case of a negative folding pulse.
  • the described direct demodulation of folding signals into the baseband presupposes the presence of dispersive filters for operation in the transmission frequency position (for example in the ISM band around 2.4 GHz). As long as these filters cannot be manufactured or can only be produced with disproportionately high expenditure, demodulation can only take place after the received signal has been converted into the IF position.
  • a prerequisite for successful demodulation of the folding pulses in the receiver is the best possible congruence of the envelopes of the compressed pulses in the receiver.
  • This congruence can only occur if the center frequency of the signal received and mixed down into the IF position matches the center frequency of the two complementary dispersive delay lines as closely as possible.
  • the carrier frequency (center frequency) is only one of many frequency components in the received chirp signal and is in no way distinguished from the others, only methods that can extract the carrier from a pure double-sideband signal can be considered for carrier recovery.
  • Carrier control with the Costas loop is based on the fact that the received signal is converted into the baseband with the help of an I / Q demodulator, the demodulator output signals are low-pass filtered and then multiplied together in order to obtain a control criterion for the phase of the reference carrier ,
  • the VCO that generates the reference frequency can be controlled directly with the product signal.
  • the output signals of the delay lines are demodulated into the baseband using detector stages and then converted into square-wave pulses using threshold comparators. These rectangular pulses are fed to a phase detector, which is followed by a controller. Its output signal influences a voltage controlled oscillator (VCO) with which the local oscillator (LO) of the system is generated.
  • VCO voltage controlled oscillator
  • compressed chirp pulses are generated at the outputs of the complementary delay lines, the time offset of which represents a measure of the deviation of the IF center frequency from the center frequency of the delay lines, and which are used as a control criterion for the frequency of the reference carrier (LO) can.
  • the phase detector checks for congruence of the demodulated compressed pulses, its output voltage varies in size and polarity depending on the time offset of the pulses.
  • the following controller changes the control voltage of the VCO 's, to the envelopes of the compressed chirp pulses are superimposed.
  • the control loop is engaged and the prerequisite for the multiplicative demodulation of the folding signals is given.
  • Frequency synchronization thus does not take place between the carrier frequency of the received signal and the reference carrier (LO), as is customary in the known methods, but rather between the IF signal and the characteristics of the dispersive filter.
  • the system does not synchronize itself to a received carrier signal, but conversely it synchronizes the received signal to a system-specific reference, the center frequency of the complementary dispersive group delay filter.
  • the incoming signal is shifted in frequency in the IF position until its center frequency and the center frequency of the dispersive filters lie one above the other. This means that the system also easily compensates for changes in the filter center frequency due to heating, aging or other influences.
  • a data sequence can be preceded by a preamble of folding pulses, which is used especially for the oscillation of the frequency control loop.
  • the synchronization achieved is also maintained during the subsequent transmission of the data pulses, it is irrelevant whether positive or negative folding pulses or longer sequences of the same polarity are received. If folding pulses occurring in bursts are received with the receiver arrangement shown, then each data burst must be preceded by a preamble for synchronization.
  • a preamble of folding pulses is first transmitted before the transmission of a data burst.
  • the VCO actuating voltage is sampled with a sample and hold element and recorded for the duration of the data burst.
  • the structure of the receiver device allows both the reception of folded signals and simple chirp signals (e.g. up-chirp / down-chirp). In the latter case, the control loop described can be switched off. It is then sufficient to use a simple PLL circuit with a quartz reference to generate the local oscillator.
  • a data sequence consisting of up-chirp pulses (logically HIGH) and down-chirp pulses (logically LOW) is preceded by a preamble of folding pulses which are used for frequency synchronization serves, after the frequency control loop has snapped in, the VCO control voltage is sampled and recorded for the duration of the data burst.
  • a preamble of folding pulses which are used for frequency synchronization serves, after the frequency control loop has snapped in, the VCO control voltage is sampled and recorded for the duration of the data burst.
  • a further embodiment of the invention is an automatic frequency control for an upchirp / downchirp transmission system, illustrated by way of example in FIG. 4.
  • a series of alternating upchirp and downchirp pulses are transmitted in a preamble. Rectangular pulses appear at the inputs of the phase detector in the symbol cycle, which are shifted in time from input to input.
  • this offset is exactly half a symbol period, that is 180 °.
  • the phase detector is designed for this case so that its output signal in size and polarity reflects the instantaneous phase shift and accordingly disappears in the locked state.
  • the frequency control loop shown in FIG. 4 can then also be used for frequency control of up / down chirp systems. Initially, this only applies to the preamble. For the duration of the subsequent data sequence, the VCO input signal must be clamped back to the voltage value of the locked state.
  • the phase detector can then be made switchable so that both types of transmission can work with the same frequency control loop.
  • the frequency control described can only be used for up / down chirp transmission systems if up and down chirp symbols are received alternately at least until it snaps into place, for example within a preamble preceding the data burst.
  • the subsequent data signal is generally characterized by the irregular sequence of upchirp signals (for example logically HIGH) and downchirp signals (in the example correspondingly logically LOW). This also includes longer pulse trains of the same polarity. If the symbol period is known, however, it is possible to insert the missing symbols as dummy symbols in two branches between two symbols of the same polarity, which are staggered by more than one period. For this purpose, a block “restore sequence” is connected upstream of the phase detector in FIG. 4.
  • the uninterrupted symbol sequences thus generated in both branches are then fed to the phase detector, the rest of the control loop operates in the known manner.
  • the prerequisite for this method is that the temporal To ensure this, the symbol sequences in the transmitter can be scrambled accordingly before the transmission, with the aim that the number of successive symbols of the same polarity does not exceed a fixed value k.
  • the frequency synchronization established within a preamble can also be maintained during the subsequent transmission of data sequences of any length.
  • the transmission of digital data sequences requires not only frequency synchronization on the receiver side, but generally also clock synchronization.
  • the aim is to derive the symbol clock from the received signal with correct frequency and phase.
  • Technically common methods are clock derivation with synchronous demodulator for frequency-modulated signals or clock recovery from the demodulated baseband signals, in which the low-pass filtered baseband signals are summed and then the clock frequency is filtered out from the sum signal with a bandpass filter.
  • Still other methods provide a separate PLL circuit for clock recovery.
  • transceiver With the transceiver according to the invention it is possible to send data sequences consisting of upchirp / downchirp pulses or data sequences consisting of folding pulses and to demodulate them asynchronously on the receiver side.
  • FIG. 5 shows a receiver device for up / down chirp transmission with subsequent clock derivation.
  • the chirp signals arriving at the receiver input are first converted into the IF position, automatically and asynchronously compressed with complementary dispersive delay lines and demodulated into the baseband with detector circuits.
  • the square-wave pulses arising at the outputs of the subsequent threshold value comparators only have to be linked to one another by a suitable logic element (for example an EXCLUSIVE OR gate) in order to derive the symbol clock.
  • the symbol clock (CLOCK) is fed to the clock input of a JK flip-flop, the inputs J and K are connected accordingly to the comparator outputs.
  • the particular advantage of the method for the asynchronous derivation of the symbol clock is that the receiver device immediately follows every change of the symbol rate and therefore the symbol clock on the transmitter side, without special switching procedures or reinitializations in the receiver being necessary. This is the first time that a smooth variation in the data rate of a transmission system is possible.
  • the symbol clock can be derived in the same way, assuming the steady state of the frequency control.
  • Fig. 6 shows a receiving device for the folding pulse transmission.
  • the input circuit of the receiving device again consists of a converter and the two dispersive filters.
  • the output signals of both delay lines are multiplied directly with one another, producing a bipolar baseband signal.
  • a variant for the derivation of the symbol clock is the two-way rectification of this baseband signal and the subsequent evaluation with a threshold value comparator. Its output signal also carries the symbol clock (CLOCK).
  • FIG. 7 Another variant of the clock derivation for folding pulses is shown in FIG. 7.
  • This circuit takes advantage of the fact that when receiving folding pulses and when the frequency control is in a steady state, the comparator output signals carry the symbol clock in both branches equally, so that the clock derivation can be limited to one of the branches.
  • it is advantageous to link both comparator output signals via a logic AND gate.
  • the system clock (CLOCK) is present at the output of this AND gate.
  • FIG. 8 Another variant of the clock derivation for folding pulses is shown in FIG. 8.
  • the output signals of both delay lines are multiplied directly with one another, producing a bipolar baseband signal. This signal is compared on two threshold comparators, each with a positive and a negative threshold.
  • the output signals of the Comparators are linked with each other via a logical OR gate to derive the system clock (CLOCK).
  • CLOCK system clock
  • the transceiver In both operating modes, the transceiver according to the invention allows the comparator output signals to be gated on the receiver side. This gating is based on the operating case with a fixed symbol rate or the symbol rate known to the recipient. A circuit part for clock derivation is still required.
  • 9 shows a variant of the gating which is used in the transceiver. 10 shows an example of the associated signals.
  • FIG 9 shows a schematic representation of a switch which is actuated via the “time control” block.
  • the CLOCK signal g8 has been generated in an upstream stage for clock derivation.
  • the switch With the signal g9, the switch is opened and closed.
  • the switch is initially closed in the rest position
  • the first arriving symbol clock pulse is recognized by the time control and, after a short time delay (controlled by signal g9), the switch is opened and thus further impulses which are blocked within of a certain interval, which is smaller than a symbol period.
  • the switch is closed again.
  • the next (expected) symbol clock pulse can happen and triggers the blockage again.
  • interference pulses that occur within a symbol interval are suppressed.
  • This variant is particularly suitable for starting the system. If, after activation of the system, an interference pulse first undesirably triggers the blockage, then the gate is opened again after a time that is shorter than a clock period. The system does not remain in the blocked state and can already process the upcoming symbol clock pulse.
  • FIG. 11 An exemplary embodiment for this arrangement is shown in FIG. 11.
  • a logical AND gate takes over the function of the switch, and a monoflop determines the length of the blockage interval.
  • a special form of the gating according to the invention is that a blockage interval of variable length is used.
  • the blockage interval can be particularly short; in the steady state, it is possible to switch to a longer blockade interval, which in extreme cases is only a little shorter than the symbol duration itself.
  • a further characteristic is that a symbol clock pulse closes the gate for the duration of a blockage interval, then opens for the duration of an opening interval (within which the next symbol clock pulse is expected) and then closes again for the duration of a blockage interval - a process that continues repeated.
  • This variant is suitable for operation in the steady state.
  • a chirp pulse is received by the receiving device of the transceiver, converted into the IF position and fed to complementary dispersive filters, then not only does a compressed pulse arise at the output of one of the two filters, but also a stretched chirp pulse on the complementary chirp -Filter.
  • the stretched chirp pulses appear in each of the two branches as intrinsic interference signals that must be taken into account in the detection and subsequent discrimination.
  • the detected signals are compared with a threshold value in each path. The effect described requires that the threshold value of the comparator is always in the range between the peak values of the stretched pulse and the compressed pulse. This alone limits the dynamics of signal detection.
  • the receiving system should also be able to react to changes in performance at the detector input.
  • FIG. 1 A receiving device according to the invention is shown in FIG. 1
  • the incoming signal is first converted to the IF position and fed to the inputs of two complementary dispersive filters.
  • the compressed chirp pulses at the output of each of the two filters are fed to an envelope detector, an average value detector and a peak value detector in both branches.
  • a threshold value for the subsequent comparator is derived from the output signals from the mean value detector and peak value detector.
  • the threshold value can variably assume any value between the peak and mean value of the detected signal.
  • the position of the threshold value can be controlled digitally within this interval.
  • the output signals of the envelope detectors are compared with the threshold values thus generated in both branches.
  • the signals COMPJJP and COMP_DOWN are available at the outputs of the two comparators for further digital processing.
  • the threshold value comparator In the event that there are no received signals, the threshold value comparator must offer the highest possible sensitivity, but the background noise of the receiver device must not lead to the switching of the comparator. Therefore, in a special embodiment of the invention, the lower limit of the threshold value is set such that the threshold value in the idle state (in the state in which it is ready to receive) is always higher than the detection signal of the background noise of the receiver device. For this purpose, a voltage U jmin is added to the threshold value formed from the mean value and peak value in both branches, with the result that the threshold value at the comparator input is always higher than the noise amplitude at the detector output.
  • the transceiver with the inventive combination of detector and comparator with adaptive threshold defines the threshold values of the amplitude discrimination in such a way that reliable detection of complementary chirp signals is possible even when the signal changes at the detector input.
  • the NANONET transceiver is shown in the variant which is advantageous here as a highly integrated circuit which is provided for the transmission of digital data sequences and which is a complete transmitter (from digital input to RF- Power amplifier), a complete analog receiver (from the antenna input to the output for the demodulated and digitized received data), a programmable analog and a programmable digital control device.
  • the analog control device consists of power management, analog / digital converter, power sources, battery charge monitoring, alarm signaling and other components. All essential functions of this function block can be initialized and controlled by the application software.
  • the programmable digital control device which communicates with external microcontrollers via a serial peripheral interface (SPI), provides various control functions for the analog part of the IC.
  • SPI serial peripheral interface
  • this block already takes over important functions of the protocol stack up to the MAC layer, error correction, real-time clock, wake-up management, interrupt requests, automatic generation of acknowledge signals and other tasks. All functions of this block are initialized and controlled via the application software on an external microcontroller.
  • TX Transmitter
  • the Analogue Sensor Interface (1) is used to record the sensor data in several channels. This module also provides a power source for supplying the connected sensors.
  • the application software starts reading out the connected sensors, the sensor data is A / D converted by the Analogue Sensor Interface and transferred to the block control register (x) of the digital part.
  • the sensor data can be transferred to the application via the lines DilO1, ... DiO4 shown.
  • the centerpiece of the transmitter is the I / Q modulator (2).
  • the digital symbols to be transmitted are mapped in the block pulse sequence (3) to pre-stored bit sequences which represent the real part and the imaginary part of the transmission signal in the baseband. These bit sequences are band-limited with the low-pass filters (3) and (4) and fed to the inputs of the I / Q modulator (2).
  • the carrier signal for the I / Q modulator is generated in the block frequency synthesis (5). This frequency synthesizer generates either the carrier for the transmitter-side direct modulation in the transmission frequency band or the carrier for the receiver-side downmixing in the IF position.
  • the analog switch (6) is controlled by the signal RX / TX and carries out the carrier switching between transmit and receive operation.
  • the output signal of the I / Q modulator (2) is fed to a preamplifier stage (7) and then to the power amplifier (8).
  • the output power of the power amplifier can be controlled by the digital part via the block Power Control (9).
  • the power amplifier can be switched off for the duration of the reception period via the RX TX signal.
  • the block diagram on the transmitter side also shows an internal oscillator OSC (10), prepared for the connection of an external quartz, and Battery Management (11) for monitoring the state of charge of the battery.
  • the received signal from a connected antenna is coupled into the Low Noise Amplifier (LNA) (12).
  • LNA Low Noise Amplifier
  • the LNA can be switched off with the signal RX / TX for the duration of the transmission period. Its gain is controlled by the AGC block (13).
  • the LNA is followed by the downmixer (14), which converts the received signal into the intermediate frequency position.
  • the downstream amplifier (15), like the LNA, is included in the automatic gain control (AGC). Its output signal is extracted from the transceiver.
  • AGC automatic gain control
  • the circuit is prepared in such a way that a SAW component can be connected directly to the IF amplifier (15), which consists of two dispersive delay lines with complementary group delay characteristics.
  • the output signals of both delay lines are coupled into the IC at the inputs of the multi-stage regulated amplifiers (16) and (17).
  • each of the input amplifiers (16) and (17) in the circuit is followed by a detector stage (18) or (19) and downstream low-pass filters (20) or (21).
  • a threshold comparator (22) or (23) follows each of the two low-pass filters.
  • the threshold values for both comparators are adaptive and are determined in the block threshold (24) from the TP output signals themselves.
  • the comparator output signals are processed in the digital part, initially in the bit decoder.
  • a multiplier (25) is available in the receiving section for demodulating folding pulses, with which the output signals of the dispersive filter are multiplied.
  • the multiplier is followed by an amplifier stage 26 and two threshold value detectors (27) and (28) for the detection of the bipolar folding signals.
  • the threshold values for both comparators are determined adaptively within the Threshold (24) block.
  • the output signals of the two comparators are processed further in the digital part.
  • the microcontroller interface (29) serves to transmit the send and receive data and control information between the external microcontroller and the transceiver chip. It also synchronizes the data communication between the two components.
  • the FIFO (30) buffers received or transmitted data and realizes the time decoupling of the processes in the transceiver chip and the external microcontroller.
  • the MAC state machine (31) controls analog and digital blocks depending on the access method used (CSMA / CA, TDMA), controls the sequence of the send and receive processes and evaluates received protocol information (packet type, automatic destination address comparison, determination of the Package length etc.).
  • the data to be sent or received are processed in the digital bit processing unit (32) (frame synchronization, checksum generation and control, forward error correction, scrambling / descrambling, optional encryption / decryption).
  • the symbols received by the analog part are detected by the bit detector (33) and the bits are synchronized.
  • the power management (34) switches off external and internal power supplies (power saving mode) and controlled again by internal events (wake-up timer, battery management).
  • the microcontroller management (35) deactivates the power supply and all connections to the external microcontroller. After switching on the power supply by the power management, the start-up of the microcontroller is controlled here.
  • the real-time clock (36) contains a real-time clock, which is used for controlling the access method (TDMA) and the power saving mode. It is also used for time recording for applications. The wake-up timer saves the time for exiting the power saving mode for power management.
  • the analog blocks of the transceiver are controlled or queried via the control register (37).
  • the DilOs digital input / output
  • Corresponding external SAW (Surface Acoustic Wave) components have been used regularly to receive chirp signals.
  • the present invention also allows the chirp signal reception and its detection to be carried out without corresponding external SAW modules.
  • the chirp signal passes through a differential comparator upon receipt and the received signal is processed in a shift register which is connected to a corresponding exclusive or interconnected reference shift register.
  • an external SAW module - FIG. 16 - can be dispensed with, which makes the receiver very inexpensive and simple.
  • DDDL digital differential dispersive line
  • the invention is not limited to the disclosed transceiver alone, but also the chirp signal reception, as disclosed in FIGS. 15 and 16, can be implemented independently of the transmission unit of the transceiver.
  • the transceiver described above can operate in the ISM band at about 2.4 GHz.
  • a chirp pulse with a bandwidth is used for each symbol transmitted of 80 MHz (with a roll-off factor of 0.25 results in an effective bandwidth of 64 MHz).
  • the energy which is distributed over the wide frequency range of 80 MHz is "collected” again in the receiver, so that a very short and high pulse is generated (sin x / x function).
  • an external SAW filter surface acoustic wave
  • FIGS. 15 and 16 either an external SAW filter (surface acoustic wave) can be used in the receiver or the solution as described in FIGS. 15 and 16, so that only those energy parts that belong to the chirp pulse are "stacked" on top of one another. while all others (e.g. interference signals) pass the filter.
  • the actual signal stands out clearly from the background.
  • This "system gain” can be freely selected within wide limits by increasing or reducing the length of the chirp pulse. In the method described above, a chirp pulse duration of 1 ⁇ sec and an effective bandwidth of 64 MHz (at 18 db) are sufficient.
  • the system uses extremely little power, about 5 mA in initial operation and 33 mA when sending 10 mW.
  • the reason for this is the largely analog signal processing, which does not require any complex digital signal processors for echo suppression.
  • the described transceiver chip can be implemented in silicon germanium technology, but also in CMOS technology.
  • the special applications of the described transceiver are in factory automation, for example for monitoring machines.
  • Another good area of application is intelligent access control with wireless keys (e.g. chip cards, active RFID) to identify people, animals or goods wirelessly.
  • the active RFID logistics tags have a greater range and can also be reprogrammed.
  • the use for alarm systems is very suitable, in particular also alarm systems for fire or movement, and bidirectional communication between a transceiver and a corresponding chirp sensor is possible.
  • an application for networking computers is also possible, for. B. the network between a personal computer and a PDA or between a personal computer and the periphery (mouse, keyboard).
  • the DDDL consists of an input shift register, which receives the output signal of a differential comparator.
  • Each cell of the input shift register is linked to an exclusive OR block, which is also connected to the output of a memory which contains a stored reference for an upchirp signal and / or a stored sequence of a downchirp signal.
  • the individual results of the large number of exclusive OR blocks are summed up and made available to the correlator output.
  • the total result is processed in a comparator module for "UP" or for "DOWN” and then the corresponding chirp signal is detected at the output of the comparator and the result is made available.
  • the comparator also receives a threshold signal and outputs a chirp-detected signal at the output if the comparison result between the correlator output signal and the threshold signal can be detected accordingly.

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  • Engineering & Computer Science (AREA)
  • Computer Networks & Wireless Communication (AREA)
  • Signal Processing (AREA)
  • Radar Systems Or Details Thereof (AREA)
  • Transceivers (AREA)

Abstract

L'invention concerne un dispositif émetteur-récepteur caractérisé en ce que des signaux chirp différents en termes de produit BT et/ou de caractéristique temps-fréquence peuvent être enregistrés dans une mémoire de façon à pouvoir être appelés sélectivement et placés dans la bande de fréquences d'émission directement convertis en liaison montante. Lors de cette opération, aucune bande de fréquence-image n'est produite, ce qui permet de faire l'économie de filtres passe-bande dans la gamme de fréquences porteuses. Une démodulation directe et automatique dans la bande de base est également possible dans le récepteur, cette démodulation dépendant de la faisabilité des filtres dispersifs à fonctionnement asynchrone (par exemple sous forme de composants SAW) pour la bande de fréquences porteuses.
EP03747095A 2002-04-10 2003-04-08 Dispositif emetteur-recepteur Withdrawn EP1497929A2 (fr)

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DE2002115798 DE10215798A1 (de) 2002-04-10 2002-04-10 Sende-Empfangsvorrichtung
DE10215798 2002-04-10
DE10252626 2002-11-11
DE10252626 2002-11-11
PCT/EP2003/003617 WO2003092183A2 (fr) 2002-04-10 2003-04-08 Dispositif emetteur-recepteur

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JP (1) JP2005522970A (fr)
KR (1) KR20040111501A (fr)
CN (1) CN1653709A (fr)
AU (1) AU2003226794A1 (fr)
CA (1) CA2480846A1 (fr)
IL (1) IL164343A0 (fr)
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AU2003226794A1 (en) 2003-11-10
IL164343A0 (en) 2005-12-18
KR20040111501A (ko) 2004-12-31
JP2005522970A (ja) 2005-07-28
WO2003092183A3 (fr) 2004-03-25
CA2480846A1 (fr) 2003-11-06
WO2003092183A2 (fr) 2003-11-06
CN1653709A (zh) 2005-08-10

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