EP2870660B1 - Système d'antennes pour communication satellite large bande dans la plage de fréquences ghz, doté d'un réseau d'alimentation - Google Patents

Système d'antennes pour communication satellite large bande dans la plage de fréquences ghz, doté d'un réseau d'alimentation Download PDF

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EP2870660B1
EP2870660B1 EP13734662.3A EP13734662A EP2870660B1 EP 2870660 B1 EP2870660 B1 EP 2870660B1 EP 13734662 A EP13734662 A EP 13734662A EP 2870660 B1 EP2870660 B1 EP 2870660B1
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EP
European Patent Office
Prior art keywords
antenna system
antenna
microstrip line
waveguide
networks
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German (de)
English (en)
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EP2870660A1 (fr
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Joerg Oppenlaender
Michael Wenzel
Alexander MOESSINGER
Michael Seifried
Christoph Haeussler
Alexander Friesch
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Lisa Draexlmaier GmbH
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Lisa Draexlmaier GmbH
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    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q21/00—Antenna arrays or systems
    • H01Q21/06—Arrays of individually energised antenna units similarly polarised and spaced apart
    • H01Q21/061—Two dimensional planar arrays
    • H01Q21/064—Two dimensional planar arrays using horn or slot aerials
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q13/00—Waveguide horns or mouths; Slot antennas; Leaky-waveguide antennas; Equivalent structures causing radiation along the transmission path of a guided wave
    • H01Q13/02—Waveguide horns
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q13/00—Waveguide horns or mouths; Slot antennas; Leaky-waveguide antennas; Equivalent structures causing radiation along the transmission path of a guided wave
    • H01Q13/02—Waveguide horns
    • H01Q13/025—Multimode horn antennas; Horns using higher mode of propagation
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q13/00—Waveguide horns or mouths; Slot antennas; Leaky-waveguide antennas; Equivalent structures causing radiation along the transmission path of a guided wave
    • H01Q13/02—Waveguide horns
    • H01Q13/0275—Ridged horns
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q15/00—Devices for reflection, refraction, diffraction or polarisation of waves radiated from an antenna, e.g. quasi-optical devices
    • H01Q15/02—Refracting or diffracting devices, e.g. lens, prism
    • H01Q15/08—Refracting or diffracting devices, e.g. lens, prism formed of solid dielectric material
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q15/00—Devices for reflection, refraction, diffraction or polarisation of waves radiated from an antenna, e.g. quasi-optical devices
    • H01Q15/24—Polarising devices; Polarisation filters 
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q19/00—Combinations of primary active antenna elements and units with secondary devices, e.g. with quasi-optical devices, for giving the antenna a desired directional characteristic
    • H01Q19/06—Combinations of primary active antenna elements and units with secondary devices, e.g. with quasi-optical devices, for giving the antenna a desired directional characteristic using refracting or diffracting devices, e.g. lens
    • H01Q19/08—Combinations of primary active antenna elements and units with secondary devices, e.g. with quasi-optical devices, for giving the antenna a desired directional characteristic using refracting or diffracting devices, e.g. lens for modifying the radiation pattern of a radiating horn in which it is located
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q21/00—Antenna arrays or systems
    • H01Q21/0006—Particular feeding systems
    • H01Q21/0025—Modular arrays
    • H—ELECTRICITY
    • H01—ELECTRIC ELEMENTS
    • H01Q—ANTENNAS, i.e. RADIO AERIALS
    • H01Q21/00—Antenna arrays or systems
    • H01Q21/0006—Particular feeding systems
    • H01Q21/0075—Stripline fed arrays

Definitions

  • the invention relates to an antenna system for broadband communication between earth stations and satellites, in particular for mobile and aeronautical applications.
  • the weight and size of the antenna system are very important as they reduce the payload of the aircraft and cause additional operating costs.
  • the problem is therefore to provide antenna systems that are as small and light as possible, which nevertheless meet the regulatory requirements for transmitting and receiving when operated on mobile carriers.
  • envelopes envelope curves or masks
  • envelopes envelope curves or masks
  • the values specified for a certain distance angle must not be exceeded when the antenna system is transmitting. This leads to strict Requirements for the angle-dependent antenna characteristics.
  • the antenna gain must drop sharply as the distance from the target satellite increases. Physically, this can only be achieved by very homogeneous amplitude and phase assignments on the antenna. Therefore, parabolic antennas are typically used which have these properties.
  • parabolic mirrors are, however, only very poorly suited because of their size and because of their circular aperture.
  • the antennas are mounted on the fuselage and must therefore only be as low as possible because of the additional air resistance.
  • Antennas that are designed as sections of paraboloids (“banana-shaped mirrors") are possible, but their efficiency is very low due to their geometry.
  • Antenna fields which are made up of individual radiators and have suitable feed networks, can, on the other hand, be implemented in any geometry and any length to aspect ratio without the antenna efficiency suffering.
  • antenna fields of very low height can be implemented.
  • the problem arises in particular when the reception frequency band and the transmission frequency band are far apart (e.g. in the Ka band with reception frequencies around 18 GHz - 21 GHz and transmission frequencies around 28 GHz - 31 GHz) that the Individual radiators in the fields must support a very large bandwidth.
  • horn radiators are by far the most efficient individual radiators in fields.
  • horn antennas can be designed as broadband.
  • reception and transmission frequencies are also far apart in terms of frequency and the distance between the beam centers has to be designed according to the minimum useful wavelength of the transmission band for regulatory reasons, the horns are regularly so small that the reception band can no longer be supported by them.
  • the minimum useful wavelength is only approx. 1 cm. So that the radiating elements of the antenna field are close together, i.e. no parasitic side lobes (grating lobes) occur, the aperture area of a square horn antenna may only be approx. 1 cm x 1 cm.
  • conventional horns of this size only have a very low efficiency in the reception band at approx. 18 GHz - 21 GHz, since they have to be operated close to the cut-off frequency due to the finite opening angle. Such horns can no longer support the Ka reception band or their efficiency decreases very sharply in this band.
  • the horn radiators should generally support two orthogonal polarizations, which further restricts the geometric scope, since an orthomode signal converter, so-called transducer, is necessary at the horn output.
  • An implementation of the orthomode signal converter in waveguide technology regularly fails because there is not enough installation space available at higher GHz frequencies.
  • feed networks for fields from horn radiators which are implemented using hollow conductor technology, generate only very low dissipative losses.
  • the individual horn radiators of the fields are fed by waveguide components and the entire feed network also consists of waveguide components.
  • the reception and transmission bands are far apart in terms of frequency, however, the problem arises that conventional waveguides can no longer support the frequency bandwidth required.
  • the required bandwidth in the Ka-band is more than 13 GHz (18 GHz - 31 GHz).
  • Conventional rectangular waveguides cannot efficiently support such a large bandwidth.
  • grating-lobe-free antenna diagrams can be achieved if the phase centers of the individual radiators are less than a wavelength of the maximum useful frequency apart, as is the case, for example, in FIG EP 1 921 709 A1 is revealed. It is also known that parabolic amplitude assignments of such antenna fields can suppress the side lobes of the antenna diagram (e.g. JD Kraus and RJ Marhefka, "Antennas: for all applications", 3rd ed., McGraw-Hill series in electrical engineering, 2002 ). Using special amplitude assignments, an antenna diagram that is optimally adapted to the regulatory mask can be achieved for a given antenna size (e.g. DE 10 2010 019 081 A1 ; Seifried, Wenzel et. al.).
  • GB 2,426,876 A shows an antenna system with several horn antennas, which is constructed in several layers, but is not connected to a separation of microstrip lines supporting different polarizations.
  • U.S. 5,453,751 A shows the handling of circularly polarized signals in antenna fields.
  • US 6,201,508 B1 , WO 2008/069369 A1 , GB 2 247 990 A and EP 2 006 956 A2 show further design options for antenna systems with multiple horn antennas, but none for the with reasonable manufacturing effort Satellite communications in aircraft can generate sufficient performance and compliance with regulatory requirements.
  • the object of the invention is to provide a broadband antenna system in the GHz frequency range, in particular for aeronautical applications, which allows a regulatory-compliant transmission operation with maximum spectral power density with minimal dimensions and at the same time has high antenna efficiency and low inherent noise in reception operation.
  • the antenna system consists of at least two modules, each module containing at least two individual radiators, and microstrip line networks to feed the individual radiators within a module and waveguide networks are used to feed the modules, the individual radiators having a first and a second polarization and the two polarizations orthogonal to each other are.
  • microstrip lines are used where there is very little installation space available (the point at which the individual radiators are fed). Although microstrip lines have significantly higher dissipative losses than waveguides, they require much less installation space. In addition, the losses can be severely limited here by only combining as many primary horn antennas in the modules as are necessary to obtain sufficient installation space for waveguide components. The length of the microstrip lines thus remains comparatively short.
  • the inter-modular feed networks are then designed as very lossless waveguides.
  • the manufacture of densely packed antenna systems can be greatly facilitated by the fact that they are made up of several layers and the microstrip line networks of the two orthogonal polarizations are located between different layers.
  • the modules of the antenna system can then be assembled from a few layers.
  • the layers are advantageously made of aluminum or similar electrically conductive materials, which can be structured with the known structuring processes (milling, etching, lasers, wire eroding, water cutting, etc.).
  • the microstrip line networks are structured on a substrate using known etching processes.
  • the first and second polarizations are linear polarizations.
  • the signals of the two orthogonal polarizations are carried in separate feed networks, which has the advantage that, with the help of appropriate components, such as polarizers or 90 ° hybrid couplers, both linearly polarized signals and circularly polarized signals can be sent or received.
  • the antennas can have the smallest possible size and still a regulatory-compliant transmission operation with maximum spectral power density is possible, it is also provided according to an advantageous further development of the invention that at least some of the individual radiators are dimensioned so that for direct adjacent individual radiators the distance between the phase centers of the individual radiators is less than or equal to the wavelength of the highest transmission frequency at which no parasitic side lobes (grating lobes) may occur (reference frequency in the transmission band).
  • At least one direction is defined by the antenna field, see above that for this direction the distance between the phase centers of the individual radiators is less than or equal to the wavelength of the highest transmission frequency at which no parasitic side lobes (grating lobes) may occur.
  • grating lobes In this direction, preferably along a straight line through the antenna field, directly adjacent individual radiators are then close together, so that no parasitic side lobes ("grating lobes") can occur in the corresponding section through the antenna diagram. Otherwise, these grating lobes would lead to a strong reduction in the spectral power density permitted by regulations.
  • all known radiating elements that support two orthogonal polarizations can be used as single radiators. These are, for example, rectangular or round horn radiators, patch antennas, single dipoles offset by 90 °, crossed dipoles, or appropriately arranged slot radiators.
  • Such rectangular modules can be combined to form antenna fields in a space-saving manner.
  • the rectangular modules can be fed in a relatively simple manner with binary microstrip line networks.
  • the individual radiators In order to implement antennas with the lowest possible dissipative losses, it is advantageous to design the individual radiators as horn radiators, which are among the antennas with the lowest losses. Both horn radiators with a rectangular and a round aperture opening can be used. If no grating lobes should appear in any section through the antenna diagram, horn radiators with a square aperture opening are advantageous, the size of the aperture opening being selected so that the distance between the phase centers of directly adjacent horn radiators is less than or equal to the wavelength of the highest transmission frequency as the reference frequency, where no grating praises are allowed.
  • the individual radiators are designed as horn radiators in such a way that they are equipped with symmetrical geometric constrictions, i.e. constrictions, in the two polarization planes and at their output for each of the two orthogonal polarizations separately via the the geometric constriction belonging to the respective polarization direction are fed.
  • geometric Constrictions can greatly increase the range of the horns.
  • the horns can advantageously also be designed as dielectrically filled horns.
  • the effective wavelength increases in the horns and these are able to support much larger bandwidths than would be the case without the filling.
  • dielectric fillings lead to parasitic losses through the dielectric, these losses remain comparatively small, especially with very small horns.
  • a dielectric filling with a relative permittivity of approx. 2 is sufficient. With horns only a few centimeters deep, this leads to losses of ⁇ 0.2 dB when using suitable materials.
  • the horn radiators are designed as stepped horns ("stepped horns"). By setting the width and length of the steps and the number of steps, the antenna can then be optimally adapted to the relevant frequency bands.
  • the horn antennas are designed in such a way that they support two orthogonal linear polarizations. With such horn antennas, isolations of far more than 40 dB can be achieved. Isolation values of this kind are required in particular for signal encodings with high spectral efficiency.
  • a further improvement in the reception performance, particularly in the case of very small horn radiators, can be achieved by equipping the individual horn radiators with a dielectric cross septum or a dielectric lens.
  • the insertion loss (S 11 ) in the receiving band can be significantly reduced by such structures, even if the aperture areas of the individual radiators are already so small that a free-space wave would be almost completely reflected without these additional dielectric structures.
  • the horn radiators of the antenna array are fed in parallel according to a further advantageous further development of the invention. This is most effective when the microstrip lines and the waveguides are constructed as binary trees, as the number of required Power splitter in the general case of any values of the total number of individual radiators N and any values of the number of individual radiators in a module N i is so minimal.
  • the binary trees are neither completely nor completely symmetrical.
  • the feed networks of the antenna system can be designed as complete and completely symmetrical binary trees and all individual radiators can have feed lines of the same length, ie also attenuations of the same type.
  • microstrip lines are located on a thin substrate and are guided in closed metallic cavities, the cavities typically being filled with air.
  • a substrate is typically referred to as thin if its thickness is smaller than the width of the microstrip lines.
  • the cavities through which the microstrip lines are guided are advantageously structured directly with the metallic layers. If the cavities are designed as notches or depressions in the metallic layers lying above and below the microstrip line, then the microstrip line lies together with its substrate in a cavity which consists of two half-shells.
  • the walls of the cavity can be electrically closed by providing the substrate with electrical through-contacts (vias). "Fences" from Vias can almost completely prevent the loss of electromagnetic power in such arrangements.
  • reception and transmission bands of the antenna are very far apart in terms of frequency, then it may be the case that standard hollow conductors (rectangular hollow conductors) can no longer support the required bandwidth.
  • waveguides with geometrical constrictions (constrictions) along the direction of propagation of the electromagnetic wave.
  • constrictions can greatly increase the useful bandwidth.
  • the number and arrangement of the constrictions depend on the design of the antenna system.
  • an advantageous further development of the invention consists in that dielectrically filled waveguides are used for the waveguide feed networks.
  • Such waveguides require significantly less installation space than air-filled waveguides.
  • a part or a whole waveguide network can also consist of dielectrically filled waveguides. Partial filling is also possible.
  • the feed networks For further processing of the signals, for example by coupling a low-noise amplifier ("Low-Noise Amplifier”, LNA) to the receiving feed network and / or a power amplifier ("High Power Amplifier", HPA) to the transmitting feed network, it can be advantageous the feed networks must be equipped with frequency diplexers. Such frequency diplexers separate the receive from the transmit band.
  • Waveguide diplexers are particularly advantageous because they can achieve a very high level of insulation and also have very little attenuation.
  • each module of the antenna field is equipped with a diplexer directly at its output or input. At the input or output of this diplexer, all signal combinations are then available in pure form: polarization 1 in the receiving band, polarization 2 in the receiving band, polarization 1 in the sending band and polarization 2 in the sending band.
  • the modules can then be connected to one another by four corresponding waveguide networks. This embodiment has the advantage that the waveguide feed networks do not have to be very broadband in terms of frequency because they only have to be suitable for signals of the receiving or transmitting band.
  • the frequency diplexers are only attached to the input or output of the waveguide networks. Such an embodiment saves installation space, but typically requires a broadband design of the waveguide networks.
  • both the intra-modular microstrip line networks and the inter-modular waveguide networks are designed in such a way that they can support the transmit and receive bands simultaneously.
  • the antenna is provided with frequency diplexers which are connected to a suitable high-frequency switching matrix ("switching matrix"), dynamic switching between the orthogonal polarizations is possible (“polarization switching").
  • switching matrix a suitable high-frequency switching matrix
  • Such embodiments are particularly advantageous when the antenna is to be used in satellite services that work with what is known as “spot beam” technology.
  • spot beam coverage areas (cells) of a relatively small area are created on the earth's surface (typical diameter in the Ka-band approx. 200km -300km).
  • frequency re-use neighboring cells are only differentiated by the polarization of the signals.
  • the antenna When the antenna is used on fast-moving carriers, in particular on aircraft, there are typically very many and very fast cell changes and the antenna must be able to quickly switch the polarization of the received or transmitted signals.
  • the antenna is used in satellite services where the polarization of the received or transmitted signal is fixed and does not change either geographically or temporally, then it is advantageous if the first intra-modular microstrip line network and the associated inter-modular waveguide network are connected to the Reception band of the antenna, and the second intra-modular microstrip line network and the associated inter-modular waveguide network are designed for the transmission band of the antenna system.
  • This embodiment has the advantage that the respective feed networks can be optimized for the respective usable frequency band, and thus a very low-loss antenna system with very high performance is created.
  • the feed networks are equipped with so-called 90 ° hybrid couplers.
  • 90 ° hybrid couplers are four-gates which convert two orthogonal linearly polarized signals into two orthogonal circularly polarized signals or vice versa. With such arrangements it is then possible to also send and receive circularly polarized signals.
  • the antenna field can also be equipped with a so-called polarizer for receiving and transmitting circularly polarized signals.
  • these are suitably structured metallic layers ("layers") which lie in a plane approximately perpendicular to the direction of propagation of the electromagnetic wave.
  • the metallic structure acts in such a way that it acts capacitively in one direction and inductively in the orthogonal direction. For two orthogonally polarized signals, this means that a phase difference is impressed on the two signals. If the phase difference is now set so that it is just 90 ° when passing through the polarizer, then two orthogonal linearly polarized signals are converted into two orthogonal circularly polarized signals or vice versa.
  • the polarizer advantageously consists of several layers which are attached at a certain distance (typically in the region of a quarter wavelength) from one another.
  • a particularly suitable embodiment of the polarizer is a multi-layer meander polarizer.
  • metallic meander structures of suitable dimensions are structured on a typically thin substrate using the usual structuring processes.
  • the substrates structured in this way are then glued to foam panels or laminated to form sandwiches.
  • Low-loss, closed-cell foams such as Rohacell or XPS can be used as foams.
  • very high usable bandwidths and high cross-polarization isolations are achieved if the polarizer is not attached exactly perpendicular to the direction of propagation of the electromagnetic wave in front of the antenna field, but is tilted slightly.
  • the typical distance between the polarizer and the aperture surface of the antenna field is in the range of one wavelength of the useful frequency and the tilt angle with respect to the aperture plane is in the range from 2 ° to 10 °.
  • the antenna pattern of the antenna system in the transmission band must lie under a mask prescribed by the regulations, and with small antennas it can only be transmitted with high spectral power densities if the diagram is as close as possible to the mask
  • aperture amplitude tapering Particularly in the case of flat aperture openings, parabolic amplitude assignments of the aperture are particularly suitable for this purpose. Parabolic amplitude assignments are characterized in that the power contributions of the individual radiators increase from the edge of the antenna field towards the center and z. B. results in a parabolic course.
  • the amplitude occupancy of the antenna system is preferably designed so that it is at least along the direction through the antenna system in which the Radiant elements are close together, works.
  • the radiating elements are close together in the direction in which the distance between the phase centers of the individual radiators is less than or equal to the wavelength of the highest transmission frequency, at which no significant parasitic side lobes (grating lobes) may occur.
  • Fig. 1 represents an exemplary embodiment of an antenna module of an antenna according to the invention.
  • the individual radiators 1 are designed here as rectangular horn radiators which can support two orthogonal polarizations.
  • the intra-modular microstrip line networks 2, 3 for the two orthogonal polarizations are located between different layers.
  • the dimensions of the individual radiators and the size of their aperture areas is selected so that the distance between the phase centers of the individual beam elements along both main axes is less than ⁇ min , where ⁇ min denotes the wavelength of the highest useful frequency. This distance ensures that parasitic sidelobes, so-called “grating lobes", cannot occur in any direction in the antenna diagram up to the highest useful frequency (reference frequency).
  • Both microstrip line networks represent a 64: 1 power splitter, as they combine the signals from 64 individual radiators.
  • An exemplary internal organization of the two microstrip line networks is shown in Fig. 2 shown.
  • the modules comprise a smaller or larger number of horn radiators.
  • K / Ka-band antennas e.g. 4 x 4 modules are optimal.
  • the microstrip line networks then represent a 16: 1 power splitter that combines the signals from 16 individual radiators. In this case, the microstrip lines become relatively short and their noise contribution therefore remains small.
  • an antenna with optimal performance parameters can be constructed by appropriately designing the module sizes.
  • the modules are advantageously made only as large as necessary in order to be able to feed them with waveguides. The parasitic noise contribution of the microstrip lines is thereby minimized.
  • the two microstrip line networks 2, 3 couple the combined signals, each separated according to polarization, in microstrip-to-waveguide couplings 4, 5, as shown in FIG Figure 1b is shown.
  • microstrip-to-waveguide couplings 4, 5 any number of modules can be coupled with the aid of waveguide networks efficiently and with little attenuation to form an antenna system according to the invention.
  • Fig. 2 shows two exemplary microstrip line networks 2, 3 for feeding the individual radiators 1 of the 8 ⁇ 8 antenna module of FIG Fig. 1 . Both networks are designed as binary 64: 1 power dividers.
  • the orthogonally polarized signals are coupled into and out of the individual horn radiators of the 8 ⁇ 8 module.
  • the sum signal is coupled into or out of the waveguide at the waveguide couplings 4a and 5a. Since the two microstrip line networks 2, 3 are typically in two planes one above the other, there are waveguide bushings 4b and 5b also on the corresponding board in order to create a breakthrough and the connection to the waveguide couplings 4a and 5a.
  • microstrip line networks 2, 3 can be produced using all known methods. Low-loss substrates are particularly suitable for antennas.
  • Fig. 3 shows by way of example how different antenna modules 8 can be coupled to form antenna systems according to the invention.
  • Antenna systems according to the invention consist of a number M of modules, where M must be at least two.
  • modules can also be arranged in a circle.
  • not all modules have to be the same size (number of individual radiators).
  • the modules 8 are now networked with one another with the aid of the waveguide networks 9, 10.
  • the corresponding waveguide coupling points 11, 12 of the waveguide networks 9, 10 are connected to the corresponding waveguide couplings 4, 5 (cf. Figure 1b ) of the individual modules 8 connected.
  • the waveguide networks 9, 10 themselves each represent an M: 1 power splitter, so that the two orthogonally polarized signals can be fed into the antenna system via the summing ports 13, 14 or decoupled from the antenna system.
  • waveguide networks 9, 10 can be used for the waveguide networks 9, 10, such as conventional rectangular or round waveguides or broadband, toothed ("ridged") waveguides.
  • Dielectrically filled waveguides are also conceivable.
  • the waveguide network that is directly connected to the waveguide coupling 4, 5 can be advantageous to fill the part of the waveguide network that is directly connected to the waveguide coupling 4, 5 with a dielectric.
  • the dimensions of the dielectrically filled hollow conductors are then reduced considerably, so that their space requirements are minimal.
  • the antenna shown is thus constructed in accordance with claim 1:
  • the antenna consists of an antenna field of N individual radiators 1, each individual radiator 1 being able to support two independent orthogonal polarizations and N designating the total number of individual radiators 1 of the antenna field.
  • the individual radiators 1 are dimensioned in this way (see Fig. 1 ) that for at least one direction through the antenna field the distance between the phase centers of the horn antennae is less than or equal to the wavelength of the highest transmission frequency at which no grating lobes may occur.
  • the individual radiators 1 are fed separately by microstrip lines for each of the two orthogonal polarizations (see Fig. Fig. 2 , Microstrip-to-waveguide couplings 6, 7).
  • microstrip lines of one orthogonal polarization are connected to the first intra-modular microstrip line network 2 and the microstrip lines of the other orthogonal polarization are connected to the second intra-modular microstrip line network 3.
  • the first intra-modular microstrip network 2 is coupled to the first inter-modular waveguide network 9 and the second intra-modular microstrip network 3 is coupled to the second inter-modular waveguide network 10, so that the first inter-modular waveguide network 9 receives all signals from the one orthogonal Polarization at the first sum port 13 and the second inter-modular waveguide network 10 brings together all signals of the other orthogonal polarization at the second sum port 14.
  • microstrip line networks 2, 3 and the waveguide networks 9, 10 are constructed here as complete and completely symmetrical binary trees, so that all individual radiators 1 are fed in parallel.
  • the Figures 3c and 3d show a physical implementation of a corresponding antenna system.
  • the modules 8 consist of individual radiators 1 and have two different sizes, ie the number of individual radiators 1 per module 8 is not the same for all modules 8.
  • the middle four modules 8 each have 8 individual radiators 1 more than the other four modules 8. This means that the height of the antenna system on the left and right edge is lower than in the middle area.
  • Such embodiments are particularly advantageous when the antenna system has to be optimally adapted to an aerodynamic radome.
  • the modules 8 are fed separately with two waveguide networks 9 and 10 for each polarization.
  • the waveguide networks 9, 10 are in two separate layers behind the modules and the modules are connected to the waveguide networks 9, 10 by the coupling points 11, 12, which are coupled to the waveguide couplings of the modules 4, 5.
  • Both waveguide networks 9, 10 are implemented here as millings.
  • the transmit and receive bands of the antenna system are far apart in terms of frequency, the case may arise that the dimensions of the individual radiators 1 of the field must be so small that the lower of the two frequency bands comes close to the limit frequency of the individual radiators 1, or even below.
  • Conventional horn radiators, for example, can then no longer support this frequency band or their efficiency drops sharply.
  • the reception frequency band is approx. 19GHz - 20GHz and the transmission frequency band is approx. 29GHz - 30GHz.
  • the aperture of the individual radiators 1 must be at most 1 cm ⁇ 1 cm ( ⁇ min is 1 cm).
  • the primary individual radiators 1 are designed as toothed ("ridged”) horn radiators.
  • Such horn antennas can be opposite conventional horn radiators have a greatly expanded frequency bandwidth.
  • the impedance matching of such toothed horns to the free space is then carried out using antenna physics methods.
  • the toothed horns can be designed so that they can support two orthogonal polarizations. This is achieved, for example, by the fact that the horns are toothed four times symmetrically ("quad-ridged").
  • the signals of the orthogonal polarizations are fed in and out through separate microstrip line networks 2, 3.
  • FIG 4a shows schematically the detailed structure of a horn antenna equipped with symmetrical geometric constrictions using the example of a quadruple toothed horn antenna 1.
  • the horn antenna 1 consists of three segments (layers), with the two microstrip line networks 2, 3 between the segments.
  • the horn radiators 1 are equipped with symmetrical geometric constrictions 15, 16 corresponding to the orthogonal polarization directions, which extend along the direction of propagation of the electromagnetic wave.
  • Such horns are referred to as "toothed" horns.
  • Is shown in Figure 4a an exemplary quadruple-toothed single horn that can broadly support two orthogonal polarizations.
  • horn antennas 1 can be implemented which can also support transmission and reception bands that are far away in terms of frequency without significant losses in efficiency.
  • An example of this are K / Ka band satellite antennas.
  • the reception band is 18 GHz - 21 GHz and the transmission band 28 GHz - 31 GHz.
  • the depth, width and length of the steps depends on the desired frequency bands and can be determined using numerical simulation methods.
  • the coupling and decoupling of the signals to the microstrip line networks 2, 3 typically takes place at the narrowest point of the constrictions 15, 16 for the respective polarization direction, which allows a very broadband impedance matching.
  • Figure 4d shows schematically part of the longitudinal section through a toothed horn at the location of two opposing constrictions 16.
  • the constrictions 16 are stepped and the distance d i between opposing steps decreases from the aperture of the horn antenna (above) to the horn end (below).
  • the horn itself is stepped (cf. Figures 4a-c ), so that the edge length a i of the horn opening in the corresponding cross section from the aperture of the horn antenna to the horn end also decreases at each step.
  • the distances d i and the associated edge lengths a i , or at least a part thereof, are now designed so that the associated lower limit frequency of the respective toothed waveguide section is below the lowest useful frequency of the horn antenna. Only if this condition is met can the electromagnetic wave of the corresponding wavelength penetrate into the horn antenna up to the waveguide-to-microstrip line coupling, and be coupled in or out there.
  • the distances d i and the associated edge lengths a i are advantageously chosen so that a sufficient distance from the limit frequency remains and the attenuation does not become too high.
  • FIG. 5 the structure according to the invention of a 2 x 2 antenna module is shown schematically, which consists of four quadruple-toothed horn antennas 1, four couplings 17 to the microstrip line networks 2, 3, two microstrip line networks 2, 3, separate for each of the two orthogonal polarizations, and couplings of the microstrip line networks 2, 3 on the waveguide coupling 4, 5 has.
  • the constrictions as symmetrical teeth 15, 16 of the horn antenna 1 are also shown.
  • the two orthogonally polarized signals pol 1 and pol 2 are fed into or extracted from the corresponding microstrip line network 2, 3 through the coupling-out or coupling-in points 17.
  • microstrip line networks 2, 3, in turn, are designed as binary 4: 1 power dividers and couple the sum signals into the waveguides 4, 5.
  • the distance between the phase centers of two adjacent horn antennae 1 in the vertical direction is smaller than ⁇ min , so that at least in this direction in the antenna diagram no unwanted parasitic side lobes ("grating lobes") can occur and the horn antennae are close in this direction.
  • phase centers of the horn radiators 1 fall in the in Fig. 5 shown example with the beam centers of the horn antenna 1 together. In general, however, this is not necessarily the case.
  • the position of the phase center of a horn antenna 1 of any geometry can, however, be determined using numerical simulation methods.
  • microstrip lines are particularly suitable because of their known broadband nature.
  • microstrip lines require very little installation space, so that highly efficient, broadband horn antenna systems whose antenna diagrams do not show any parasitic sidelobes ("grating lobes") can also be implemented for very high frequencies (e.g. 30 GHz - 40 GHz).
  • the antenna modules are constructed from dielectrically filled horn radiators 18.
  • the horn radiators 18 filled with a dielectric 19 are arranged here, for example, in an 8 ⁇ 8 antenna field and are coupled to one another via the microstrip line networks 2 and 3.
  • the microstrip line networks 2, 3 couple the sum signals into the waveguide couplings 4, 5.
  • the dielectric filling body (dielectric) 19 also consists of three segments, which are each defined by the microstrip line networks 2, 3.
  • the individual radiators 1 can support two frequency bands that are far apart, they are designed stepped inside, as shown in the sections Figures 7b-c is shown by way of example.
  • the coupling out or coupling in of the highest frequency band typically takes place at the narrowest or lowest point through the microstrip line network 3, which is furthest from the Aperture opening of the individual radiator 1 is removed.
  • the lower-lying frequency band is coupled out or coupled in at a point further towards the aperture opening by a microstrip line network 2.
  • the depth, width and length of the steps depends on the desired frequency bands and can also be determined here using numerical simulation methods.
  • the horn antenna 1 can, however, also be designed so that both coupling-in and coupling-out points can support both the transmit and receive frequency bands.
  • the dielectric filler body 19 is also designed in a stepped manner correspondingly with a precisely fitting fit.
  • the shape of the filling body 19 on the aperture surface depends on the electromagnetic requirements for the antenna diagram of the individual radiator 1.
  • the filling body 19 can be made flat at the aperture opening, as shown. However, other designs, e.g. curved inwards or outwards, are also possible.
  • a wide variety of known materials such as Teflon, polypropylene, polyethylene, polycarbonate or polymethylpentene can be used as dielectrics.
  • a dielectric with a relative permittivity of about 2 is sufficient (e.g. Teflon, polymethylpentene).
  • the horn antenna 18 is completely filled with a dielectric 19.
  • embodiments with only partial filling are also possible.
  • the advantage of using dielectrically filled horns is that the horns themselves have a much less complex internal structure than in the case of toothed horns.
  • Figure 7d shows the view of the horn from above (plan view) with the aperture edges k 1 and k 2 , as well as the longitudinal sections through the horn antenna along the lines AA 'and B-B'.
  • the horn antenna is now designed so that there is a first rectangular cross section through the horn, the opening of which has a long edge k E , and a second cross section through the horn, the opening of which has a long edge k s .
  • the horn antenna can support the receiving band.
  • edge k s is also chosen so that the associated lower limit frequency of a dielectrically filled waveguide with a long edge k S is below the lowest useful frequency of the transmission band of the antenna system, then the horn can also support the transmission band, and this also applies if the receiving band and broadcast band are far apart.
  • edge k s is orthogonal to the edge k E , two orthogonal linear polarizations are supported simultaneously by such a horn antenna, since the corresponding waveguide modes are linearly polarized and orthogonal to one another.
  • Horn radiators designed in this way can also be operated accordingly without or only with a partial dielectric filling and that the in Figure 7d
  • the illustrated embodiment can be expanded to any number of rectangular horn cross-sections and thus to any number of useful bands.
  • the edge lengths k 1 and k 2 of the rectangular aperture of the horn radiators are chosen so that both k 1 and k 2 are less than or at most equal to the wavelength of the reference frequency, which is in the transmission band of the antenna.
  • the available installation space is then optimally used and a maximum antenna gain is achieved.
  • Fig. 8 shows an exemplary 2 ⁇ 2 antenna module that consists of four dielectrically filled horn radiators 18. As in Fig. 7b-c the coupling-in and coupling-out in the microstrip line networks 2, 3 are shown here completely embedded in the dielectric 19. Otherwise the module does not differ from the corresponding module of toothed horn antennas, as shown in Fig. 5 is shown, the microstrip line networks 2, 3 are each connected to the waveguide couplings 4, 5.
  • a further advantageous embodiment is shown.
  • the module is equipped with a dielectric grating 20 extending over the entire aperture opening.
  • Such dielectric grids 20 can greatly improve the impedance matching, in particular at the lower frequency band of the individual radiators 1, by reducing the effective wavelength in the vicinity of the aperture openings of the individual radiators 1.
  • dielectric crosses are located above the centers of the aperture openings of the individual radiators.
  • other embodiments such as cylinders, spherical bodies, cuboids, etc. are also possible.
  • the dielectric grid 20 does not have to be regular or periodic.
  • the grid for the horn antenna 1 at the edge of the antenna has a different geometry than for the horn antenna 1 in the center. This can be used to model edge effects, for example.
  • Figures 10a-b represents an exemplary module that is built up using layer technology. This technology allows modules according to the invention to be produced particularly inexpensively. In addition, the reproducibility of the modules is guaranteed even at very high frequencies (high tolerance requirements).
  • the first layer consists of an optional polarizer 21, which is used for circularly polarized signals.
  • the polarizer 21 converts linearly polarized signals into circularly polarized ones and vice versa, depending on the polarization of the incident signal. In this way, circularly polarized signals incident on the antenna system are converted into linearly polarized signals so that they can be received by the horn radiators of the module without loss. On the other hand, the linearly polarized signals emitted by the horn antennas are converted into circularly polarized signals and then emitted into the free space.
  • the next two layers form the front part of the horn antenna field, which includes the primary horn structures 22 without an input or output unit.
  • the following layers 23a, 2 and 23b form the coupling and decoupling of the first linear polarization from the horn radiators of the field.
  • the microstrip line network 2 of the first polarization and its substrate are embedded in metallic supports (layers) 23a, 23b.
  • the carriers 23a, 23b have recesses (notches) at the points where a microstrip line runs (cf. also Figure 11d , Reference number 25).
  • microstrip line network 3 of the second, orthogonal polarization is embedded with its substrate in the carriers 23b, 23c.
  • the waveguide terminations 24 of the horn radiators and the waveguide couplings 4 and 5 are.
  • the primary horn structures 22, the carriers 23a-c and waveguide terminations 24 are electrically conductive and can be manufactured cost-effectively with the known methods of metalworking, e.g. from aluminum (e.g. milling, laser cutting, water jet cutting, electrical discharge machining).
  • the layers from plastic materials, which are then completely or partially covered with an electrically conductive layer (e.g. galvanically or chemically).
  • an electrically conductive layer e.g. galvanically or chemically.
  • the known injection molding processes for example, can also be used to produce the plastic layers.
  • Such embodiments have the advantage over layers made of aluminum or other metals that a considerable weight reduction can result, which is particularly advantageous when the antenna system is used on aircraft.
  • the layer technique described can be used in the same way for antenna modules made of toothed horns as well as for modules made of dielectrically filled horns.
  • FIGs 11a-d show the detailed structure of the microstrip line networks 2, 3 embedded in the metallic carrier.
  • the recesses (notches) 25 are designed so that the microstrip lines 26 of the microstrip line networks 2, 3 run in closed metallic cavities. This minimizes microwave losses.
  • the substrates circuit board
  • the substrates are provided with metallic vias 27 on the edges of the notches so that the metallic carrier are galvanically connected, and so the cavities are completely closed electrically. If the plated-through holes 27 lie sufficiently close along the microwave lines 26, then no more microwave power can escape.
  • the plated-through holes 27 preferably end flush with the metallic walls of the cavity 25. If a thin, low-loss substrate (circuit board material) is used, the electromagnetic properties of such a structure are similar to those of an air-filled coaxial line. In particular, a very broadband microwave line is possible and parasitic higher modes cannot propagate. In addition, the tolerance requirements are low even at very high GHz frequencies.
  • the vias can also be dispensed with under certain circumstances, since virtually no microwave power can escape through the very narrow slots even without vias.
  • the horn antenna couplings and outcouplings 6, 7 are integrated directly into the metallic carrier.
  • Fig. 12 shows the vacuum model of an exemplary 8 x 8 antenna module.
  • the horn radiators 1 are tightly packed and yet more than sufficient space remains for the microstrip line networks 2, 3, as well as for the waveguide terminations 28 of the individual radiators 1 and the waveguide couplings 4, 5.
  • a dielectric grating 20 is attached in front of the aperture plane.
  • the waveguide networks which couple the modules to one another are made up of toothed waveguides. This has the advantage that toothed waveguides can have a much larger frequency bandwidth than conventional waveguides or can be designed specifically for different useful bands.
  • FIG. 13 An exemplary network of double-toothed waveguides is shown in Fig. 13 shown schematically.
  • the rectangular Waveguides are provided with symmetrical geometric constrictions 29, which are supplemented by vertical constrictions 30 at the location of the power dividers.
  • the design of the toothed waveguide and the corresponding power splitter can be carried out using the numerical simulation of such components, depending on the requirements of the network.
  • Double-toothed waveguides do not necessarily have to be used.
  • waveguides with single or four teeth are also conceivable.
  • the waveguides of the inter-modular waveguide networks are completely or partially filled with a dielectric.
  • Such fillings can significantly reduce the installation space requirement compared to unfilled waveguides with the same useful frequency. This then results in very compact, space-optimized antennas, which are particularly suitable for applications on aircraft.
  • Both standard waveguides and waveguides with geometric constrictions can be filled with a dielectric.
  • the antenna is equipped with a multilayer meander polarizer.
  • Fig. 14 shows an example of a position of such a polarizer.
  • multilayer meander polarizers are used.
  • Fig. 14 several of the in Fig. 14
  • the layers shown are arranged one above the other in parallel planes. Between the layers there is a low-loss layer of foam material (eg Rohacell, XPS) with a thickness in the range of a quarter of a wavelength.
  • a low-loss layer of foam material eg Rohacell, XPS
  • fewer layers can be used if the demands on the aspect ratio are lower.
  • more layers can be used if the demands on the axis ratio are high.
  • An advantageous arrangement is a 4-layer meander polarizer with which axis ratios of less than 1 dB can be achieved, which is usually sufficient in practice.
  • the design of the meander polarizers is based on the usable frequency bands of the antenna system and can be carried out using methods of numerical simulation of such structures.
  • the meander lines 31 lie in the embodiment of FIG Fig. 14 at an angle of about 45 ° to the main axes of the antenna.
  • the meander polarizer is a linear component
  • the process is reciprocal, i.e. left and right circularly polarized signals are converted into linearly polarized signals in the same way.
  • the polarizer 21 can be attached in front of the aperture opening. It is thus possible in a relatively simple manner to use the antenna both for linearly polarized signals and for circularly polarized signals without the internal structure having to be changed for this purpose.
  • the antenna is equipped with a parabolic amplitude assignment, which is implemented by a corresponding design of the power dividers of the feed networks. Since the antenna diagram must lie under a mask prescribed by the regulations, very much higher maximum permitted spectral EIRP densities can be achieved in transmission mode with such amplitude assignments than without such assignments. This is of great advantage, in particular for antennas with a small aperture area, since the maximum regulatory-compliant spectral EIRP density is directly proportional to the achievable data rate and thus to the costs of a corresponding service.
  • the amplitude occupancy only runs parabolically in the area around the antenna center, but increases again when approaching the edge, so that a closed curve exists around the antenna center and the power contributions of the individual radiators from the center of the antenna to each point of this curve fall off.
  • Such amplitude assignments can be particularly advantageous for non-rectangular antennas.
  • FIG 15b is an example of the maximum regulatory compliant spectral EIRP density (EIRP SD) resulting from a parabolic amplitude occupancy of a rectangular 64 x 20 Ka-band antenna in both main axis directions as a function of the angle of rotation about the main beam axis ("skew"). Without parabolic assignment, the EIRP SD would be around 8 dB lower in the range from 0 ° skew to approx. 55 ° skew and around 4 dB lower in the range from approx. 55 ° skew to approx. 90 ° skew.
  • EIRP SD maximum regulatory compliant spectral EIRP density
  • Fig. 16-18 show the basic structure of a number of antenna systems according to the invention with different functional scope in the form of block diagrams.
  • the antenna system whose basic structure is in Fig. 16 is particularly suitable for applications in the K / Ka band (reception band approx. 19.2GHz -20.2GHz, transmission band approx The polarization direction of the signals does not change).
  • a polarizer 21 is initially provided. This is followed by an antenna array 32, which is constructed either from four-toothed ("quad-ridged") horn radiators or from dielectrically filled horn radiators.
  • the aperture openings of the individual horn antennas typically have dimensions of less than 1 cm x 1 cm in this frequency range.
  • the antenna array 32 is organized in modules, each individual radiator having two microstrip line couplings or couplings 33 separated according to polarization, which in turn are connected separately according to polarizations to form two microstrip line networks 36.
  • the polarization of the transmitted and received signals is fixed and is typically orthogonal to one another Provided here to design the microstrip line network 36 of one polarization on the transmission band and the microstrip line network 36 of the other polarization on the reception band.
  • the polarizer 21 is oriented so that the signals in the transmission band 34 are right-handed circularly and the signals in the reception band 35 are left-handed circularly polarized.
  • the signals of the two microstrip line networks 36 of the individual modules, separated according to polarization and frequency band, are now coupled with microstrip line-to-waveguide couplings 37 in two waveguide networks 38.
  • the two waveguide networks 38 are optimized for the corresponding band that they are intended to support.
  • different waveguide cross-sections can be used for the receive-band waveguide network and the transmit-band waveguide network.
  • enlarged waveguide cross-sections can be used, which can greatly reduce the dissipative losses in the waveguide networks and thus considerably increase the efficiency of the antennas.
  • a reception band frequency filter 39 is provided in order to protect the low-noise reception amplifier, which is typically mounted directly on the reception band output of the antenna, from being overloaded by the strong transmission signals.
  • an optional transmission band filter 40 is also provided. This is necessary, for example, if the transmission band power amplifier (HPA), not shown, does not have a sufficient filter at its output.
  • HPA transmission band power amplifier
  • the illustrated structure of an antenna system according to the invention has a further, very important advantage, particularly for satellite antennas. Since the transmission band feed network and the reception band feed network are completely separated from one another both at the level of the microstrip lines and at the level of the waveguides, it is possible to use different amplitude assignments for the two networks.
  • the receiving band feed network can be assigned homogeneously, i.e. the power contributions of all horn antennae of the antenna are the same in the receiving band and all power dividers on the level of the receiving band microstrip line network as well as on the level of the receiving band waveguide network are symmetrical 3 dB power dividers, if that Feed network is constructed as a complete and completely symmetrical binary tree.
  • the antenna Since homogeneous amplitude assignments lead to the maximum possible antenna gain, it is achieved that the antenna is maximally efficient in the receiving band and the ratio of antenna gain and self-noise G / T of the antenna is maximal.
  • the transmission band feed network can be provided with a parabolic amplitude allocation, independently of the reception band feed network, in such a way that the regulatory-compliant spectral EIRP density becomes maximum.
  • the essential performance characteristics of satellite antennas, especially of small satellite antennas, are the G / T and the maximum regulatory compliant spectral EIRP density.
  • the G / T is directly proportional to the data rate that can be received via the antenna.
  • the maximum regulatory compliant spectral EIRP density is directly proportional to the data rate that can be sent with the antenna.
  • both performance features can be optimized independently of one another.
  • a hyperbolic amplitude occupancy at least in a partial area of the antenna system can thus achieve that the strength of the interference signals received by the antenna from neighboring satellites decreases and the effective G / T increases in such an interference scenario.
  • Fig. 17 the structure of an antenna system according to the invention is shown in the form of a block diagram, which allows simultaneous operation with all four possible polarization combinations of the signals.
  • the antenna system initially consists of an antenna field 41 of broadband, dual polarized horn radiators, for example four-toothed horn radiators, which according to the invention are organized in modules.
  • each horn antenna receives or sends two orthogonal linear polarized signals, which, however, also contain the complete information when operated with circularly polarized signals.
  • All signals 42 of the same polarization are combined after coupling 33 from the antenna field in the first microstrip line network, all signals of the orthogonal polarization 43 in the second microstrip line network.
  • the two microstrip line networks 36 are designed in such a way that they support both the transmission band and the reception band. Optimizing the feed networks on one of the belts is only possible to a limited extent here. However, all four polarization combinations are simultaneously available for this.
  • microstrip line networks 36 are due to their construction (coaxial line-like structure) typically already broadband enough that they can support the receiving and transmitting bands simultaneously, after the transition 37 microstrip-to-waveguide the waveguide networks 44 must do so if very large bandwidths are required specially designed. This can be done, for example, by the in Fig. 13 toothed waveguide described. However, it is also possible to use, for example, dielectrically filled waveguides.
  • Two frequency diplexers 45, 46 are provided to separate the reception band and transmission band signals.
  • the frequency diplexers 45, 46 are, for example, low-loss waveguide diplexers.
  • two 90 ° hybrid couplers 47, 48 When operating with circularly polarized signals, two 90 ° hybrid couplers 47, 48, one for the reception 49 and one for the transmission band 50, are also provided, with the aid of which the linear polarized signals present at the output of the frequency diplexer 45, 46, circularly polarized signals can be combined.
  • the 90 ° hybrid couplers 47, 48 are e.g. low-loss waveguide couplers.
  • the antenna system can also be operated simultaneously with four different linear and four different circularly polarized signals be used. Many other possible combinations and the corresponding antenna configurations are also possible.
  • Fig. 18 the structure of an antenna system according to the invention is shown in the form of a block diagram, which has the same scope of functions as in Fig. 16 antenna shown, but is organized differently.
  • a polarizer 21 is used instead of the 90 ° hybrid coupler 47, 48 according to the structure Fig. 17 .
  • the feed networks 36, 44 again process two orthogonal polarizations separately from one another (here left circular and right circular) and are each designed with a corresponding broadband for the reception band and the transmission band.
  • the four polarization combinations of circularly polarized signals are then directly present at the output of the frequency diplexers 45, 46.
  • the frequency diplexer 45 for the first circular polarization the signal in the receive and transmit band
  • the frequency diplexer 46 for the second (orthogonal to the first) circular polarization the signal in the receive and transmit band.
  • Fig. 18 The advantage of building according to Fig. 18 lies in the fact that no 90 ° hybrid couplers are required when operating with circularly polarized signals. Depending on the application, this can save installation space or weight, for example. Under certain circumstances, there may also be cost advantages.
  • the advantage of building according to Fig. 17 lies in the fact that when operating with circularly polarized signals, the axial ratio of the circularly polarized signals can in principle be freely adjusted via the respective power contributions at the input of the 90 ° hybrid couplers 47, 48.
  • radomes can exhibit polarization anisotropies due to the radome material and the radome curvature, which lead to the axial ratio of circularly polarized signals being greatly changed when they pass through the radome.
  • a structure of the antenna according to Fig. 17 now allows the axial ratio of the circularly polarized signals, e.g. im Transmit mode to be set so that a subsequent polarization distortion caused by the radome passage is compensated. A degradation of the cross-polarization isolation thus effectively does not take place.

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Claims (13)

  1. Système d'antennes, comprenant au moins deux modules (8), dans lequel chaque module (8) contient au moins deux éléments rayonnants à cornet (1), et les éléments rayonnants à cornet (1) prennent en charge une première et une deuxième polarisation, et les deux polarisations sont orthogonales l'une à l'autre, et les modules (8) sont réalisés à partir de différentes couches (22, 23a, 23b, 23c, 24), dans lequel le système d'antennes comprend au moins un réseau de lignes à microrubans (2, 3) et ledit au moins un réseau de lignes à microrubans (2, 3) sert à l'alimentation des éléments rayonnants à cornet (1) à l'intérieur d'un module (8), et dans lequel le système d'antennes comprend au moins un réseau de guides d'ondes creux (9, 10) et ledit au moins un réseau de guides d'ondes creux (9, 10) sert à l'alimentation des modules (8), dans lequel les réseaux de lignes à microrubans (2) présentant une polarisation et les réseaux de lignes à microrubans (3) présentant l'autre polarisation sont disposés séparément les uns des autres entre les couches des modules (8) et les éléments rayonnants à cornet (1) possèdent une alimentation constituée d'une première ligne à microrubans pour la première polarisation et d'une deuxième ligne à microrubans pour la deuxième polarisation, les lignes à microrubans pour la première polarisation et les lignes à microrubans pour la deuxième polarisation forment respectivement des réseaux de lignes à microrubans intramodulaires séparés (2, 3) qui sont respectivement couplés à un réseau de guides d'ondes creux (9, 10), dans lequel le premier réseau de guides d'ondes creux (9) est conçu pour combiner tous les signaux présentant la première polarisation et le deuxième réseau de guides d'ondes creux (10) est conçu pour combiner tous les signaux présentant la deuxième polarisation, dans lequel les réseaux de lignes à microrubans intramodulaires séparés (2, 3) et les réseaux de guides d'ondes creux (9, 10) sont réalisés sous la forme d'arbres binaires et sont conçus pour alimenter en parallèle les éléments rayonnants à cornet (1), dans lequel les éléments rayonnants à cornet (1) sont dimensionnés de telle sorte que, pour au moins quatre éléments rayonnants à cornet (1) directement adjacents, qui sont situés dans au moins deux modules (8) directement adjacents, la distance entre les centres de phase des éléments rayonnants à cornet (1) soit inférieure ou au plus égale à la longueur d'onde d'une fréquence de référence qui se situe dans une bande d'émission du système d'antennes.
  2. Système d'antennes selon la revendication 1, caractérisé en ce que les première et deuxième polarisations sont des polarisations linéaires.
  3. Système d'antennes selon l'une des revendications précédentes, caractérisé en ce qu'une partie des éléments rayonnants à cornet (1) ou la totalité des éléments rayonnants à cornet (1) sont pourvus d'une cloison transversale diélectrique (20) et/ou d'une lentille diélectrique pour réduire la perte d'insertion (S11) dans une bande de réception (35) du système d'antennes.
  4. Système d'antennes selon l'une des revendications précédentes, caractérisé en ce que les réseaux de lignes à microrubans (2, 3) sont situés sur un substrat mince et les lignes à microrubans (26) sont amenées à passer dans des cavités (25) dont les parois sont au moins partiellement électriquement conductrices.
  5. Système d'antennes selon l'une des revendications précédentes, caractérisé en ce que les différentes couches (22, 23a, 23b, 23c, 24) sont constituées de métal, et les lignes à microrubans (26) des réseaux de lignes à microrubans intramodulaires séparés (2, 3) sont amenées à passer dans des cavités qui sont réalisées sous la forme d'encoches (25) dans les couches (23a, 23b, 23c), dans lequel, respectivement, une encoche (25) se situe au-dessus et une autre se situe en-dessous de la ligne à microrubans (26).
  6. Système d'antennes selon la revendication 4, caractérisé en ce que le substrat des lignes à microrubans (26) est pourvu de trous métallisés traversants (27) de manière à établir un contact électrique entre les parois des cavités (25).
  7. Système d'antennes selon l'une des revendications précédentes, caractérisé en ce qu'au moins une partie des réseaux de guides d'ondes creux (9, 10) présente au moins un rétrécissement géométrique (15, 16) dans la direction de propagation de l'onde électromagnétique pour la prise en charge de la largeur de bande requise, de sorte que la largeur de bande utile peut être fortement augmentée.
  8. Système d'antennes selon l'une des revendications précédentes, caractérisé en ce que les réseaux d'alimentation sont pourvus de duplexeurs de fréquence (45, 46) qui sont conçus pour séparer les signaux de la bande d'émission (34) des signaux de la bande de réception (35) afin qu'ils puissent être traités séparément.
  9. Système d'antennes selon l'une des revendications précédentes, caractérisé en ce que les dimensions des lignes à microrubans des réseaux de lignes à microrubans intramodulaires séparés (2, 3) et des guides d'ondes creux des réseaux de guides d'ondes creux (9, 10) sont sélectionnées de manière à prendre en charge à la fois la bande d'émission et la bande de réception (35) du système d'antennes.
  10. Système d'antennes selon l'une des revendications 1 à 8, caractérisé en ce que les dimensions des lignes à microrubans et du guide d'ondes creux sont sélectionnées de manière à ce que le premier réseau de lignes à microrubans intramodulaire (2) et le réseau de guides d'ondes creux intermodulaire associé (9) soient conçus pour la bande de réception (35) du système d'antennes, et à ce que le deuxième réseau de lignes à microrubans intramodulaire (3) et le réseau de guides d'ondes creux intermodulaire (10) associé soient conçus pour la bande d'émission du système d'antennes.
  11. Système d'antennes selon l'une des revendications 1 à 10, présentant une excursion d'amplitude homogène dans la bande de réception (35) de sorte que les contributions de puissance de tous les éléments rayonnants à cornet sont approximativement égales, et présentant une excursion d'amplitude dans la bande d'émission telle qu'au moins pour une partie des éléments rayonnants à cornet (1), les contributions de puissance des éléments rayonnants à cornet (1) augmentent du bord vers le centre du système d'antennes.
  12. Système d'antennes selon l'une des revendications 1 à 10, caractérisé en ce que l'excursion d'amplitude, au moins pour les éléments rayonnants à cornet (1), suit une courbe approximativement parabolique au moins dans la bande de fréquence d'émission, de sorte que la contribution en puissance des éléments rayonnants à cornet (1) situés sur le bord du système d'antennes est inférieure à la contribution en puissance des éléments rayonnants à cornet (1) situés au centre du système d'antennes.
  13. Système d'antennes selon l'une des revendications précédentes, comprenant des coupleurs hybrides à 90° (47, 48) dans les réseaux de guides d'ondes (9, 10), de sorte que des signaux polarisés circulairement peuvent être générés à partir de signaux polarisés linéairement.
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AMENDOLA G ET AL: "Hybrid waveguide–stripline feeding network for dual polarised arrays at K band", IET MICROWAVES, ANTENNAS & PROPAGA, THE INSTITUTION OF ENGINEERING AND TECHNOLOGY, UNITED KINGDOM, vol. 5, no. 13, 21 October 2011 (2011-10-21), pages 1568 - 1575, XP006041966, ISSN: 1751-8725, DOI: 10.1049/IET-MAP.2011.0018 *
KILIC O ET AL: "Flat antenna design considerations for satellite-on-the move and satellite-on-the-pause (SOTM/SOTP) applications", MILCOM 2001. PROCEEDINGS. COMMUNICATIONS FOR NETWORK-CENTRIC OPERATIONS: CREATING THE INFORMATION FORCE. MCLEAN, VA, OCT. 28 - 30, 2001; [IEEE MILITARY COMMUNICATIONS CONFERENCE], NEW YORK, NY : IEEE, US, vol. 2, 28 October 2001 (2001-10-28), pages 790 - 794, XP010579116, ISBN: 978-0-7803-7225-2, DOI: 10.1109/MILCOM.2001.985946 *

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US10211543B2 (en) 2019-02-19
CN104428948A (zh) 2015-03-18
CN104428949B (zh) 2017-05-24
US20150188236A1 (en) 2015-07-02
WO2014005699A1 (fr) 2014-01-09
EP2870659A1 (fr) 2015-05-13
EP2870658B1 (fr) 2019-10-23
US9716321B2 (en) 2017-07-25
EP2870658A1 (fr) 2015-05-13
CN104428950B (zh) 2017-04-12
CN104428948B (zh) 2017-07-11
ES2856068T3 (es) 2021-09-27
WO2014005691A1 (fr) 2014-01-09
CN104428950A (zh) 2015-03-18
ES2763866T3 (es) 2020-06-01
US20150188238A1 (en) 2015-07-02
EP2870660A1 (fr) 2015-05-13
WO2014005693A1 (fr) 2014-01-09
US20150162668A1 (en) 2015-06-11
US9660352B2 (en) 2017-05-23
CN104428949A (zh) 2015-03-18
EP2955788A1 (fr) 2015-12-16

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