JP2000262088A - Synchronous motor drive - Google Patents

Abstract

(57) [Summary] PROBLEM TO BE SOLVED: To provide a low-cost structure by not using a position sensor.
Synchronous motor drive that enables stable and stable driving
Providing equipment. SOLUTION: A CPU 101 has an angular velocity command value ω_m ^*,
Set values of armature resistance and various gains, and γ of armature current
Component i_γ, Δ component i_δTo the synchronous motor 104 based on
The voltage to be applied is calculated and the corresponding command value voltage v_u ^*~
v_w ^*Is output to the inverter device 102. Respond to this
Drive from the inverter device 102 to the synchronous motor 104
Dynamic voltage signal v_u~ V_wIs supplied, and the synchronous motor 104
Threshold voltage v _u ^*~ V_w ^*Is performed.
Drive voltage signal v_v, V_wIs detected by the sensor CT
Current i_v, I_wIs input to the coordinate converter 103. seat
From the target converter 103, the coordinate-converted current i_γ, I_δBut
This is fed back to the CPU 101.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a synchronous motor driving device for driving a synchronous motor.

[0002]

2. Description of the Related Art Synchronous motors have been used in various electric appliances. Compared to induction motors, synchronous motors have features such as small size, light weight, high efficiency, enabling servo lock, and always rotating at a synchronous speed regardless of the size of the load. It is desired that the synchronous motor can be driven and controlled by a simple and simple driving device.

[0003] One of the characteristics required for a synchronous motor is a torque characteristic. In general, when the generated torque and _{τ, τ = n p · Λ} o · i a ( where, n _p: number of pole pairs, lambda _o: the armature according to the field flux linkage, i _a: armature current) in Given, such a linear characteristic becomes an important characteristic in the control system. An inverter-driven synchronous motor that achieves this is a direct current (DC) brushless motor.

[0004]

However, in the conventional DC brushless motor, a complicated position sensor for detecting a relative position between the stator and the rotor is used, which is expensive and requires a signal line. There was also a problem with respect to the reliability of the device, such as the incorporation of noise into the device. In addition to the torque characteristics described above, the synchronous motor must be able to operate smoothly without any turbulence or step-out, be able to start and stop smoothly, and be a system that is robust to constant changes. ,
And other stable characteristics are required.

SUMMARY OF THE INVENTION An object of the present invention is to provide a synchronous motor driving device which can be configured at a low cost without using a position sensor and can drive a synchronous motor stably.

[0006]

SUMMARY OF THE INVENTION A synchronous motor drive according to the present invention.
A control means for outputting a command value signal;
Inver that outputs a drive signal according to the signal to a synchronous motor
Control means, wherein the control means controls the power of the synchronous motor.
Armature resistance is R, armature winding inductance is L, impedance
The angular frequency of the barter means is ω₁, I_δ= Λ_δ/ L (Λ_δ:
The number of flux linkages of the armature δ winding) and the γ-axis component of the armature current
i_γAnd the δ-axis component of the armature current is i_δ, Armature current δ axis
Component command value is i_δ ^*, The current error (i_δ ^*−i_δ)
Feedback gain K_γ, K_δAs v_γ ^*= Ri_γ+ Lω₁I_δ+ K_γ・ (I_δ ^*−i_δ)
/ Ω₁  v_δ ^*= Ri_δ+ K_δ・ (I_δ ^*−i_δThe command value voltage v that satisfies the relationship_γ ^*, V_δ ^*is connected with
Output a signal to the inverter means as the command value signal
It is characterized by doing. The control means includes:_γ ^*= Ri_γ+ Lω₁I_δ+ K_γ・ (I_δ ^*−i_δ)
/ Ω₁  v_δ ^*= Ri_δ+ K_δ・ (I_δ ^*−i_δ) Voltage satisfying the relationship_γ ^*, V_δ ^*Signals related to
It is calculated and output to the inverter means as a command value signal.
The inverter means is configured to drive a drive corresponding to the command value signal.
Supply the signal to the synchronous motor.

Further, the synchronous motor driving device of the present invention includes control means for outputting a command value signal, and inverter means for outputting a drive signal corresponding to the command value signal to the synchronous motor, wherein the control means comprises: with phase difference angle φ is when greater than the predetermined value for controlling said inverter means to decrease the angular frequency omega ₁ of the inverter unit, when the phase difference angle φ is smaller than a predetermined value so as to raise the angular frequency omega ₁ It is characterized in that the inverter means is controlled.

At this time, the control means includes an inverter device.
The angular frequency of the_l, Ω₁ ^*= N_pω _m ^*(Ω_m ^*Is a rotor
Angular velocity command value), n_pIs the number of pole pairs, K_mIs feedback
Gain, T_mIs the time constant of the first-order lag system, and P = d / dt
And the following equation ω_l= Ω_l ^*-K_m・ ｛T_mP / (1 + T_mP)｝ ・ i
_γ  Control the synchronous motor so as to satisfy
Is also good.

The control means may provide a pre-excitation period prior to the start of the synchronous motor and immediately before stopping the synchronous motor, and during the pre-excitation period, v _γ ^* = 0,
i _δ ^* = I _δs (> 0, current of the δ winding during the pre-excitation period), and ω ₁ = 0.

Further, the control means includes: (i)_δ ^*−
i_δ) To increase or decrease the estimated value of armature resistance << R >>
You may comprise so that it may be. Furthermore, the control means
Is i_γAnd i_δOf field strength based on
<< I _o>> may be increased or decreased.

[0011]

Embodiments of the present invention will be described below. In this embodiment, by driving the synchronous motor with a variable frequency / variable voltage (VVVF) inverter, the position sensor can be omitted and the synchronous motor can be driven at a specified angular velocity (inverter angular frequency ω ₁ ). Also, speed fluctuations can be suppressed even when the load increases or decreases. Furthermore, (in the following description with reference to the torque current i _gamma.) Torque τα current i is used primary flux control method to obtain the characteristics of the. That is, a method of controlling the voltage so that the number of flux linkages of the armature becomes a predetermined value is used.

First, control for obtaining a torque τ proportional to the armature current without using a position sensor (without a position sensor) will be described. As shown in FIG. 6, when a balanced three-phase current having an angular frequency ω ₁ is supplied to the winding of an armature (a stator in FIG. 6) 601 of a synchronous motor, an angular velocity ω ₁ / n _p (n _p is the number of pole pairs) Is formed in the gap.

Now, suppose that the rotor 602 is rotating clockwise at the electrical angular velocity ω _me . At this time, the axis of the field of the rotor (d axis: It is known that the torque generated by the synchronous motor is expressed by the following equation (1), assuming that the shaft rotates with the internal phase difference angle δ. τ = n _p · Λ _o · sin δ · i _γ (1) where Λ _o is the number of flux linkages of the armature by the field, and i _γ is the γ-axis component of the armature current. From equation (1), δ is π / 2
It can be seen that the torque can be generated most efficiently for the same current by selecting (at the time of motor operation) or -π / 2 (at the time of regenerative braking). δ is controlled to the above position.)

Next, according to the primary magnetic flux control method,
To explain how it is possible to obtain a torque τ proportional to the torque current i _gamma. Fig. 7 shows the number of primary magnetic flux linkages in the synchronous motor.
FIG. 3 is a diagram in which (hereinafter, referred to as a primary magnetic flux) is drawn based on dq coordinates of a rotor. Here, the q-axis is an axis that rotates 90 degrees clockwise from the d-axis and is orthogonal to the d-axis. In FIG. 7, since the magnetic axis of the field is selected as the d-axis, λ _d
= Λ _o , λ _q = 0. The γ-δ axis is the electrical angular angular velocity ω ₁
, And the coordinate system rotates by an angle φ with respect to the dq axes. Also, the γ-axis component of the armature current is shown as a torque current _iγ .
Note that there is a relationship of φ = (δ−π / 2) between the phase difference angle (the angle between the γ-δ axis and the dq axis) φ and the internal phase difference angle δ in FIG.

Now, as shown in FIG. 7, the γ-axis component and the δ-axis component of the primary magnetic flux and the armature current are λ _1γ , λ _1δ and i _γ , i _δ , respectively. When the δ-axis components are λ _2γ and λ _2δ , respectively, and the inductance of the armature winding is described as L, two pairs of the following formulas (2) are used: λ _1γ = Li _γ + λ _2γ , λ _2γ = −Λ _o sin φ λ _{1δ = Li δ + λ 2δ,} λ 2δ = Λ o cosφ ··· (2) is obtained.

When the various quantities defined above are used, the generated torque τ is given by τ = n _p (λ _1δ i _γ -λ _1γ i _δ ) according to the equations (1) and (2). If the magnetic flux is controlled so as to satisfy a pair of the following equations (4) λ _1γ = 0 λ _1δ = （ _δ (set value) (4), the following equation (5) can be obtained: τ = n _p · Λ _δ · i _γ ··· (5) is satisfied, the synchronous machine can generate a torque that is proportional to the torque current i _gamma irrespective of the rotational speed.

The principle of the primary magnetic flux control method has been described above. In order to control the primary magnetic flux stably and accurately to the value of equation (4), the present embodiment employs the following method. First, λ _1γ / L = λ _γ ′, λ _1δ / L = λ _δ ′, I _δ = Λ
_{When δ} / L and I _o = Λ _o / L are defined, in general, the voltage equation of the synchronous machine is represented by the following equation (6): v _γ = Ri _γ + LPλ _γ ′ + Lω ₁ λ _δ ′ v _δ = Ri _δ + LPλ _δ It is given by the _{'-Lω 1 λ γ' ··· (} 6). However, a P = d / dt (differential operator) Further, as apparent from FIG. 7, a _{λ γ '= i γ -I o} sinφ λ δ' = i δ + I o cosφ.

Incidentally, ideal primary magnetic flux control (λ _γ ′
= 0, λ _δ '= I _δ )
The axis component and the δ-axis component are i _γ ^* , i _δ ^*, and in the above equation, λ _γ ′ = 0 and λ _δ ′ = I _δ are substituted and rearranged.
The following equation (7) is obtained. ^{_{i γ * = I o sinφ i}} δ * = I δ -I o cosφ ··· (7)

To control the primary magnetic flux to the above value, the current
To i_γ= I_γ ^*, I_δ= I_δ ^*(At this time, i_γ ^*, I
_δ ^*Is equivalent to the command current, and “*” is a symbol representing the command value
We use as. ))
It can be realized by doing. Therefore, a pair of the following formulas
Consider applying the voltage represented by (8) to a synchronous machine.
You. v_γ ^*= Ri_γ+ Lω₁I_δ+ K_γ・ (I_δ ^*−i_δ) / Ω₁  v_δ ^*= Ri_δ+ K_δ・ (I_δ ^*−i_δ(8) Note that the third term on the right side of the first equation of equation (8) and the third term on the right side of the second equation
The two terms are current feedback terms and K_γ, K_δIs
Command value current i of δ-axis component of armature current_δ ^*And the actual current
i_δError (i_δ ^*−i_δ) Is the command value voltage v_γ ^*When
v_δ ^*This is the gain at the time of feedback. Also,
The armature resistance R is an estimated value, as will be described later,
Automatic correction is performed to prevent the occurrence of an error due to the error.

Considering the stability of the control according to the equation (8), as will be described later, when the inverter for driving the synchronous machine outputs the command value voltages v _γ ^* and v _δ ^*, , V _γ = v _γ ^* , v _δ = v _δ ^* , so rearranging as equation (6) = equation (8) gives the following equation (9).

[0021]

(Equation 1)

On the other hand, the aforementioned λ_γ’And λ_δ’Relational expression
From <λ_δ’> = Λ_δ'-I_δ  = (I_δ+ I_ocosφ) -I_δ= I_δ− (I_δ-I_ocosφ) <λ_γ’> = Λ_γ'-0 = i_γ-I_osinφ (where “<>” is a symbol representing the deviation
I'm using ), I_δ ^*The ideal primary flux control is performed
Point (<λ_γ’> = <Λ_δ’> = 0)
And the phase difference angle φ is very small, i_γ/ I_o= Sinφ ≒
φ, cosφ ≒ (1-φ^Two/ 2) holds because
(10) holds, and as a result, equation (9) becomes the following equation (1)
It can be expressed as 1). i_δ ^*= I_δ-I_ocosφ ≒ I_δ-I_o+ I_γ ^Two/ 2I_o ... (10)

[0023]

(Equation 2)

From the above equation (11), the characteristic of the magnetic flux control system
The equation is expressed in the time domain as P^Two+ K_δ/ LP + (ω₁ ^Two+ K_γ/ L) = 0, and K_γAnd K_δBy properly choosing₁All of
Performs stable primary magnetic flux control over the entire region (all operating regions)
It becomes possible. For example, even in the low speed range,
Since L is extremely small in a synchronous motor, K_γ/ L and K_δ/
The value of L is very large and ω₁Even if the root changes,
It is possible to perform stable operation without being affected by
You. As described above, the operating point of the synchronous motor
When it deviates from the operating point of the primary magnetic flux control, the effect is i
_δ≠ i_δ ^*And this deviation is represented by v_γ ^*And v_δ ^*To
Feedback provides an ideal operating point
It is possible to return to perform a stable operation.

As described above, the equation (8) is applied to the synchronous machine.
When a voltage that satisfies is satisfied, stable primary magnetic flux control can be performed, and a torque proportional to the armature current can be generated stably without a position sensor. Note that the command value voltages v _γ ^* and v _δ ^* in equation (8) are coordinate-converted into v _u , v _v , and v _w as described later, and are supplied to the synchronous motor via the inverter device.

By the way, the armature resistance R of the equation (8) and the strength I _o (= Λ ₀ / L) of the magnet (field) of the equation (10) are
Both are functions of temperature. For example, in a general synchronous machine, when the temperature of the windings 10 degrees C increases, the armature resistance R is increased by about 5%, on the contrary, the strength lambda _o of the magnet decreases as the temperature rises. Therefore, if errors occur in these,
Errors also occur in λ _γ ′ and λ _δ ′, making it difficult to perform a desired stable operation. Therefore, as described below, by automatically correcting the intensity I ₀ of the armature resistance R and the magnet, to carry out a stable operation.

First, an automatic correction method of the armature resistance R will be described.
To explain, for convenience, the estimated value of the armature resistance R (Equation (8))
Is the estimated value.
Is used as a symbol for ) As R in formula (8)
And considering the steady state, Equations (6) and (8)
Where P = 0 and Equation (6) = Equation (8), Ri_γ+ Lω₁<Λ_δ′> = << R >> i_γ+ K_γ(I_δ ^*
−i_δ) / Ω₁  Holds.

Here, (R-<< R >>) = △ R, <λ _δ '
> ≒ (i _δ −i _δ ^* ), the following equation is obtained. ω ₁ i _γ · {R} (K _γ + Lω ₁ ² ) (i _δ ^* −i _δ ) Therefore, when ω ₁ i _γ > 0, if R><< R >>, then (i _δ ^* −i _δ ) > 0. Therefore, if the current value of the resistor R << R (kT) >> is known, t = kT to (k + 1) T
By using the fact that the value of the resistor R in the next period can be calculated from the current error (i _δ ^* −i _δ ) and the sign of ω ₁ i _γ , the following equation (12) is satisfied. Is automatically corrected.

[0029]

(Equation 3)

Here, sgn (ω ₁ i _γ ) is a sign function, and takes a value of ₁ when ω ₁ i _γ ≧ 0 and a value of −1 when ω ₁ i _γ <0. As described later, a central processing unit (CPU)
Is used to calculate the equation (12), and the cycle T represents a sampling cycle of data input to the CPU. Further, the temperature change of the armature winding is slow, and in the transient state, even if << R >> = R, (i _δ ^* −i _δ ) = 0.
Because it may not be the value of the integral gain K _R is better to select smaller.

Next, the field strength I_o(= Λ₀/ L)
The dynamic correction method will be described. (I) in equation (8)_δ ^*−
i_δ) Is passed through an integrator (see equation (12)).
I in normal state_δ ^*= I_δIt is. Also, <λ_γ’> = I
_γ-I_osinφ ≒ 0, i_δ ^*= I_δ-I_ocosφ
Therefore, if φ is eliminated from these two equations, i_γ ^Two+ (I_δ ^*-I_δ)^Two= I_γ ^Two+ (I_δ-I_δ)^Two=
I_o ^Two  Becomes Therefore, I_oEstimated value << I_o> Is calculated by the following equation (1)
The correction may be made as in 3).

[0032]

(Equation 4)

According to equation (13), the current value << I _o (k
T) >>, the value of Io in the next period can be calculated. Also in this case, the change of _Io is gradual, and in the transient state, << _Io >>
Even if = I _o , the integral term in the above equation may not be 0, so it is better to select the integral gain K _I to a small value. That is, as shown in equation (13), when the synchronous motor operates near the ideal operating point, the error of _<< I _o >> is i _γ
And i _δ . Therefore, when << I _o >><I _o , i _γ ² + (i _δ −I _δ ) ² > I _o ² holds, so (i
^{_{γ 2 + (i δ -I δ}} ) 2) = until I _o ^2, may be controlled to increase the gradually "I _o" by choosing smaller K _I. In equation (13), the squared value is integrated. Since the integral term does not include a sine function, a cosine function, a square root, or the like, the equation (13) can be used without imposing a large load on the CPU. 13) It is possible to perform the arithmetic processing.

Next, a description will be given of a method of controlling a synchronous motor stably without any irregularity in the rotational speed without using a sensor for detecting the rotational speed. As is well known, since the synchronous machine can rotate only at the synchronous speed, the phenomenon that the internal phase difference angle δ in FIG. 6 (phase difference angle φ in FIG. 7) fluctuates when the load fluctuates or the like, that is, turbulence occurs. Further, if the turbulence is remarkable, a step-out occurs, and there is a risk that the motor loses its rotational force and stops suddenly.

[0035] In order to solve this problem, in FIG. 7, when phase difference angle phi is larger than a predetermined value (i γ ₌ I _o sinφ phase angle satisfying the phi) is the corner of the inverter device frequency omega ₁ Should be slightly lower than the current value. dφ /
dt = ω _l -n _p ω _m or _{φ = ∫ (ω l -n p} ω m)
Therefore, the phase difference angle φ gradually decreases and shifts to a predetermined value. Conversely, when the phase difference angle φ is small, ω ₁ may be increased.

Specifically, when the phase difference angle φ is small, in the above equation i _γ = I _o sin φ, φ is set to s
Assuming that the approximation of inφ ≒ φ is small, i
_γ ≒ I _o φ, and i _γ is proportional to φ. From this, by multiplying i _γ by K _m instead of the phase difference angle φ and feeding it back to ω ₁ , the angular frequency ω _l
give. Furthermore, in order to stabilize the system to prevent _hunting, it adds a K _m T _m P / term (1 _{+ T} m P) as a feedback term, the angular frequency omega _l of the inverter device, satisfies the following expression (14) Give to give.

[0037]

(Equation 5)

Where ω ₁ ^* = n _p ω _m ^* , ω _m ^* is the angular velocity command value of the rotor, n _p is the number of pole pairs, K _m is the feedback gain, T _m is the time constant of the first-order lag system, P = d / dt. As shown in equation (14), the angular frequency ω _l of the inverter device
In the steady state where i _γ is constant,
Since the time derivative Pi _γ of i _γ becomes 0, ω ₁ = ω ₁ ^* , and the synchronous machine rotates at the angular velocity according to the command value regardless of the load.

Further, as in the case of acceleration / deceleration or a sudden change in load, ω
₁And i_γIs rapidly changing, T_mP / (1 + T_mP) T to the extent that ≒ 1 holds_mEquation (14) becomes
become that way. ω₁≒ ω₁ ^*-K_mi_γ  Thus, for example, ω_l ^*Is increased, dφ
/ Dt> 0 and φ increases, but i increases in proportion to φ._γ
Also increases, so ω_lNegative feedback works to suppress the increase in
Therefore, stable operation can be achieved. Yo
Therefore, based on Expression (14), ω₁By controlling
This allows stable operation without disturbance,
It is possible to rotate the synchronous motor at the same speed
You.

Next, a control method for improving the starting characteristics of the synchronous machine will be described. In a synchronous sensor driving device without a position sensor, since φ (= φ _S ) at the time of stop cannot be determined, the synchronous motor may temporarily rotate in the opposite direction to ω _m ^* . That is, when the synchronous motor is stopped, the positional relationship between the rotor and the stator (δ in FIG. 6 or φ in FIG. 7) cannot be detected. If i _γ (> 0) flows in the state of δ <0, the rotor rotates counterclockwise and stops at the position of δ ≒ 0. Thereafter, δ gradually increases with the rise of ω ₁ , and the torque τ increases accordingly (see equation (1)). The rotor rotates in the normal clockwise direction. In addition, it is necessary to devise a method of avoiding a reversal phenomenon at the beginning of starting which is not preferable for practical use.

When the synchronous motor has δ = π / 2 (φ = 0),
Generates the largest torque. Therefore, in a state where φ is set at this position, it is preferable to start the _engine by supplying the current _iγ . For this purpose, the current (excitation current) i _δ of the δ winding at a position perpendicular to the γ winding is utilized.

To explain this in detail, the generated torque τ of the synchronous machine is λ _1γ (= Lλ _γ ′), λ _1δ (= Lλ _δ ′)
By substituting Equation (2) into Equation (2), it can be expressed as Equation (15) below. _{τ = n p L (λ δ} 'i γ -λ γ' i δ) = n p LI O (cosφ · i γ + sinφ · i δ) ··· (15) phase angle at the time of start-up φ (= φ _s) since is optional, it is necessary to shift the phi _S by some method to the position of normal operation angle (φ ≒ 0).

Therefore, the pre-excitation period is set before starting.
And during that period, v_γ ^*= 0, i_δ ^*= I_δs(> 0), ω₁= 0. Where I_δSIs the δ winding at the phase difference angle φ
Current (excitation current). During this period, the synchronous motor
Works like That is, λ_γ’= I_γ-I_osinφ_s  n_pω_m+ Dφ / dt = ω₁= 0, and from equation (6), v_γ= Ri_γ+ LPλ_γ'= (R + LP) i_γ+ N_pL
I_ocosφ_s・ Ω_m  Holds.

Here, assuming that v _γ = v _γ ^* = 0 and the LPi _γ term in the above equation is smaller than the others (P ≒ 0) and is ignored, the following equation (16) is obtained. _{cosφ s · i γ ≒ -n p} LI o cos 2 φ s / R · ω m ··· (16) formula As can be seen from (16), the first of the right-hand side of equation (15) 1
The torque by the term always acts in the direction opposite to ω _m (braking force).

For example, if the pre-excitation is performed in the state of 0 <φ _s ≦ π, since ω _m = 0 at the beginning of the pre-excitation, only the torque of the second term on the right side of the equation (15) acts. The synchronous motor rotates in the forward direction (ω _m > 0). If ω _m rises during this time, a braking force by i _γ acts, so that the synchronous motor shifts φ _s to 0 in an over-braking state. Also, 0> φ _s >
In the case of -π, the torque acts in the opposite direction to shift φ _s to 0, and the pre-excitation ends when φ _S becomes 0.
Then, given the angular frequency omega ₁ of the inverter device as in equation (14) when this operation has been completed, may be accelerated to a desired value.

As described above, in this embodiment, the synchronous motor is started from the best position using the pre-excitation. At this time, since v _γ = 0 (the γ winding is equivalently short-circuited), i _γ according to ω _m flows, and the rotor oscillates toward φ _S = 0. Transitions smoothly.

On the other hand, when stopping the synchronous motor, ω ₁
^{* The} above-mentioned pre-excitation period is provided from the time when ≒ 0. By this operation, the synchronous motor can be reliably stopped near φ _s = 0. Since the synchronous motor has a statically stable operating point at φ _s = 0, φ _s ≒ 0 at the next start-up
Starting from is possible. By performing the above operation, there is a risk that the rotor rotates in the reverse direction only once at the beginning during the pre-excitation period, and the rotation of the rotor in the reverse direction can be prevented.

FIG. 1 is a block diagram showing an embodiment of the present invention.
In the figure, a synchronous motor drive for realizing each of the above control operations is shown.
It is a block diagram of an apparatus. In FIG. 1, the calculation means
The central processing unit (CPU) 101 has an angular velocity command value.
ω_m ^*And the set value are input. As the setting value,
Armature resistance R, field strength I_o, Armature winding inductor
Tance L, δ-axis component LI of armature flux linkage number _δ, Pole pairs
n_p, Various gains K_γ, K_δ, K_m, K_R, K_I, Time constant T
_mThere is. However, armature resistance R and field strength I_oIs the temperature
Therefore, a nominal value is given. Also,
The CPU 101 has a coordinate converter 1 as coordinate conversion means.
03 to i_γ, I_δIs entered. CPU101 and coordinates
Converter 103 constitutes control means. Each of the above
Gain K_γ, K_δ, K_m, K_R, K_IAnd time constant T_mIs control
It can be selected as appropriate according to the goals, etc.
Rated output is 2kW and rated voltage is 115V, 133Hz
For a 2,000 rpm synchronous motor, K_γAs
1.070 [V / AS], K_δ2.60 [V /
A], K_m3.60 [1 / AS] K_RAs 0.
15 [V / A²S], K_I0.03 [1 / A
S], T_mValue of about 0.10 [S] can be used
You.

From the CPU 101, the three-phase command value voltage v
u *, v v *, v with w * is output to the inverter device 102 to be described later, theta 1 with respect to the coordinate converter 103 is outputted. Inverter device 10 as inverter means
2, the three-phase drive signal v
u, v v, with v w is supplied, the motor current iv of this time, iw is detected by the sensor CT, it is outputted to the coordinate converter 103. Coordinate converter 103, the detection current i v, and i w by coordinate transformation, the corresponding current i gamma, the i [delta] and outputs it to the CPU 101.

FIG. 2 is a flowchart showing the processing of the CPU 101.
It is a chart. Hereinafter, this embodiment will be described with reference to FIGS.
The operation of the embodiment will be described. First, the CPU 101_m
^*, I_γ, I_δ(Step S201), and ω_m
^*And i_γFrom ω_l ^*(= N_pω_m ^*) Is calculated (step
Step S202). ω_m ^*And ω_l ^*Determine if is zero
(Step S203), ω_m ^*And ω_l ^*Is zero
In the case (at the time of starting), a pre-excitation period is provided and v_γ
^*= 0, ω _l= 0, v_δ ^*= Ri_δ+ K_δ・ (I_δS−
i_δ) And the process proceeds to step S207.

On the other hand, in step S203, ω_m ^*
And ω_l ^*If is not zero (during operation), the expression
Using (8), (10), and (14), ω_l, I_δ ^*,
v_γ ^*, V_δ ^*Is calculated (step S204). Next
And Δω_l ^*= 0 is determined (step S20).
5), Δω_l ^*If not = 0, go to step S207
Transition, Δω_l ^*= 0 and << I >>₀》 Automatic
After performing the correction, the process proceeds to step S207 (step S207).
S206). In step S207, θ_l= ∫ω
_ldt is calculated, converted to a binary value, and output to the coordinate converter 103.
Power.

Further, v _γ ^* and v _δ ^* are _subjected to coordinate transformation to obtain v
_u to v _w to calculate the (voltage to be applied to the synchronous machine), the command value voltage v _u of the inverter device 102 ^*, _v ^v *, v seek _w ^* outputted to the inverter device 102 (step S20
9). Thereafter, the above operation is repeated.

Incidentally, as shown in FIG.
Is used to perform coordinate conversion from i _u , i _v , i _w to i _γ , i _{δ in order} to convert an AC transient into a DC so as to be treated as a DC transient. This not only greatly simplifies the handling of expressions,
The handling of the control system becomes convenient. Also, i _u , i _v , i _w
A coordinate converter 103 having a hardware configuration is used for the coordinate conversion from i to i _γ and i _δ .
_γ and i _δ are DC, and the PWM of the inverter device 102
This is because there is an advantage that current ripple due to control can be easily removed.

The command value voltage v _u obtained as described above
^{_{*, V v *, v w}} * is is output to the inverter device 102, theta ₁ is outputted to the coordinate converter 103, the inverter device 102, the command value voltage ^{_{v u *, v v *,}} v w *
To the synchronous motor 104 based on the three-phase drive signal v _u ,
v _v , v _w are issued. Thereby, the synchronous motor 104 is driven stably.

Next, a command value for realizing each of the above operations will be described.
Explains an example of an inverter device that can output different drive signals
I do. FIG. 3 shows details of the inverter device 102 of FIG.
In the circuit diagram, one of the three phases of the inverter device 102 is shown.
FIG. 4 shows a circuit diagram of a minute (u phase). In FIG.
Not charged by rectifier circuit, make up DC power supply
The capacitor 301 includes a transistor T
_u ⁺And transistor T_u ⁻They are connected in series. Tiger
Transistor T_u ⁺Constitutes first switching means, and
Transistor T_u ⁻Constitutes the second switching means.
You.

[0056] transistor _T ^{u +} collector - the emitter is connected the feedback diode _D ^{u +} Further, the transistor _{T u} ^- collector - emitter between the feedback diode D
_u ^- are connected. Transistor T _u ⁺ emitter of the transistor T _u ^- the connection point of the collector together is connected to a load, it constitutes a voltage divider circuit resistors R ₁ and R
₂ in series.

On the other hand, the command value voltage v from the CPU 101_u
^*Constitutes a differential signal output means via the multiplication circuit 306
Command value is input to the positive input section of the subtraction circuit (addition point) 303
η_u ^*Is input.
Further, a resistor R is provided at a negative input portion of the subtraction circuit 303.₁And resistance R
₂Are connected, whereby the subtraction circuit 303
The output potential v_uIs the resistance R₁And R₂With partial pressure
Potential η_uIs entered.

The output of the subtraction circuit 303 is connected to the input of an integration circuit 304 as storage means. The output of the integrating circuit 304 is connected to the input of a hysteresis comparator circuit 305 as a hysteresis comparator means. Hysteresis comparator circuit 3
05 compares the output voltage e of the integration circuit 304 with a predetermined threshold voltage, and outputs an H level or L level control signal S according to the comparison result.

The output of the hysteresis comparator circuit 305 has a well-known delay circuit 302 as a delay means for delaying the rise of the control signal S by _Td.
Is connected. From the delay circuit 302, the drive signal is a signal delayed by a rising of the T _d of the control signal S is the transistor T _{u ^+,} T _u ^- is configured to be supplied to the bases of.

FIG. 4 is a diagram for explaining the operation principle of the inverter device 102 shown in FIG.
In contrast, the temporal change of the output voltage e of FIG. 04 differs depending on the presence or absence of the delay circuit 302. The case where the delay circuit 302 is not provided, that is, the case where T _d = 0 where the control signal S and the change timing of the drive signal coincide, is indicated by a solid line. When the delay circuit 302 is provided, that is, the rise of the drive signal is controlled. The case where the signal is delayed by _Td from the signal S is indicated by a broken line.

Hereinafter, the operation of the inverter device 102 will be described with reference to FIGS. The output potential V _u is _divided by the resistors R ₁ and R ₂ , and as a signal related to the output, the potential η _u (= Kv _u , where K = R ₂ / (R ₁ + R)
₂ )) is input to the negative input of the subtraction circuit 303. From the subtraction circuit 303, the potential η _u and the command value η _u ^* (= Kv _u
^* ) A difference signal corresponding to the difference is supplied to the integration circuit 304. The difference signal is time-integrated by an integration circuit 304, and an output voltage e expressed by the following equation (17) is output to a hysteresis comparator circuit 305.

[0062]

(Equation 6)

The output voltage e is compared with a predetermined threshold value by a hysteresis comparator circuit 305.
As will be described later, a control signal S of H level or L level is output according to the comparison result. The control signal S is the control signal S is controlled to turn on the transistor T _u ⁺ at H level, when the L level, the transistor T _u ^- is a signal for controlling the on to.

When T _d = 0 where no delay circuit 302 is provided, a drive signal having the same waveform as the control signal S is supplied to the transistors _Tu ⁺ and _Tu ⁻ . Accordingly, as soon as the output of the hysteresis comparator circuit 305 changes, the output potential v _u changes, and in response to this, the output voltage e of the integration circuit 304 changes, so that the output voltage e becomes as shown by the solid line in FIG. It changes between the first threshold value ΔH of the hysteresis comparator circuit 305 and the second threshold value −ΔH. Therefore, assuming that the time of one cycle of the output voltage e is T,
The following equation (18) holds.

[0065]

(Equation 7)

Since the cycle of the command value v _u ^* is sufficiently larger than the cycle of the output voltage e, control satisfying the following equation (19) can be realized with practically sufficient accuracy.

[0067]

(Equation 8)

[0068] On the other hand, when a delay circuit 302 as shown in FIG. 3, after the rise of the control signal S is delayed by _{T d,} Totanjisuta _T ^u as a drive signal +, _{T u}
⁻ , The transistors _Tu ⁺ , _Tu ⁻
It turns on with a delay of _Td from the control signal S.
In this case, when the load current i _u is positive, the transistor _Tu
^{Since +} is turned on with a delay of _{Td, the} output voltage e operates beyond the threshold value ΔH by this delay as shown by the broken line in FIG. However, when the load current i _u is positive, the output voltage e does not drop below −ΔH even if the turning on of the transistor _Tu ⁻ is delayed by _Td because the feedback diode _Du ⁻ turns on first. . Note that when the load current i _u is negative, FIG. 4 is a diagram that is inverted upside down around the time axis t.

By the way, the transistor T_u ⁺, T_u ⁻,
Feedback diode D_u ⁺, D_u ⁻Ignore the voltage drop of
If the instantaneous potential of the output terminal is_u ⁺But
V during the period of_dConstant, transistor T_u ⁻Or da
Iod D_u ⁻Is 0 and constant during the ON period. did
Therefore, the output potential v_uOf the periods T and T '_uIs T_dWhen = 0: E_u= (Τ + / T) · V_d, T_dWhen is provided: E_u= (Τ + '/ T') · V_d  Becomes

The period T or T 'is extremely small as compared with the period of the command value v _u ^* , and the change of the output voltage e during this period can be regarded as a straight line, so that τ + / T = τ +' / T 'is sufficiently accurate for practical use. It can hold. That is, even if there is a time delay due to _Td , the integration circuit 304 stores the voltage error between the potential η _u and the command value η _u ^* during this time, and the transistor _Tu ⁺ is used to compensate for this in the next cycle. , T _u ^- becomes the oN time or oFF time to automatically adjust the command value v _u
An output Eu corresponding to ^* is obtained. Therefore, the command value v _u ^*
Can be obtained as an output v _u of a PWM waveform corresponding to.
In the above description, the voltage drop of the transistors _Tu ⁺ and _Tu ⁻ and the feedback diodes _Du ⁺ and _Du ⁻ and the time delay of the operation are ignored. However, these are added to the potential η _u. Therefore, it is possible to obtain the output as the command value v _u ^* .

Next, the overall operation of the inverter device shown in FIG. 3 will be described with reference to the timing chart of FIG. FIG. 5 shows a command value η _u ^* , a potential η _u , an output voltage e of the integration circuit 304, a control signal S, and transistors _Tu ⁺ and _Tu ^−.
3 shows on / off timings of. The command value v
_u ^* and the output v _u are signals obtained by multiplying the command value η _u ^* and the potential η _u by 1 / K, and have waveforms that change at the same timing as these. FIG. 5 shows a case where the load current i _u is positive.

At time T ₁ , when the output voltage e becomes equal to the threshold value −ΔH, the hysteresis comparator circuit 305 detects this and outputs an L level control signal S. Thus, transistor T _u ⁺ off, 0 and will also potential eta _u, output voltage e starts increasing. T _d
After the transistor T _u ^- is turned on. Output voltage e
Reaches the threshold value ΔH, the hysteresis comparator circuit 305 detects this, and the H level control signal S
Is output. Thus, transistor _{T u} ^- it is turned off. The output voltage e continues to increase thereafter.

[0073] If after _{T d} elapses transistor _T ^{u +} is turned on, the potential eta _u is also becomes KV _d, the output voltage e starts decreasing. When the output voltage e reaches the threshold value-△ H,
The hysteresis comparator circuit 305 detects this and outputs an L level control signal S. Thus, transistor T _u ⁺ off, 0 and will also potential eta _u, output voltage e starts increasing. Thereafter, this operation is repeated.

At this time, the output voltage e is obtained by integrating (η _u ^* −η _u ) with respect to time as shown by the equation (18), and the gradient (de / dt) is (η _u ^* − η _u ), and becomes steeper as the potential difference between the command value η _u ^* and the instantaneous value of the potential η _u becomes larger. Therefore, as the potential difference becomes larger, the pulse width of the control signal S becomes narrower, and the time during which the transistor _Tu ⁺ is turned on also becomes shorter. Accordingly, the pulse width of the potential η _u also becomes narrow, and a potential η _u having a narrow pulse width corresponding to the command value η _u ^* is obtained. Conversely, the smaller the potential difference, the wider the pulse width of the potential η _u and the command value η _u
A potential η _u having a wide pulse width corresponding to ^* is obtained.

[0075] Mean value _{E u} of the period T obtained by repeating the above operation, as shown in equation (20) becomes equal to the command value ^{_{v u * (= η u *}} / K), as a result, the command Output v _{u of the} PWM waveform equal to the value v _u ^* (= η _u / K)
Is obtained. Therefore, with a simple configuration in which a storage means such as an integration circuit is provided, it is possible to compensate for the influence of the arm short-circuit prevention time, the voltage drop of the element, and the like, and obtain an output according to the command value.

In order to store the voltage error between the potential η _u , which is a signal related to the output, and the command value η _u ^* , the integration circuit 30
4 is used, but when other signals are digitally processed, for example, a digital storage device such as a random access memory (RAM) is used to store the voltage error caused by _Td , and The compensation may be made in cycles. Further, although the voltage is divided using the resistors R ₁ and R ₂ in order to generate the potential η _u , various changes are possible, such as voltage division using a plurality of capacitors connected in series. Further, although the potential η _u is used as a signal related to output, the output potential v _u may be directly used as the signal. Further, although a transistor is used as the switching means, another switching element such as an IGBT or a thyristor can be used.

As described above, according to the present embodiment,
Since a position sensor can be used, a low-cost configuration can be realized. Further, it is possible to stably generate a torque proportional to the armature current. Further, it is possible to perform a stable operation without any disturbance. Further, it is possible to start and stop satisfactorily. Further, even if the armature resistance and the strength of the field change, a stable operation can be performed by performing automatic correction.

[0078]

According to the present invention, the position sensor can be eliminated, so that the configuration can be made at low cost.
There is an effect that stable driving becomes possible.

[Brief description of the drawings]

FIG. 1 is a block diagram according to an embodiment of the present invention.

FIG. 2 is a flowchart for explaining an operation according to the embodiment of the present invention.

FIG. 3 is a block diagram of an inverter device used in the embodiment of the present invention.

FIG. 4 is a timing chart for explaining an operation of the inverter device used in the embodiment of the present invention.

FIG. 5 is a timing chart for explaining an operation of the inverter device used in the embodiment of the present invention.

FIG. 6 is a diagram for explaining an operation of the exemplary embodiment of the present invention.

FIG. 7 is a diagram for explaining an operation of the exemplary embodiment of the present invention.

[Explanation of symbols]

 101: CPU constituting control means 102: Inverter device as inverter means 103: Coordinate converter constituting control means 104: Synchronous motor

 ──────────────────────────────────────────────────続 き Continuing on the front page (72) Minoru Senba 3-1-1, Yoshino-cho, Nishibiwajima-cho, Nishi-Kasugai-gun, Aichi Prefecture Inside the Technical Development Center of Takatake Seisakusho Co., Ltd. No. 1-1, Takahama Works Hamamatsu Techno Center F-term (reference) 5H576 BB10 DD05 EE01 EE15 EE18 FF01 FF05 GG04 GG05 HB02 JJ03 JJ04 JJ09 JJ22 LL22 LL24 LL40 MM12

Claims (6)

[Claims] A control means for outputting a command value signal;
Outputs a drive signal corresponding to the command value signal to the synchronous motor.
Converter means, wherein the control means comprises:
The armature resistance of the machine is R, and the inductance of the armature winding is
L, the angular frequency of the inverter means is ω₁, I_δ= Λ_δ/ L
(Λ_δ: Δ-axis component of the number of magnetic flux linkages of the armature caused by the field)
Let the γ-axis component of the armature current be i_γAnd the δ-axis component of the armature current
i_δThe command value of the δ-axis component of the armature current_δ ^*, Incorrect current
Difference (i_δ ^*−i_δ) Feedback gain_γ, K
_δAs v_γ ^*= Ri_γ+ Lω₁I_δ+ K_γ・ (I_δ ^*−i_δ)
/ Ω₁  v_δ ^*= Ri_δ+ K_δ・ (I_δ ^*−i_δThe command value voltage v that satisfies the relationship_γ ^*, V_δ ^*is connected with
Output a signal to the inverter means as the command value signal
A synchronous motor driving device. 2. A control device for outputting a command value signal; and inverter means for outputting a drive signal corresponding to the command value signal to a synchronous motor, wherein the control device is configured to control when a phase difference angle φ is larger than a predetermined value. Is the angular frequency ω of the inverter means
Controls said inverter means to lower the _1, wherein when phase difference angle φ is smaller than a predetermined value and controls the inverter unit to raise the angular frequency omega ₁ synchronous motor driving device. 3. The control means sets the angular frequency of the inverter device to ω ₁ , ω ₁ ^* = n _p ω _m ^* (where ω _m ^* is the angular velocity command value of the rotor), n _p is the number of pole pairs, and K _m is Feedback gain,
3. The synchronous motor driving device according to claim 2, wherein the synchronous motor is controlled such that _Tm is a time constant of a first-order lag system and P = d / dt so as to satisfy the following expression. ω ₁ = ω ₁ ^* −K _m · {T _m P / (1 + T _m P)} · i
_γ 4. The control means provides a pre-excitation period before starting the synchronous motor and immediately before stopping the synchronous motor, and v _γ ^* = 0, i _δ during the pre-excitation period.
^{_* =} I ^δs (> 0, the current of δ winding in pre-excitation period), ω ₁ = 0, by controlling the synchronous motor drive system of claim 1, 2 or 3 wherein. 5. The synchronization according to claim 1, wherein the control means increases or decreases the estimated value << R >> of the armature resistance based on (i _δ ^* −i _δ ). Motor drive. 6. The control unit according to claim 1, wherein the control unit increases or decreases the estimated value of the field strength << I _o >> based on i _γ and i _δ. Synchronous motor drive.