JPH0131835B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION Field of the Invention The present invention relates to a television synchronization receiver that can be used in television receivers and VTR video tuners.

Conventional configurations and their problems In recent years, so-called electronic tuners that use variable capacitance diodes as tuning elements have been widely used in television receivers and VTR video tuners. Electronic tuners have advantages such as being non-contact, so there is no problem of contact failure, and being electronically controllable, making it convenient for multi-functions such as remote control. However, due to variations in the characteristics of variable capacitance diodes and the need for an inductor for tuning, it is difficult to manufacture them without adjustment or to automate their production.

Therefore, in order to configure a receiver that is easy to integrate without using a tuning circuit using a variable capacitance diode and an inductor, it is possible to use a synchronous reception method. Although there are various synchronous reception methods, a synchronous carrier regeneration method is suitable for phase-synchronizing a synchronous carrier wave with a weak television signal. This method is called Costas loop.
known as the method.

FIG. 1 is a main part block diagram showing the configuration of a conventional synchronous carrier regeneration type synchronous receiver using a Costas loop. 1 is a first synchronous detector that synchronously detects the in-phase component of a modulated carrier input; 2 is a second synchronous detector that synchronously detects a quadrature component; 3 and 4 are each of these two synchronous detectors 1 and 2; A low-frequency wave generator that outputs a low-frequency wave, 5 indicates these two low-frequency wave generators 3
and a phase detector that detects the phase of the synchronized carrier wave with respect to the modulated carrier wave by multiplying the outputs of 4 and 4 by voltage, 6 is a low-pass filter that converts the output of this phase detector into a low-pass filter, and 7 is this low-pass filter 6 The voltage controlled oscillator 8 is controlled by the output of the voltage controlled oscillator 7, and 8 is a 90° phase shifter that shifts the phase of the output of the voltage controlled oscillator 7 by 90°.

In this Costas loop type synchronous receiver, the first
The signals of the in-phase and quadrature components obtained from the second synchronous detectors 1 and 2 are applied to the phase detector 5, and from this phase detector 5, the modulated carrier wave which is the receiver input and the synchronous signal which is the output of the voltage controlled oscillator 7 are applied to the phase detector 5. By obtaining a voltage proportional to the phase error with the carrier wave and feeding this voltage back to the voltage controlled oscillator 7, the phase error is controlled to be zero.

If the conventional example shown in FIG. 1 is applied as is to a television receiver, the output of the low frequency filter 6 will contain noise components because the television signal has residual sideband characteristics. This noise component uses a voltage controlled oscillator to phase modulate the synchronous carrier wave and frequency modulate the audio intermediate frequency signal. Therefore, demodulated video and audio signals are disturbed by noise components.

OBJECTS OF THE INVENTION It is an object of the present invention to provide a television synchronization receiver capable of removing interference caused by noise components remaining in the output of a low-band phase detector on demodulated video and audio signals. It is in.

Structure of the Invention In order to achieve the above object, the present invention provides a voltage controlled oscillator and a phase changer of the output of the voltage controlled oscillator.
A 90° phase shifter that shifts the phase by 90°, the output of the voltage controlled oscillator, and the output of the 90° phase shifter are respectively synchronous carrier waves, and these two synchronous carrier waves are inputted from the receiver input section. First and second synchronous detectors detect the in-phase and quadrature components of the video carrier signal, and the outputs of the first and second synchronous detectors are used to detect low frequency components at the video signal baseband and audio intermediate frequency signals. first and second low frequency filters for converting the frequency portions having both sideband characteristics in the residual sidebands of the television signals output from the first and second low frequency filters to low frequency waveforms; 3rd and 4th
a phase detector that detects the phase difference between the video carrier wave and the synchronous carrier wave from the outputs of the third and fourth low-frequency wave generators, and the output of this phase detector is fed back to the voltage controlled oscillator. This makes it possible to remove noise components from the control voltage that controls the voltage controlled oscillator, even though the television signal has residual sideband characteristics. Therefore, interference caused by noise components to the demodulated video signal and audio signal can be eliminated.

DESCRIPTION OF EMBODIMENTS Hereinafter, embodiments of the present invention will be described with reference to the drawings. FIG. 2 shows a block diagram of essential parts of a television synchronization receiver according to an embodiment of the present invention. In FIG. 2, 9 is a high frequency input section, 10 is a first synchronous detector, 11 is a second synchronous detector, 12 and 13 are first and second low frequency filters, and 14 and 15 are signal amplifiers. , 16 and 17 are third and fourth low frequency filters, 18 is a phase detector, 19 is a Costas loop low frequency filter,
20 is a voltage controlled oscillator, 21 is a 90° phase shifter,
These constitute the Costas loop. 22 is an audio intermediate frequency amplifier, 23 is a frequency discriminator, 24
25 is a voltage subtractor, and 25 is a low-frequency wave generator, which constitutes a frequency pull-in circuit, whose output is sent to the voltage adder 26, which is a low-frequency wave generator 19 of the Costas loop.
is added to the output of 27 is a voltage storage device, 28
2 is a voltage selector, and 29 is a control input device, which constitute a channel selection voltage generation circuit. The output voltage of the voltage selector 28 is also summed with the output of the Costas loop low frequency filter 14 in a voltage adder 26. 30
3 is a video signal amplifier, 31 is a video output device, and 32 is an audio output device.

The operation of the television synchronous receiver of this embodiment constructed in this manner will be described below. It is assumed that the video carrier signal of the channel to be received, which is input to the high frequency input unit 9, is υ _V (t), and the audio carrier signal is υ _S (t). Since υ _V (t) is subjected to residual sideband modulation, it can be expressed as follows.

υ _V (t)=R _e {[I(t)+jQ(t)]expj[ω _V t+
_V )}=I(t)cos( _ωVt + _V )−Q(t)sin( _ωVt + _V )...(1) Here, R _e is the real part of the expression in { }.
I(t) is a signal having an in-phase component with respect to the carrier wave signal, and includes a video signal. Q(t) is a signal of a component orthogonal to the carrier signal, ω _V is the angular frequency of the video carrier signal, and _V is the phase of the video carrier signal.

Furthermore, the narrowband Gaussian noise n(t) is set as n(t)=n _C (t) cos(ω _V t+ _V ) −n _S (t) sin(ω _V t+ _V ) ……(2), and the above υ _V (t) and this n(t) are applied to one terminal of each of the first and second synchronous detectors 10 and 11.

Now, let the output of the voltage controlled oscillator 20 be υ ₀ (t)=A ₀ cos (ω ₀ t+ ₀ )...(3), and apply this to the other terminal of the first synchronous detector 10 consisting of a voltage multiplier. Then, its output υ _PV
(t) is υ _PV (t)=A ₀ [υ _V (t)+n(t)]cos(ω ₀ t+
₀ ) = A ₀ /2 [I (t) + n _C (t)] {cos [(ω _V + ω ₀
)t+ _V + ₀ ]+cos[( _ωV − _ω0 )t+ _V−
0 ]}−A ₀ /2[Q(t)+n _S (t)]{sin[(ω _V +
ω ₀ )t + _V + ₀ 〕+sin〔(ω _V -ω ₀ )t+ _V − ₀ 〕
}...(4) When the voltage controlled oscillator output is synchronized with the video carrier wave, ω ₀ = ω _V , so υ _PV (t) = A ₀ /2 [I (t) + n _C (t)] {cos (2ω
_V t+ _V + ₀ ) + cos ( _V − ₀ )} −A ₀ /2 [Q(t)+n _S (t)] {sin(2ω _V t
+ _V + ₀ ) − sin ( _V − ₀ )}...(5) When the 2ω _V signal is removed by the low-pass filter 12, υ _PV (t) = A ₀ /2 [I(t) + n _C (t )〕cos −A ₀ /2〔Q(t)+n _S (t)〕sin……(
6) where _V = ₀ and is the phase difference between the video carrier and the voltage controlled oscillator output. If = 0, υ _PV (t) = A ₀ /2 [I (t) + n _C (t)] ...(7). In other words, the signal and noise of the in-phase component with respect to the video carrier wave are obtained as the detection output. but,
Orthogonal components are not detected. This detection output is amplified by a signal amplifier 14 via a low-frequency amplifier 12 as a video detection output, and is outputted to a video output device 31 via a video signal amplifier 30.

The wave characteristics of the low frequency filter 12 are shown in FIG. The video signal is waved at the baseband as shown in this figure. When a television signal is received using the conventional superheterodyne reception method, the overall baseband frequency characteristic can be considered flat due to the Nyquist wave characteristics of the intermediate frequency amplifier. The reception method must be considered as shown in Figure 4a. That is, the voltage gain in the low frequency range is twice the gain in the high frequency range. Therefore, in the embodiment shown in FIG. 2, the frequency characteristics of the video signal amplifier 30 are corrected as shown in FIG. 4b.

Since the audio carrier signal υ _S (t) of television broadcasting is frequency modulated, υ _S (t) = A _S cos [{ω _S + s (t)} t + _S ]...(8
). where A _S is the amplitude of the audio carrier signal,
ω _S is the angular frequency of the audio carrier signal, s(t) is the audio signal, and _S is the phase of the audio carrier signal.

This υ _S (t) and υ ₀ (t) of equation (3) are calculated by the synchronous detector 10
, the output is υ _PS (t)=A _S cos [{ω _S +s(t)}t+ _S ]A ₀ cos
(ω ₀ t+ ₀ )=A _S A ₀ /2cos [(ω _S +ω ₀ )t+s(t
)t + _S + ₀ ] + A _S A ₀ /2cos [(ω _S −ω ₀ ) t
+s(t)t+ _S − ₀ ]...(9) When the frequency component of ω _S +ω ₀ is removed by the low-pass filter 12, υ _PS (t)=A _S A ₀ /2cos [(ω _S −ω ₀ )t +s(t)t+ _S − ₀ ] ...(10) If ω _IF =ω _S −ω ₀・ω ₀ =ω _V , then υ _PS (t)=A _S A ₀ /2cos [｛ω _IF +s (t)}t+ _S
− ₀ ] ...(11) υ _PS (t) in equation (10) is nothing but the audio carrier signal shown in equation (8) converted into an audio intermediate frequency signal with an angular frequency ω _IF .

The wave characteristics of the low frequency converter 12 are designed to cover the frequency ω _IF of the audio intermediate frequency signal, as shown in FIG. The audio intermediate frequency signal passes through the low frequency amplifier 12 and is amplified by the signal amplifier 14 and the audio intermediate frequency amplifier 22. The output is demodulated by a frequency discriminator 23 to obtain an audio signal s(t). s(t) is supplied to the audio output device 32.

In the above, the phase of the video carrier wave signal υ _V (t) and the synchronized carrier wave signal υ ₀ (t) of the output of the voltage controlled oscillator 20 are
Although the explanation has been made on the assumption that there is no difference between the phase of

The output υ _Q (t) of the 90° phase shifter is the voltage controlled oscillator 20
Since it has a phase _{difference of} 90 _° with the _{output of} In addition to the second synchronous detector (11), its output υ _PQ
(t) is passed through the low-pass filter 13, υ _PQ (t)=-A ₀ /2 [I(t)+n _C (t)] cos − A ₀ /2 [Q(t)+n _S (t)] sin ... (13) However, it is assumed that ω ₀ =ω _V. This υ _PQ (t) is amplified by a signal amplifier 15 and applied to a phase detector 18.

In the phase detector 18 consisting of a voltage multiplier, υ _PV
(t) and υ _PQ (t) are voltage multiplied, resulting in a control voltage υ _C (t).

υ _C (t)=υ _PV (t)・υ _PQ (t)=−A ₀ ² /8 {[I
(t)+n _C (t)] ² − [Q(t)+n _S (t)] ² }sin
θ −A ₀ /4 [I(t)+n _C (t)][Q(t)+
n _S (t)] cosθ...(14) Here, θ=2. However, the amplification degrees of the first and second signal amplifiers are assumed to be 1 here.

The video carrier signal υ _V (t) is transmitted through vestigial sideband transmission, but its transmission characteristics are different from normal vestigial sideband transmission and consist of a double sideband transmission part and a single sideband transmission part. It's on. That is, the residual sideband characteristic of the video carrier signal υ _V (t) shown in FIG. 5a is the superposition of the double sideband characteristic shown in FIG. 5b and the single sideband characteristic shown in FIG. 5c. .

A signal by double sideband transmission consists only of in-phase components with respect to the phase of the carrier wave, and a signal by single sideband transmission consists of in-phase components and orthogonal components. Now I _L (t)
is the in-phase component of the signal υ _V (t) due to double-sideband transmission,
I _U (t) is the in-phase component of the signal υ _V (t) by single sideband transmission, and Q _U (t) is the signal υ _V (t) by single sideband transmission.
Equation (14) becomes υ _C (t)=−A ₀ ² /8{[I _L (t)+I _U (t)+n _C (t
)] ² − [Q _U (t) + n _S (t)] ² }sinθ −A ₀ /4 [I _L (t) + I _U (t) + n _C (t)] [
Q _U (t) + n _S (t)] cos θ...(15).

If the low frequency characteristics of the low frequency filters ₁₆ and ₁₇ are shown in FIG. ₆ or narrower than that shown in ^FIG . _C ′(t)〕 ² −
[n _S ′(t)] ² }sinθ −A ₀ ² /4[I _L (t)+n _C ′(t)][n _S ′(
t)] cos θ (16) where n _C ′(t) and n _S ′(t) are the synchronous and orthogonal components of narrowband Gaussian noise n(t) after passing through the low-band filter 16.

If I _L (t)≧n _C ′(t) and I _L (t)≧n _S ′(t), υ _C (t)=A ₀ ² /8 [I _L (t)] ² sinθ −A ₀ ² /4 [I _L (t)] [n _S ′ (t)] cosθ……(17
) [I _L (t)] ² ≠ 0, so if the loop bandwidth is narrow enough to remove the second term component in equation (17),
The voltage controlled oscillator 20 is controlled so that θ=0. In other words, the phase error between the video carrier signal υ _V (t) and the output υ ₀ (t) of the voltage controlled oscillator 20 is:
=0.

Here, even if the loop bandwidth is made narrow enough to be 0, the average value of will be 0, and a certain amount of the noise component represented by the second term of equation (17) will remain. This noise component is generated by the voltage controlled oscillator 2.
Gives fluctuation to the output phase and output frequency of 0.

However, when comparing the second term of equation (17) with the second term of equation (15), the difference in amplitude is significantly large.
This is because Q _U (t)≧n _S (t), _S () ² > _S ′ () ² , and I _U (t) is not included in equation (17).
However, _S () ² and _S ′ () ² are each n _S (t)
and the variance of n _S ′(t).

In this way, the low frequency amplifiers 16 and 17 are
By inserting as shown in the figure, the influence of the noise component, ie, the second term of equation (15) or the second term of equation (14), can be significantly reduced.

Furthermore, if the bands of the low frequency filters 16 and 17 are made narrower, the dispersion _S '() ² of n _S '(t) becomes smaller in proportion to the band. The voltage controlled oscillator 20
The fluctuations in the output phase and output frequency become smaller.

The operation in which the television receiver of this embodiment selects a desired channel for reception and enters the reception state is as follows. The voltage storage device 27 corresponds to the desired reception channel input from the control input device 29.
The voltage selector 28 selects the channel selection voltage stored in the voltage selector 28 and applies it to the voltage adder 26. The voltage controlled oscillator 20 is controlled by this channel selection voltage, and a synchronous carrier signal υ ₀ (t) is generated. The audio carrier signal υ _S (t) and the synchronous carrier signal υ ₀ (t) are applied to a synchronous detector 10, resulting in the generation of the audio intermediate frequency signal υ _PS (t). The frequency of the audio intermediate frequency signal υ _PS (t) is changed by the frequency pull-in circuit to
The carrier frequency ω _V of the broadcast video carrier signal υ _V (t) and the carrier frequency ω _S of the audio carrier signal υ _S (t)
The frequency of the synchronous carrier signal υ ₀ (t) is controlled so that it becomes equal to the difference between ω IF and ω _IF . When this frequency falls into the frequency pull range of the Costas loop, the Costas loop rapidly enters a state of phase locking. When the Costas loops are phase synchronized, the synchronous detector 10 obtains a video signal υ _PV (t) and an audio intermediate frequency signal υ _PS (t). These signals pass through a low frequency converter 12, etc., and the video signal is sent to a video output device 31.
Next, the audio intermediate frequency signal is demodulated by the frequency discriminator 23, and the audio signal that is the demodulated signal is output to the audio output device 32.

Effects of the Invention As is clear from the above description, according to the present invention, the outputs of the first and second synchronous detectors of the Costas loop are used to convert the video signal baseband and the audio signal into the first and second low-band detectors. The frequency portion of the residual sideband of the television signal of the output of the first and second low-pass filters having the double-sided band characteristic is transmitted to the third and fourth frequency bands. The configuration is such that the phase difference between the video carrier wave and the synchronized carrier wave is detected from the outputs of the third and fourth low frequency waveformers. This has the effect of significantly reducing phase and frequency fluctuations of the voltage controlled oscillator output.

[Brief explanation of drawings]

Fig. 1 is a block diagram of the main part of the conventional example, Fig. 2 is a block diagram of the main part of an embodiment of the present invention, Fig. 3 is a frequency characteristic diagram of a low frequency filter that waves the output of the synchronous detector, and Fig. 4 is a diagram of the main part of the conventional example. Figure a is a baseband frequency characteristic diagram of a video signal, Figure 4 b is a frequency characteristic diagram of a video signal amplifier, Figure 5 a is a characteristic diagram of vestigial sideband transmission of a television signal, and Figure 5 b is a diagram of a television signal. Characteristic diagram showing double sideband transmission during vestigial sideband transmission of a signal, No. 5
FIG. c is a characteristic diagram showing single sideband transmission during vestigial sideband transmission of a television signal, and FIG. 6 is a frequency characteristic diagram of the third and fourth low-band filters. DESCRIPTION OF SYMBOLS 10...First synchronous detector, 11...Second synchronous detector, 12...First low-pass filter, 13...Second low-pass filter, 16...Third low-pass filter, 17... Fourth
18...phase detector, 20...voltage controlled oscillator, 21...90° phase shifter.

Claims (1)

[Claims] 1. A voltage controlled oscillator, a 90° phase shifter that shifts the phase of the output of this voltage controlled oscillator by 90°, and the output of the voltage controlled oscillator and the output of the 90° phase shifter, respectively, as synchronous carrier waves. first and second synchronous detectors that detect in-phase and quadrature components of a video carrier signal input from the receiver input section using two synchronous carrier waves; first and second low-pass filters that generate low frequencies at the frequencies of the signal baseband and the audio intermediate frequency signal; and residual sidebands of the television signal output from the first and second low-pass filters. 3rd and 4th low frequency filters that lower the frequency portion having both sideband characteristics of A television synchronization receiver comprising a phase detector for detecting the phase and means for feeding back the output of the phase detector to the voltage controlled oscillator.