JPH0413959B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION [Object of the Invention] (Industrial Application Field) The present invention relates to a high-speed, large-capacity AC motor drive device used in a steel rolling mill, a water pump, a tunnel exhaust blower, or the like.

(Prior Art) Motors can be broadly divided into DC motors and AC motors. The former has the advantages of small torque ripple, excellent control performance, and ease of handling, and has been used in a wide range of fields. However, because maintenance of brushes and commutators is time-consuming and there are limits to increasing speed and capacity, there is a recent trend toward replacing them with AC variable speed motors.

Typical AC motors are induction motors and synchronous motors. Other motors include reluctance motors and hysteresis motors, but their application fields are quite limited.

A motor that naturally commutates a thyristor inverter using the back electromotive force of a synchronous motor is generally known as a non-commutator motor. Since this commutatorless motor uses natural commutation, it is easy to increase the capacity and the control performance is similar to that of a DC motor, so it has been applied to various fields. However, since it requires field poles,
The main body of the electric motor is large, and the overload capacity is small due to the limitations of natural commutation.

Induction motors, particularly squirrel cage induction motors, have the advantage of being simple in structure, robust, and easy to handle. On the other hand, a self-excited inverter is required, and there are restrictions from the converter.

Recently, self-extinguishing elements such as transistors and gate turn-off thyristors have been made to have larger capacities and have come to be used in the above self-excited inverters.
In particular, a pulse width modulation control (PWM) inverter can supply a sinusoidal current to the motor, making it possible to achieve an AC variable speed motor with low torque ripple and low noise. Further, as a constraint method, techniques such as V/f=constant control, slip frequency control, vector control, etc. have been established, and it is also known that characteristics comparable to a DC machine can be obtained.

Further, a cycloconverter is a typical example of a natural commutation using the voltage of an AC power source. This cycloconverter can supply a sinusoidal current to a motor, and has the advantage of being easy to increase capacity because of natural commutation. Particularly recently, a reactive power compensating type cycloconverter (Japanese Patent Publication No. 14988/1988), which always controls the input power factor at the receiving end to 1, has been attracting attention.

(Problems to be Solved by the Invention) The conventional AC motor drive devices described above are utilized in various fields by taking advantage of their respective advantages.

However, when it comes to a device that drives a large-capacity, high-speed electric motor, the current situation is that it cannot be easily achieved using the above-mentioned conventional techniques.

That is, since the cycloconverter uses natural commutation, it is easy to increase the capacity, but the output frequency is low and high speed cannot be achieved. Furthermore, self-excited inverters require self-extinguishing elements such as transistors and gate turn-off thyristors, making the device expensive and causing problems such as difficulty in increasing the capacity.

In addition, since commutatorless motors have natural commutation, it is possible to increase the capacity and it is relatively easy to increase the speed.
Since the electric motor itself is complicated and large, and rectangular wave current is supplied to the armature winding, there are problems such as large torque ripple. Furthermore, there remain problems with commutation during startup, overload capacity, etc.

On the other hand, as the capacity of electric motors increases, the effects of reactive power and harmonics generated on the power supply side cannot be ignored. Fluctuations in reactive power cause fluctuations in power system voltage, which has various negative effects on electrical equipment connected to the same system. Further, harmonic currents cause inductive disturbances in televisions, radios, and communication lines, and the third, fifth, and seventh harmonics generated by converters are particularly disliked as they are difficult to remove.

Reactive power compensation type cycloconverter
-14988, etc.) is a power converter that can always maintain the input power factor at the receiving end at 1, and is an effective means of solving the above reactive power problem, but harmonic current that depends on the output frequency is generated on the input side. Because it appears, we have to take great care to deal with it.

Furthermore, recently, power converters (such as Japanese Patent Application Laid-open No. 59-61475) that have both the functions of an AC/DC power converter and an active filter have been announced, and AC motor drive devices that combine the above-mentioned automatic inverter + induction motor device have been announced. is attracting attention.

This method has the advantage that the input current is controlled to a sine wave that is in phase with the power supply voltage, so there are few harmonics and the input power factor can always be maintained at 1. It has to be constructed with self-extinguishing elements, making it difficult to increase the capacity and being economically unsatisfactory.

The present invention has been made in view of the above-mentioned problems, and has a sine frequency of 0 to several hundred Hz for AC motors (induction motors, synchronous motors, reluctance motors, etc.) with respect to the commercial power frequency (50 Hz or 60 Hz). It is an object of the present invention to provide a high-speed, large-capacity AC motor drive device that supplies wave current, has a power supply whose input power factor is always 1, and has few harmonics.

[Means for Solving the Problems] In order to achieve the above object, the present invention comprises an AC power supply, a non-circulating current type cycloconverter whose output side terminal is connected to the AC power supply via a power transformer, and the non-circulating current type cycloconverter. A phase advancing capacitor connected to the input side terminal of a circulating current type cycloconverter, a circulating current type cycloconverter having the input side terminal connected to the phase advancing capacitor via an isolation transformer, and an output side of the circulating current type cycloconverter. and an AC motor connected to a terminal, the non-circulating current type cycloconverter converts the current supplied from the AC power supply into the power supply voltage so that the voltage peak value of the phase advance capacitor is approximately constant. Control is performed so that the power is controlled to be an in-phase (input power factor = 1) sine wave (with small harmonic components), and the circulating current type cycloconverter supplies variable voltage variable frequency power to the AC motor. It is characterized by

(Function) In other words, a non-circulating current type cycloconverter converts power between an AC power supply (constant frequency of 50Hz or 60Hz) and a phase advance capacitor (constant frequency of 500Hz, for example). The current supplied from the AC power supply is controlled so that the applied voltage value is approximately constant.
At this time, by controlling the current supplied from the power source to be a sine wave in phase with the power source voltage, the input power factor is always 1, and the current can have few harmonic components.

In addition, the circulating current type cycloconverter converts power between the phase advance capacitor and the AC motor, and the frequency of the phase advance capacitor is 500Hz, while the AC motor's armature winding has a frequency of about 0 to 500Hz. can supply a sinusoidal current with a frequency of .

At this time, the phase advance capacitor becomes a reactive power source for the two cycloconverters,
The frequency (500 Hz) is determined so that the lagging reactive power of the two cycloconverters and the leading reactive power of the phase advance capacitor are equal. Conversely, an external sine wave oscillator (frequency 500Hz)
By providing a phase control reference signal for the converter, a circulating current of the circulating current type cycloconverter flows such that the frequency and phase of the phase advance capacitor voltage match the frequency and phase of the oscillator.

With the phase advance capacitor voltage established in this way, both cycloconverters can perform power conversion with only natural commutation operation.Moreover, with respect to the AC power frequency of 50Hz (or 60Hz), the AC motor is It can supply sinusoidal current of several hundred Hz.

(Embodiment) FIG. 1 is a configuration diagram showing an embodiment of an AC motor drive device of the present invention.

In the figure, R, S, and T are the receiving terminals of the 3-phase AC power supply,
MWR is a power transformer, CC-1 is a non-circulating current type cycloconverter, CAP is a △ wired or wired high frequency three-phase phase advance capacitor, TR _U ,
TR _V and TR _W are isolation transformers, CC-2 is a circulating current type cycloconverter, and M is an AC motor (three-phase squirrel cage induction motor).

The non-circulating current type cycloconverter CC-1 is
It is configured for each R phase, S phase, and T phase, and each output terminal is connected to a power transformer MTR via AC reactors L _SR , L _SS , and L _ST .

The R-phase cycloconverter is a positive group converter.
It consists of SPR and negative group converter SNR.
Similarly, the S-phase cycloconverter includes a positive group converter SPS and a negative group converter SNS, and the T-phase cycloconverter includes a positive group converter SPT and a negative group converter SNT.

In addition, circulating current type cycloconverter CC-2
is configured for each U phase, V phase, and W phase, and each output terminal is connected to the armature winding of the AC motor M.

The U-phase cycloconverter is a positive group converter.
It is composed of an SPU, a negative group converter SNU, and DC reactors L _pu1 and L _pu2 , and the two converters SPU and SNU are isolated on the input side by an isolation transformer TR _u .

Similarly, the V-phase cycloconverter consists of a positive group converter SPV, a negative group converter SNV, and DC reactors L _OV1 and L _OV2 , and the W-phase cycloconverter consists of a positive group converter SPW, a negative group converter
Consists of SNW and DC reactors L _OW1 and L _OW2 ,
Each converter is isolated on the input side by isolation transformers TR _V and TR _W.

The input side terminals of CC-1 and CC-2 are connected to the high frequency phase advancing capacitor CAP.

In addition, as a control circuit, there is a rotating pulse generator PG directly connected to the motor, current detection current transformers _CTR , _CTS ,
CT _T , CT _U , CT _V , CT _W , voltage detection transformer
PT _S , PT _CAP , rectifier circuit D, voltage control circuit AVR,
Speed control circuit SPC, current control circuit ACR ₁ , ACR ₂ ,
Phase control circuits PHC ₁ and PHC ₂ are provided.

The non-circulating current type cycloconverter CC-1 is supplied from a three-phase AC power supply so that the peak value V _cap of the three-phase AC voltages V _a , V _b , and V _c applied to the high-frequency phase advance capacitor CAP is almost constant. Currents I _R , I _S ,
Control _IT .

In addition, circulating current type cycloconverter CC-2
In this example, the high-frequency phase advancing capacitor CAP is used as a three-phase voltage source, and the induction motor M is provided with a variable voltage and variable frequency three-phase voltage source.
Provides phase alternating current power. At the same time, the circulating current flowing through CC-2 is automatically adjusted so that the frequency and phase of the voltage applied to the phase advance capacitor CAP match the frequency and phase of the externally applied three-phase reference voltage. .

The detailed operation will be explained below.

First, the voltages of the phase advance capacitors CAP are V _a , V _b , V _c
The startup operation for establishing this will be explained.

The voltages of the R, S, and T phases of a three-phase AC power supply can be expressed as follows. However, V _sn is the voltage peak value, ω _s =
2πf _s is the power source angular frequency.

V _R =V _sn・sin(ω _s t) …(1) V _S =V _sn・sin(ω _s t−2π/3) …(2) V _T =V _sn・sin(ω _s t+2π/3) ...(3) If _{the frequency f cap on} the input side (phase advance capacitor side) of the cycloconverter CC-1 is sufficiently high with respect to the frequency f s of the power supply, the upper power supply voltage V _R , V _S and V _T can be replaced with DC voltages.

FIG. 2 shows how the phase advance capacitor CAP is charged through the positive group converter SPR of the R-phase cycloconverter, and shows the case where firing pulses are applied to the thyristors S ₂ and S ₄ .

The charging current I _R is expressed as one route V _R ⁺ →L _SR →S ₄
→C _ab →S ₂ →V _R ^- , and the other route is V _R ⁺ →L _SR →S ₄ →C _ca →C _bc →S ₂ →V _R ^- . As a result, the voltage V _ca = +V _R is applied to the phase advance capacitor C _ba , and the phase advance capacitor
A voltage V _bc =-(1/2) V _R is applied to C _bc , and a voltage V _ca =-(1/2) V _R is applied to C _ca.

Next, when a firing pulse is given to thyristor S ₃ , the voltage of capacitor C _bc is applied to thyristor S ₂ , and S ₂ is turned off. As a result, V _ab =
Charged to +(1/2)V _R , V _bc = +(1/2)V _R , V _ca = −V _R .

A firing pulse is then given to thyristor S ₅ , turning off thyristor S ₄ and V _ab =-(1/
2) Charged to V _R , V _bc = +V _R , V _ca = -(1/2) V _R .

Figure 3 shows the firing modes of thyristors S ₁ to _{S 6} in Figure 2, the applied voltage V _ab of capacitor C _ab , and the phase voltage V _a
shows the relationship with the substrate wave.

Since the voltage _Vab is charged via the reactor _LSR , it gradually rises as shown by the broken line. If that time is 2δ, the fundamental wave component of V _ab is delayed by δ. The phase of the phase voltage V _a lags the line voltage V _ab by (π/6) radians.

As can be seen by comparing the ignition mode and the phase voltage V _a , the phase control angle α _PR at the time of starting the converter SNR is α _PR = π−δ (radians) (4). Since δ is not very large, approximately it is operated at α _PR ≒180°.

At this time, the output voltage V _PR of the converter SPR is V _PR =k·V _cap ·cosα _PR <0 (5) and is balanced with the power supply voltage V _R . however,
w is a proportional constant, and V _cap is the phase voltage peak value of the capacitor.

However, if this continues, the phase advance capacitor CAP will
A voltage higher than the power supply voltage V _R is not charged.

Therefore, the ignition phase angle α _PR is slightly shifted in the direction of 90°. Then, the output voltage V _PR shown by equation (5)
decreases, and V _R > -V _PR . As a result, the charging current I _R increases, increasing the capacitor voltage V _cap ,
It settles down to V _R = -V _PR . At this time, I _R is zero. If it is desired to further increase V _cap , this can be achieved by further shifting the phase angle α _PR in the direction of 90° and decreasing the output voltage V _PR . α _PR
= 90°, V _PR = O ^V , and theoretically the power supply voltage
Even if V _R has a very small value, the capacitor voltage V _cap can be charged to a large value. However, since there is actually circuit loss, it is essential to supply power for that amount.

If the power supply voltage V _R fluctuates, the above phase angle α _PR
By changing the capacitor voltage accordingly,
V _cap can be kept almost constant.

The above description has been made taking the case where the power supply voltage V _R is positive as an example, but when V _R becomes a negative value, the capacitor CAP can be charged through the negative group converter SNR.

Furthermore, when the R, S, and T phases are operated simultaneously, the phase advance capacitor CAP can be charged in the same way, although the operation becomes somewhat complicated.

Next, the voltages V _a , V _b , V _c of the phase advancing capacitors CAP established in this way are applied to the three-phase reference voltages e a , V a , V c applied to the phase control circuits PHC ₁ , PHC _{2 in} FIG.
Explain that the frequencies and phases of e _b and e _c match.

FIG. 4 is an equivalent circuit diagram showing the positive group converter SPR of the R-phase cycloconverter, the phase advance capacitor CAP, and the U-phase component of the circulating current type cycloconverter CC-2. However, the isolation transformer TR _U is omitted.

The phase control circuit PHC ₁ of the non-circulating current type cycloconverter CC-1 and the phase control circuit PHC ₂ of the circulating current type cycloconverter CC-2 are both controlled by external three-phase reference voltages e _a , e _b , e _c Phase controlled.

The non-circulating current type cycloconverter CC-1 has voltages V _a , V _b , V _c applied to the phase advance capacitor CAP.
The peak value V _cap is controlled to be approximately constant.

On the other hand, circulating current type cycloconverter CC
-2, when supplying currents I _U , I _V , I _W to the AC motor M, the frequencies and phases of the voltages V _a , V _b , V _c applied to the phase advance capacitor CAP are controlled by the phase control circuit PHC. ₁ , CC is set to match the frequency and phase of the three-phase reference voltage e _a , e _b , e _c input to PHC ₂ .
-Adjust the circulating current of -2. Here, the operation will be explained using a U-phase cycloconverter as an example.

The output voltage V _PU of the U-phase positive group converter SPU and the output voltage V _NU of the negative group converter SNU are equally balanced at the intermediate tap point between the DC reactors L _OU1 and L _OU2 . The output voltages V _PU and V _NU are expressed by the following equations when the control phase angles of the respective converters are α _PU and α _NU .

V _PU =k・V _cap・cosα _PU …(6) V _NU =−k・V _cap・cosα _NU …(7) However, K is a proportionality constant, and V _cap is a phase advance capacitor.
This is the peak value of the voltage applied to CAP.

Therefore, the following relational expression holds between the control phase angles α _PU and α _NU .

α _NU = 180 _° − α _{PU … (} 8 ₎ Figure 5 shows _the positive group and The firing mode of the negative group converter is shown.

If the frequency of the voltage applied to the phase advance capacitor CAP decreases, V _a ′, V _b ′, V _c ′ as shown by the broken line
Consider the case where

Since the ignition pulse of each converter is determined based on the signals e _a , e _b , e _c , the reference pressures e _a , e _b ,
e The phase with respect to _c does not change.

However, the control phase angles α PU ′, α _NU ′ with respect to the voltages V _a ′, V _b ′, V _c ′ applied to the actual phase advance capacitor CAP change, and α _PU ′<α _PU , α _{NU ′} < α _NU
becomes.

As a result, the positive group converter shown by equation (6)
The output voltage V _PU of the SPU increases, and the output voltage V _NU of the negative group converter SPU shown in (7) decreases.
Therefore, the circulating current of the U-phase cycloconverter increases, increasing the delayed reactive power on the input side (phase advance capacitor side).

FIG. 6 shows an equivalent circuit for one phase on the input side of the cycloconverter, in which cycloconverters CC-1 and CC-2 are replaced with a variable inductance _Lcc that takes a delayed current. The resonant frequency f _cap of this circuit is given by the following equation.

f _cap = 1/(2π√ _cc・_cap ) …(9) An increase in the circulating current is equivalent to a decrease in the equivalent inductance L _cc , and the above frequency f _cap increases, and V _a ′, V _b ′ , the frequency f _cap of V _c ′ is the reference voltage
The frequency of e _a , e _b , e _c approaches f _cc .

Similarly, when f _cap >f _c , the circulating current decreases, L _cc increases, and f _cap =f _c , which settles down.

When the phase of the voltage of the phase advance capacitor CAP lags behind the phase of the reference voltage, the circulating current increases in the same way as when f _cap < f _c above, increasing f _cap and increasing the voltage of the phase advance capacitor CAP. Advance the phase.

Conversely, when the voltage phase of the phase advance capacitor CAP leads the reference voltage, the circulating current decreases in the same way as when f _cap > f _c above, and the phase advance capacitor CAP
Delays the voltage phase of CAP. In this way, the voltages V _a , V _b , V _c of the phase advance capacitor CAP are set to the reference voltage
The magnitude of the circulating current is automatically adjusted so that it has the same frequency and phase as e _a , e _b , and e _c .

Let the reference voltage be e _a = sin (ω _c・t) …(10) e _b = sin (ω _c・t−2π/3) …(11) e _c = sin (ω _c・t+2π/3) …(12 ), the voltage of the phase advance capacitor CAP is V _a = V _cap・sin (ω _c・t) … (13) V _b = V _cap・sin (ω _c・t−2π/3) … (14) V _c =V _cap・sin(ω _c・t+2π/3) (15) However, ω _c =2πf _c .

Next, returning to Figure 1, the above phase advance capacitor
The operation of controlling the voltage peak value V _cap of CAP to be constant will be explained.

Figure 7 shows the cycloconverter CC-1 shown in Figure 1.
This is a detailed representation of the control circuit shown in FIG.

First, the voltage control circuit AVR in Figure 1 is the voltage setter VR, comparator _CC , and voltage control compensation circuit in Figure 7.
Consists of G _c (S).

Further, the following current control circuit ACR ₁ and phase control circuit PHC ₁ are configured for each of the R, S, and T phases, and the R-phase control circuit R-CONT and the S-phase control circuit S- in FIG.
CONT and T-phase control circuit T-CONT.

The R-phase control circuit R-CONT includes a multiplier M _R , a comparator C _R , a current control compensation circuit G _R (S), and an inverting circuit
It is composed of INV _R , phase control circuits PHP _R , _PHNR , analog switches _ASPR , ASN _R , Schmitt circuit S _R , monomulti circuit M _R , and logic circuit _LCR .

The S-phase and T-phase control circuits are similarly configured.

The reference voltage generator O _sc in FIG. 7 corresponds to that in FIG.

First, transform the voltage of the phase advance capacitor CAP to
The three-phase voltage is detected by the PT _cap and rectified by the rectifier D. This allows the phase advance capacitor to
The voltage peak value V _cap of CAP is detected and input to the comparator _CC .

Further, a voltage command value V _cap ^* is output from the voltage setting device VR, and is compared with the detected value V _cap by a comparator _CC . The deviation ε _c =V _cap ^* −V _cap is input to the next voltage control compensation circuit G _c (S), where it is integrated or proportionally amplified. To simplify the explanation here,
Assume that only proportional amplification is performed, with G _c (S) = K _c . The output _sn of the step compensation circuit _Gc (S) serves as the peak value command of the currents _R , _S , and _T supplied from the power supply.
Multipliers for R, S, and T phase control circuits ML _R ~ ML _S
is input.

On the other hand, the power supply voltages V _R , V _S , V _T are detected by the transformer PT _s and multiplied by the reciprocal of the peak value V _sn to generate the next three-phase unit sine waves φ _R , φ _S , Find _φT .

φ _R = (V _R / V _sn ) = sin (ω _s・ t) ... (16) φ _S = (V _S / V _sn ) = sin (ω _s・ t - 2π / 3) ... (17) φ _T =(V _T /V _sn )=sin(ω _s・t+2π/3) …(18) These three-phase unit sine waves φ _R , φ _S , φ _T are applied to the above multiplier.
It is input to ML _R , ML _S , and ML _T and multiplied by the peak value I _sn . The results are the command values I _R ^* , _IS ^* , and _IT ^* of the current supplied from the power source.

I _R ^* = I _sn・sin (ω _s・t) … (19) I _S ^* = I _sn・sin (ω _s・t−2π/3) … (20) I _T ^* = I _sn・sin (ω _s ·t+2π/3) (21) The R-phase current I _R is controlled as follows.

The R phase current I _R is detected by the transformer CT _R , and the comparator
Enter C _R. The comparator C _R converts the above command value to the I _R
^* and the detected value I _R to find the deviation ε _R = I _R ^* − I _R.
The deviation ε _R is input to the current control compensation circuit G _R (S), and proportional amplification is performed. Assuming that the constant of proportionality is K _R ,
One of the output signals K _R and ε _R of the compensation circuit G _R (S) is input as V〓 _PR = K _R /ε _R to the phase control circuit PHC _{R of the positive group converter SPR, and the other is input to the phase control circuit PHC R} of the positive group converter SPR. It is input to the phase control circuit PHN _R of the negative group converter SNR via INV _R as V〓 _NR =-K _R ·ε _R.

The phase control circuits PHD _R to _PHNR are based on a known method, and output signals e _a , e _b , e _c from the three-phase reference voltage generator _Osc and the phase control input voltages V〓 _PR , V〓 _NR
The ignition pulse signal is obtained from the intersection point.

That is, when the phase control input voltage V〓 _PR is input to the phase control circuit PHP _R , the control phase angle α _PR becomes α _PR = cos ^-1 {K〓·V〓 _PR } (22). However, K〓 is a proportionality constant.

Rewriting this, we have the following relationship: cosα _PR = k〓・V〓 _PR …(23). The output voltage V _PR of the converter SPR has the relationship V _PR =k・V _cap・cosα _PR , and V _PR ∝V〓 _PR .

Similarly, the negative group converter SNR has the relationship V _NR Vα _NR .

On the other hand, the R-phase current command value I _R ^* is input to the Schmitt circuit S _R and is converted into a rectangular wave signal SG ₁ that rises at its zero-crossing point.

The rectangular wave signal SG ₁ is input to the mono multi-circuit MM _R , and the time T ₁ at the rise and fall of the signal SG ₁
Create a pulse signal SG ₂ that becomes "1" only when

The next logic circuit _LCR uses the signals SG ₁ and SG ₂ to provide the next AND outputs SG ₃ and SG ₄ .

SG ₃ = SG ₁・₂ … (24) SG ₄ = ₁・SG ₂ … (25) The above relationships are shown in Figure 8.

When the signal _SG3 is "1", the analog switch _ASPR shown in FIG. 7 is turned on and a gate signal is given to the positive group converter SPR.

Further, when the signal SG ₄ is "1", the analog switch ASN _R shown in FIG. 7 is turned on and a gate signal is given to the negative group converter SNR.

Therefore, when the R phase current command I _R ^* is a positive value,
Positive group converter SPR operates, negative group converter
SNR is paused. Conversely, when I _R ^* is a negative value, the negative group converter SNR operates, and the positive group converter SPR operates.
pauses. Furthermore, when I _R ^* changes from positive to negative or from negative to positive, both converters are stopped for a time T ₁ to prevent circulating current from flowing.

The case where I _R ^* >0 will be explained as an example.

When the R-phase current command I _R ^* becomes larger than the actual current I _R , ε _R = I _R ^* − I _R becomes a positive value, and the control compensation circuit
Increase the phase control input signal V〓 _PR via G _R (S). Therefore, the output voltage V _PR of the positive group converter SPR increases, the R-phase current I _R is increased, and the control is performed so that I _R ≈I _R ^* .

Conversely, when I _R ^* < I _R ^* , the deviation ε _R becomes a negative value, and V〓 _PR is decreased via the control compensation circuit G _R (S). Therefore, the SPR output voltage V _PR decreases, reducing the R phase current I _R. As expected, it settles down to I _R ≒ I _R ^* .

In this way, the R-phase current I _R is controlled sinusoidally in accordance with its command value I _R ^* .

When I _R ^* < 0, negative group converter SNR
Again, it is controlled so that I _R ≒ I _R ^* .

Output current waveforms I _PR and I _NR of the positive group converter SPR and negative group converter SNR shown in FIG. 8 are shown, respectively. When the current I _R switches from positive to negative or from negative to positive, both converters stop operating, so a rest period T ₁ occurs.

The S-phase and T-phase currents, I _S and I _T are also controlled in the same way.

Now, in this way, the R-phase, S-phase, and T-phase currents are controlled to match the command values I _R ^** , I _S ^* , I _T ^* , but the command values are the power supply voltage V _R , It is a sine wave that is in phase with V _S and V _T .

In other words, the input power factor is always 1, and operation with very few harmonic components is possible.

Next, the control operation of the voltage peak value V _cap of the phase advancing capacitor CAP will be explained.

When V _cap ^* > V _cap , the deviation ε _C = V _cap ^* −
V _cap becomes a positive value, and the current peak value command I _sn = K _c ,
ε _c also becomes a positive value and increases.

Therefore, the input currents I _R , I _S , and I _T of each phase of the power supply also increase,
Active power P _S expressed by the following equation is supplied from the power source.

P _s = I _R · V _R + I _S · V _S + I _T · V _T = 3/2 · V _sn · I _sn ... (26) However, V _sn is the voltage peak value, and I _sn is the current peak value.

As a result, energy P _s ·t is supplied from the power supply to the phase advance capacitor CAP, and is stored as (1/2) C _cap V _cap ² . Therefore, the voltage V _cap increases and finally
It settles down as V _cap ≒ V _cap ^* .

Conversely, when V _cap ^* <V _cap , the deviation ε _c becomes a negative value, and the current peak value command I _sn also becomes a negative value.
Therefore, the energy (1/2) C _cap V _cap ² stored in the phase advancing capacitor CAP becomes P _s ·t and is regenerated to the power supply. Therefore, the voltage V _cap decreases and again
It is controlled so that V _cap ≒ V _cap ^* . At this time, the currents I _R , I _S , and I _T of each phase of the power supply are respectively the power supply voltages V _R ,
It is controlled to a sinusoidal wave having an opposite phase to V _S and V _T , and it is possible to maintain an input power factor of 1 as well.

As mentioned above, the non-circulating current cycloconverter
CC-1 is the voltage peak value V _cap of the phase advance capacitor CAP
It controls the currents I _R , I _S , I _T supplied from the power supply so that the current command values I _R ^* , I _S ^* , I _T ^* match the command values V _cap ^* . By applying a sine wave that is in phase (or in opposite phase) to the voltage, the input power factor can always be maintained at 1.

Also, at this time, the circulating current is set so that the frequency and phase of the voltage of the phase advance capacitor CAP match the frequency and phase of the three-phase reference voltage signals e _a , e _b , e _c given by _the external oscillator Osc. The adjustment of the circulating current type cycloconverter CC-2 is as described above.

Next, the operation of controlling the current supplied to the AC motor by the circulating current type cycloconverter CC-2 will be explained.

FIG. 9 is a block diagram showing a specific embodiment of the control circuit of the circulating current type cycloconverter CC-2 shown in FIG. 1. Corresponding to the control circuit shown in FIG. 1, the result is as follows.

First, the speed control circuit SPC shown in Fig. 1 includes the comparator C _N shown in Fig. 9, the speed control compensation circuit G _N (S), the excitation current setting device EX, the calculation circuits CAL ₁ to _{CAL 3} , and the three-phase sine wave pattern generation. It consists of a PTG.

In addition, current control circuit ACR ₂ and phase control circuit
PHC ₂ is provided for each of the U-phase, V-phase, and W-phase, and is connected to the U-phase control circuit U-CONT and the V-phase control circuit V-, respectively.
CONT and W-phase control circuit W-CONT.

The U-phase control circuit U-CONT includes a multiplier M _U , a comparator C _U , a current control compensation circuit G _U (S), and an inverting circuit INV _U
and phase control circuits PHP _U and _PHNU . As mentioned above, the phase control circuits PHP _U and _PHNU are controlled based on the three-phase reference voltages _ea , _eb , and _ec from the external oscillator _Osc .

V-phase and W-phase control circuits V-CONT and W-
CONT is similarly configured.

First, the control operation of the U-phase current _IU will be explained.

Multiplier M _U multiplies current peak value command I _Ln by unit sine wave φ _u =sin {(ω _r +ω _sl ^* )·t+θ _r ^* } to provide command value I _U ^* of U-phase load current.

The U-phase current I _U is detected by the current transformer CT _U and inputted to the comparator C _U. The comparator C _U compares the command value I _U ^* and the detected value I _U and provides the deviation ε _U = I _U ^* − I _U to the next current control compensation circuit G _U (S). G _U (S) proportionally amplifies the deviation ε _U (the multiplication factor is K _U ), and one is the phase control circuit of the positive group converter SPU.
Give PHP _U as V〓 _PU = K _u・ε _U. Yet another one
One is applied to the phase control circuit PHN _U of the negative group converter SNU via the inverting circuit INV _U as V〓 _PU =K _u ·ε _U.

Therefore, the output voltage of the positive group and negative group converters V _PU
and V _NU are as follows.

V _PU =k・V _cap・cosα _PU =k′・V _cap・V〓 _PU =k′・V _cap・K _U・ε _U …(27) V _NU =k・V _cap・cosα _NU =k′・V _cap・V〓 _NU =k′・V _cap・K _U・ε _U =V _PU …(28) The U-phase winding of the motor M has the average value V _U of the output voltage of the above positive group and negative group converters. =(V _PU +
V _NU )/2 is applied.

Therefore, when I _U ^* > I _U , the deviation ε _U becomes a positive value, increasing V _U , increasing the U-phase current I _U , and increasing I _U
Control so that ≒I _U ^* .

Conversely, when I _U ^* < I _U , the deviation ε _U becomes a negative value, decreasing V _U and decreasing the U-phase current I _U , and settling down as I _U ≒ I _U ^* .

By changing the current command value I _U ^* in a sinusoidal manner, the actual current I _U is also controlled to follow it.

The V-phase and W-phase currents I _V and I _W are similarly controlled.

However, as mentioned before, the circulating current of each phase is adjusted so that the frequency of the voltages V _a , V _b , V _c of the phase advance capacitor CAP is adjusted so that the phase matches that of the three-phase reference voltages e _a , e _b , e _c Needless to say, it will be adjusted accordingly.

Next, the speed control operation of the induction motor M will be explained.

An induction motor in which the secondary current I and the excitation current _Ie are vectorally orthogonal so that they can be controlled independently is known as a vector control induction machine. Here, an example of speed control using this method is taken.

There are many literatures on vector control techniques, so a detailed explanation will be omitted and only a summary will be given.

First, a pulse train proportional to the rotational speed ω _r is extracted from a rotational pulse generator PG directly connected to the rotor of the electric motor.

The comparator C _N compares the rotational speed ω _r with its command value ω _r ^* , and inputs the deviation ε _N =ω _r ^* −ω _r to the next speed control compensation circuit G _N (S).

G _N (S) consists of a proportional element, an integral element, etc., and provides a torque current command I〓 ^* as an output.

Further, the detected rotational speed value ω _r is input to the excitation current setting device EX, and provides an excitation current command I _e ^* .

The relevant torque current command I〓 ^* and excitation current command I _e ^* are:
It is input to the calculation circuits CAL ₁ to _{CAL 2} and performs the following calculation.

That is, in the calculation circuit CAL ₁ , ω _sl ^* = R _r ^* I〓 ^* /L _r ^* I _e ^* …(29) R _r ^* : Secondary resistance L _r ^* : Slip angular frequency is determined by calculation of secondary inductance. Find ω _sl ^* .

In addition, the calculation circuit CAL ₂ calculates the phase angle of the primary current command value I _L ^* with respect to the excitation current I _e ^* by calculating θ _r ^* = tan ^-1 (I〓 ^* /I _e ^* ) ...(30) Find θ _r ^* .

Further, the calculation circuit CAL ₃ calculates the peak value I _Ln of the primary current command value I _L ^* by calculating I _Ln = √ _e ^* + 〓 ^* (31).

Figure 10 shows a current vector diagram of the induction motor. The exciting current I _e ^* and the secondary current (torque current) I ^* are in an orthogonal relationship, and the generated torque T _e of the motor is calculated by the following formula: It can be expressed as

T _e = K _e・I〓 ^*・I _e ^* …(32) Normally, the excitation current command I _e ^* is given constant, and the generated torque T _e of the motor is the secondary current command (torque current command command) I〓 ^* controlled by changing the

However, when operating the rotation speed above the rated value, field weakening control is performed and the excitation current setting
Depending on EX, the excitation current command I _e ^* may be changed depending on the rotational speed ω _r .

Slip angular frequency ω _sl obtained in this way
^* , phase angle θ _r ^* and rotational angular frequency (rotation speed detection value)
Input ω _r to a sine wave pattern generator PTG to obtain the next three-phase unit sine waves φ _u , φ _v , φ _w .

φ _u = sin {(ω _r + ω _sl ^* )・t+θ _r ^* }…(33) φ _v = sin {(ω _r +ω _sl ^* )・t+θ _r ^* −2π/3}…(34) φ _w = sin {(ω _r +ω _sl ^* )・t+θ _r ^* +2π/3} …(35) The unit sine waves φ _u , φ _v , φ _w express the frequency and phase of the primary current I _L supplied to the induction motor M. It is up to you to decide.

Multipliers ML _U , ML _V , ML _W multiply the three-phase unit sine waves φ _u , φ _v , φ _w by the peak value command I _Ln to obtain the three-phase current ( Find the command values I _u ^* , I _v ^* , I _w ^* of the primary current).

I _u ^* = I _Ln・sin {(ω _r + ω _sl ^* )・t + θ _r ^* } … (36) I _v ^* = I _Ln・sin {(ω _r + ω _sl ^* )・t + θ _r ^* −2π/3 } …(37) I _w ^* = I _Ln・sin {(ω _r +ω _sl ^* )・t +θ _r ^* +2π/3} …(38) Vector control of an induction motor is based on the excitation current I _e and the secondary current I〓 The feature is that it can be controlled independently. Therefore, by changing the magnitude of the secondary current _I while keeping the motor's excitation current Ie constant, the generated torque can be controlled, making it possible to achieve a speed control response equivalent to that of a DC machine. .

The primary current command value I _u ^* given as above,
As explained ^above , the actual currents I _u , I _v , and I _w are controlled according to I _v * and _I w ^* .

When ω _r ^* >ω _r , the deviation ε _N becomes a positive value and increases the torque current (secondary current) command I 〓 ^* via the control compensation circuit G _N (S).

As a result, 1 of the induction motor shown in FIG.
The peak value I _Ln and phase angle θ _r ^* of the next current command I _L ^* (I _U ^* , I _V ^* , I _W ^* ) are increased, and the actual currents I _U , I _V , and I _W are also controlled accordingly. be done.

Therefore, the actual secondary current I of the induction motor M increases, increasing the generated torque T _e and accelerating the motor. This increases ω _r and is controlled so that ω _r ≒ ω _r ^* .

Conversely, when ω _r ^* <ω _r , the deviation ε _N becomes a negative value, decreasing the torque current command I 〓 ^* and increasing the primary current command I _L ^* (I _U ^* , I _V ^* , I _W ^* )'s peak value I _Ln and phase angle θ _r ^* are decreased. Therefore, the generated torque T _e decreases,
The rotational speed ω _r is reduced and controlled so that ω _r ≒ ω _r ^* .

In the present invention, power exchange between the AC power source and the phase advance capacitor CAP is performed by a non-circulating current type cycloconverter CC-1. As a result, the following effects can be expected.

The input side terminals of the positive group converter and negative group converter can be directly connected to the phase advancing capacitor.
No isolation transformer is required. This allows the device to be made smaller and lighter, and the exercise efficiency is also improved.

Furthermore, it becomes possible to install two elements (thyristors) constituting the converter in the same cooling fan in the forward and reverse directions, thereby simplifying the structure of the converter and making it smaller and lighter.

A DC reactor for suppressing circulating current is not required, improving operational efficiency and reducing the size and weight of the device.

Further, in the device of the present invention, power conversion between the phase advance capacitor and the AC electric power is performed by a circulating current type cycloconverter CC-2. Therefore, the following effects can be obtained.

By using the circulating current type cycloconverter, it becomes possible to increase the upper limit value of the output frequency, and it becomes possible to operate the AC motor M at extremely high speed.

By inserting a high frequency isolation transformer on the input side, each converter can be isolated.
The number of control phases can be increased. Therefore, not only the capacity of the DC reactor becomes small, but also the current pulsation supplied to the motor becomes extremely small.

Voltage V _a applied to phase advance capacitor CAP,
The circulating current of CC-2 is automatically adjusted to match V _b , V _c to the frequency and phase of external reference voltages ea, eb, ec.

As described above, the present invention connects a non-circulating current type cycloconverter and a circulating current type cycloconverter to a phase advance capacitor at the input side terminal, and connects an AC power source (50Hz or 60Hz) to an AC motor with a variable voltage variable frequency (0 to 60Hz). It provides a system that supplies power at a frequency of several hundred Hz).

[Effects of the Invention] The AC motor drive control device as described above provides the following effects.

While the power supply frequency is 50Hz, the frequency of the current supplied to the motor is approximately 0 to 500Hz.

In other words, circulating current type cycloconverter
By setting the number of control pulses (number of control phases) of CC-2 to about 24 pulses, it is also possible to make its output frequency f ₀ higher than the input frequency f _cap .
Therefore, the frequency f _cap of the voltage of the phase advance capacitor CAP is
By setting the frequency to about 500 Hz, the frequencies of the currents I _U , I _V , and I _W supplied to the motor can be operated from about 0 to 500 Hz.

Therefore, in a two-pole AC motor, the rotation speed is
It can reach 30,000rpm, making ultra-high-speed operation possible.
Therefore, in the case of a blower motor, which conventionally required a gear or the like to increase its speed, the gear becomes unnecessary, the operating efficiency is improved, and the blower motor can be made smaller and lighter.

In addition, when the rotation speed is set to around 3000 rpm, the number of poles of the motor can be increased to 20, which not only reduces torque ripple at low speeds, but also
It becomes possible to improve the accuracy of speed control by one order of magnitude compared to the conventional method.

The currents I _U , I _V , and I _W supplied to the electric motor M are controlled to sinusoidal waves, resulting in a device with extremely small torque ripple. At the same time, electromagnetic noise is eliminated, eliminating the cause of pollution.

The current supplied from the power supply is controlled to be a sine wave in phase with the power supply voltage, the input power factor can always be controlled to 1, and harmonics contained in the input current are extremely small. Input power factor = 1 means that the reactive power is zero, which reduces the installed capacity of the power supply system, and also eliminates voltage fluctuations due to reactive power fluctuations, making it possible to create equipment that does not cause any disturbance to other electrical equipment. can do. In addition, induction interference due to harmonics is eliminated, and adverse effects on nearby communication lines are eliminated.

All converters are natural commutation converters (separately excited converters) that commutate using the AC voltage applied to the phase advance capacitor CAP.
It is possible to provide a system that does not require self-extinguishing elements such as high-power transistors or gate turn-off thyristors, is highly reliable, is resistant to overload operation, and is extremely easy to increase capacity.

As a result of input power factor = 1, the power transformer
MTR only needs to supply active power, which can significantly reduce capacity.

Since CC-1 is a non-circulating current type cycloconverter, the input terminals of the positive group converter and negative group converter can be directly connected to the phase advance capacitor CAP, simplifying the element cooling configuration. Furthermore, a DC reactor for suppressing ripples in the circulating current can be omitted, and the device can be made smaller and lighter.

In addition, CC-2 is a circulating current type cycloconverter, and each converter is isolated by an isolation transformer on the input side, so it is possible to increase the number of control phases (number of control pulses) of the cycloconverter, which is effective. As described above, it becomes possible to output an output frequency that is equal to or higher than the frequency f _cap of the voltage applied to the phase advance capacitor CAP, and ultra-high-speed operation of the AC motor M becomes possible.

[Brief explanation of drawings]

FIG. 1 is a block diagram showing an embodiment of the AC motor drive device of the present invention, FIG. 2 is an equivalent circuit diagram for explaining the starting operation of the device in FIG. 1, and FIG. 3 is the operation of the device in FIG. 2. Time chart diagram for explaining the 4th
The figure is an equivalent circuit diagram for explaining the operation of controlling the frequency and phase of the voltage of the phase advance capacitor voltage of the device of Figure 1, Figure 5 is a time chart diagram for explaining the operation of Figure 4, and Figure 6 is a time chart diagram for explaining the operation of Figure 4. The figure is an equivalent circuit diagram for explaining the operating principle of Figure 1, Figure 7 is a block diagram showing a specific example of the control circuit diagram of the non-circulating current type cycloconverter of Figure 1, and Figure 8 is a diagram of the control circuit of the non-circulating current type cycloconverter in Figure 1. Fig. 7 is a time chart for explaining the operation, Fig. 9 is a block diagram showing a specific example of the control circuit of the circulating current type cycloconverter shown in Fig. 1, and Fig. 10 is an AC diagram of the device shown in Fig. 1. FIG. 3 is a vector diagram for explaining a control operation of an electric motor. R, S, T...3-phase AC power supply terminal, MTR...
Power transformer, CC-1...Non-circulating current type cycloconverter, CAP...Phase advance capacitor, CC-
2... Circulating current type cycloconverter, TR _U ,
TR _V , TR _W ...Isolation transformer, M...AC motor, L _SR , L _SS , L _ST ...AC reactor, SPR,
SPS, SPT... Positive group converter, SNR, SNS,
SNT……Negative group converter, L _OU1 , L _OU2 , L _OV1 ,
L _OV2 , L _OW1 , L _OW2 ...DC reactor, SPU,
SPV, SPW... Positive group converter, SNU, SNV,
SNW...Negative group converter, PG...Rotary pulse generator, CT _R , CT _S , CT _T , CT _U , CT _V , CT _W ...
Current transformer, PT _S , PT _cap ... Transformer, D... Rectifier circuit, _OSC ... 3-phase reference voltage generator, PHC ₁ , PHC ₂
...Phase control circuit, ACR ₁ , ACR ₂ ...Current control circuit, AVR ...Voltage control circuit, SPC ...Speed control circuit.

Claims (1)

[Claims] 1. An AC power source, a non-circulating current type cycloconverter having an output side terminal connected to the AC power source via a power transformer, and a converter connected to the input side terminal of the non-circulating current type cycloconverter. An AC motor consisting of a phase capacitor, a circulating current type cycloconverter whose input side terminal is connected to the phase advancing capacitor via an isolation transformer, and an AC motor connected to the output side terminal of the circulating current type cycloconverter. Drive device. 2. An AC power source, a non-circulating current type cycloconverter whose output side terminal is connected to the AC power source via a power transformer, a phase advance capacitor connected to the input side terminal of said non-circulating current type cycloconverter, and said phase advance capacitor connected to the input side terminal of said non-circulating current type cycloconverter. The non-circulating current cycloconverter consists of a circulating current cycloconverter whose input terminal is connected to a phase capacitor via an isolation transformer, and an AC motor connected to the output terminal of the circulating current cycloconverter. The circulating current type cycloconverter includes a control means for controlling the current supplied from the AC power source to be a sine wave having the same phase as the power supply voltage so that the voltage peak value of the phase advance capacitor is approximately constant. , an AC electric machine driving device comprising a control means for controlling the current supplied to the AC motor.