JPH05308345A - Delay lock loop device - Google Patents

Abstract

(57) [Summary] [Object] The present invention maintains the synchronization state of pseudo noise between the transmitting side and the receiving side without alienating the frequency utilization efficiency even if the carrier frequency of the received signal fluctuates due to the Doppler effect or the like. It is to provide a delay locked loop device capable of performing. [Structure] The spectrum components of both signals obtained by despreading the spread spectrum signal by pseudo noise having a phase advance and pseudo noise having a phase delay relative to the pseudo noise used when despreading the spread spectrum signal Since the shift clock frequency of the pseudo noise is controlled by the VCXO 33 in proportion to the difference between the maximum values of the DFTs 38 and 47 capable of widening the band, the reception frequency is affected by the Doppler effect or the like. Accurately maintains the synchronization of the pseudo noise between the transmitting side and the receiving side without deteriorating the S / N even when is changed.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention is applicable to the case where the carrier frequency of the received signal changes due to the Doppler effect, the frequency fluctuation of the transmitter oscillator, or the local oscillation frequency of the receiver side. The present invention relates to a delay lock loop device capable of accurately maintaining a synchronization state between pseudo noise used when demodulating a spread spectrum signal and pseudo noise on the transmission side.

[0002]

2. Description of the Related Art In a spread spectrum communication system, an information signal to be transmitted in a relatively wide frequency band or a carrier wave containing information is spread and transmitted, so that the transmission power per unit frequency is small and it interferes with other communications. It has many characteristics such as not only giving almost no noise but also being less susceptible to the influence of external noise and having excellent confidentiality. As a method for performing spread spectrum communication via a wireless communication path, the method shown in FIG. 2 is generally used.

That is, according to this communication method, on the transmitting side T, an oscillator 1 generates a carrier wave having a frequency f0, and a multiplier 2
The carrier information is modulated by the information data D to be transmitted, and the multiplier 3 is used to multiply-modulate the modulated signal with pseudo noise (hereinafter, referred to as a PN sequence) pt having a required bit length. After the spectrum of the data D is spread, the data D is sent to the receiving side R via the wireless communication path.

On the other hand, on the receiving side R, a variable frequency oscillator VC
While generating a local signal of frequency fL at O4,
The PN sequence generator 5 generates the same PN sequence pr as that used on the transmitting side T, and the spread spectrum signal transmitted from the transmitting side T and the local signal (local oscillation signal) are frequency-divided by a mixer 6. The spread spectrum signal is converted into an intermediate frequency signal having a frequency f0-fL (hereinafter, referred to as an intermediate frequency fI) by mixing, and then the data is demodulated in each of the data demodulation unit 7 and the delay lock loop device (hereinafter, DLL) 8. Output. The data demodulation unit 7 uses the multiplier 9 to despread the intermediate frequency signal with the PN sequence pr, and has a bandwidth based on the transmission rate of the information data D from the despread signal and its center frequency. Is fI
Of the original information data D by removing the intermediate frequency component of the frequency fI after the signal component of is extracted by the bandpass filter 10 and the detected signal is envelope-detected by the detector 11.
Is detected and the intermediate frequency component of the frequency fI is removed to demodulate the original information data D.

Further, the receiving side R obtains the absolute value of the demodulated information data D by using the level detector 12 at the time of data demodulation, and integrates this absolute value signal for each data cycle of the information data D. It is determined whether or not the signal level exceeds a predetermined value, and the PN is applied until the level exceeds the predetermined value.
The PN sequence generator 5 is controlled so as to sequentially shift the sequence pr bit by bit so that the PN sequences used in the transmitting side T and the receiving side R are synchronized with each other by a so-called sliding correlator. Try to.

Further, in this method, the PN sequence pt of the transmitting side T is
After synchronizing with the PN sequence pr of the receiving side R, the receiving side R uses the DLL 8 to adjust the shift clock frequency of the PN sequence generator 5 so as to maintain this synchronization state. That is, the DLL 8 uses the multiplier 13 to despread the intermediate frequency signal with the PN sequence pr + whose phase is one chip ahead of the PN sequence pr, and the transmission rate of the information data D from the despread signal. A bandpass filter 1 which has a bandwidth based on
After extraction in 4, the detector 15 envelope-detects the extracted signal and supplies the absolute value of the detected signal to one input terminal of the adder 16. Also, using the multiplier 17, the P
The intermediate frequency signal is despread by the PN series p- whose phase is delayed by one chip from the N series pr, and the bandpass filter 18 similar to the bandpass filter 14 and the detector 15 described above is used.
The detector 19 envelope-detects the inverse expanded signal and supplies the absolute value of the detected signal to the other input terminal of the adder 16. Further, the DLL 8 supplies the signal obtained by subtracting the output signal of the detector 19 from the output signal of the detector 15 by the adder 16 to the control terminal of the VCXO 21 via the low-pass filter 20 and supplies the clock signal oscillated by the VCXO 21 to PN. By supplying it to the input terminal of the sequence generator 5, the clock frequency of the VCXO 21 is controlled in proportion to the amplitude value of the output signal of the low-pass filter 20, and the PN sequence pr
And keeps the PN sequence pt synchronized.

However, in the DLL as described above, the transmitting side,
Alternatively, when the receiving side or both move at high speed, the frequency of the received signal fluctuates due to the Doppler effect. Therefore, the frequency of the signal despread with the PN sequence pr + and pr- is the bandpass filter 14 and the bandpass filter 14 respectively. There is a problem in that it is impossible to accurately maintain the synchronization of the PN sequence between the transmitting side and the receiving side outside the pass band of 18. The same problem also occurs when the frequency of the carrier wave generated at the transmission side or the local signal generated at the reception side fluctuates, especially the transmission speed of information data is slow and the frequency of the carrier wave is also low. Was extremely high. As a method for solving this problem, it is considered to increase the pass band width of the band pass filter by adding the frequency fluctuation of the carrier wave due to the Doppler effect or the like.
Is not practical because it deteriorates.

Conventionally, a method shown in FIG. 3 has been known as a method for solving this problem. In this method, the transmitting side T transmits a continuous wave CW having a predetermined frequency separately from a carrier wave for transmitting information data. On the other hand, on the receiving side R, a Doppler detector 13 for observing the reception frequency of the continuous wave CW is provided, and the frequency of the local signal generated by the VCO 14 is corrected based on the difference between the observation frequency and the specified frequency, and The frequency of the signal despread with the PN series pr + and pr- is always set to the center frequency of the bandpass filters 14 and 18.

However, the above DLL requires a transmitter and a receiver for measuring influences such as the Doppler effect in addition to the transmitter and the receiver for communicating information data, so that the device is very effective. In addition to the above, there is a drawback that the frequency utilization efficiency is deteriorated.

[0010]

SUMMARY OF THE INVENTION The present invention provides the Doppler effect as described above,
Or, it was made to solve the problem of DLL in the case where the carrier frequency of the transmitting side is accompanied, and the effect of the frequency variation is not accompanied by the enlargement of the device and without hindering the frequency utilization efficiency. PN between sender and receiver
It is an object of the present invention to provide a DLL capable of accurately maintaining sequence synchronization.

[0011]

SUMMARY OF THE INVENTION In order to achieve the above object, the present invention is configured as follows. That is, when performing spread spectrum communication, a first signal obtained by despreading a received signal with pseudo noise whose phase is ahead of the pseudo noise on the receiving side,
The shift clock frequency of the pseudo noise on the receiving side is controlled in proportion to the difference signal based on the second signal obtained by despreading the received signal with the pseudo noise whose phase is delayed from the pseudo noise on the receiving side,
In the means for holding the synchronization state of the pseudo noise between the receiving side and the transmitting side, the difference value between the maximum value of the spectral component of the first signal and the maximum value of the spectral component of the second signal is set. The shift clock frequency of the pseudo noise on the receiving side is proportionally controlled to maintain the synchronized state of the pseudo noise on the receiving side and the transmitting side.

[0012]

The present invention will be described in detail below with reference to the illustrated embodiments. FIG. 1 is a block diagram showing an embodiment of a spread spectrum signal demodulating device equipped with a DLL according to the present invention. Note that there is no difference on the transmitting side from that shown in FIG.
Here, duplicate description is omitted.

In the figure, W indicates the entire demodulation device, which is a VCO 22 as a local oscillator for generating a local signal of frequency fL, and the same PN sequence pr 'as the transmitting side.
And a PN sequence generator 23 for generating a signal, and the mixer 24 frequency-mixes the spread spectrum signal S having the carrier frequency f0 sent from the transmitting side with the local signal having the frequency fL to generate the spread spectrum signal S as a frequency. fI
To the input terminal of the PN sequence generator 23 via the data demodulator 25 and the level detector 26. Here, the data demodulation unit 25 includes a multiplier 27, a bandpass filter 28, and a detector 29 similar to the above-mentioned conventional one, which are connected so as to demodulate the original information data D from the intermediate frequency signal. To do. The level detector 26 also integrates the absolute value signal of the demodulated information data D for each data cycle of the information data D in the same manner as described above to determine whether the signal level exceeds a predetermined value. P so that the PN sequence pr ′ is sequentially shifted bit by bit until the level exceeds a predetermined value.
The N sequence generator 23 is controlled to connect the PN sequences used on the transmitting side and the receiving side so as to synchronize with each other.

Further, the demodulator W uses the intermediate frequency signal DL
The shift clock supplied to the L30 and output from the DLL 30 is connected to the clock input terminal of the PN sequence generator 23. The DLL 30 includes a local oscillator 31 for generating a local signal of frequency fI, a phase shifter 32 for delaying the phase of the local signal by π / 2 [rad], and a period of the PN sequence pr ′ as information data. And a VCXO 33 for generating a shift clock required to make the data transmission cycle the same as that of D, and a PN sequence pr '+ whose phase is advanced by one chip by the multiplier 34 by the multiplier 34. The frequency signal is despread, and the despread signal R + and the local signal of the frequency fI are mixed in frequency by the mixer 35. The real number component signal I of the despread signal SR1 in the baseband signal is obtained.
1 is supplied to the A / D converter 37 via the anti-aliasing low pass filter 36, and the digital signal is
screte Fourier Transform
A device (hereinafter referred to as DFT) 38 is supplied to one input end. Further, the imaginary number component signal Q1 of the despread signal SR1 obtained by branching the despread signal R + and mixing the frequency of the despread signal R + and the phase-delayed local signal of the frequency fI in the mixer 39 is obtained. The signal is supplied to the A / D converter 37 via the aliasing removing low-pass filter 40, and the digital signal thereof is supplied to the other input end of the DFT 38. The DFT 38, to which the two virtual components have been input, calculates the frequency component of the despread signal SR1 and supplies the frequency component to the calculator 41. Here, the spectrum value is obtained and the maximum spectrum value M1 thereof is supplied to one of the adders 42. Supply to the input end.

On the other hand, the intermediate frequency signal is branched, and the multiplier 43 despreads the intermediate frequency signal with the PN sequence pr'- whose phase is delayed by one chip from the PN sequence pr ', and the despread signal R thereof. -And the local signal of the frequency fI are mixed by the mixer 44 in frequency, and the real number component signal I2 of the despread signal SR2 in the baseband signal obtained through the aliasing elimination low-pass filter 45
The digital signal is supplied to the D / D converter 46 and the digital signal is supplied to the second D
It is supplied to one input terminal of the FT 47. An imaginary term component signal of the despread signal SR2 obtained by branching the despread signal R- and mixing the frequency of the despread signal R- and the phase-delayed local signal of the frequency fI in the mixer 48. Q
2 is supplied to the A / D converter 46 via the anti-aliasing low pass filter 49, and its digital signal is supplied to the other input terminal of the DFT 47. The DFT 47, to which the two virtual components are input, calculates the frequency component of the despread signal SR2 and supplies the frequency component to the calculator 50. Here, the spectrum value is obtained and the maximum spectrum value M thereof is obtained.
2 is supplied to the other input terminal of the adder 42.

Further, the DLL 30 adds a control signal proportional to the difference value signal obtained by subtracting M2 from the maximum spectrum value M1 by the adder 42 to the VC through the low pass filter 51.
It is supplied to the control terminal of XO33 and its VCXO3
The output clock signal No. 3 is supplied to the clock input terminal of the PN sequence generator 23, and the PN sequence on the transmitting side and the PN sequence pr 'are held in synchronization.

The demodulating device thus configured operates as follows. The operations other than the DLL 30 are almost the same as those described in the prior art, so a detailed description thereof will be omitted. However, since the despread signals SR1 and SR2 demodulate the information data D respectively as described above. PN series p
The level detector 2 is a signal obtained by despreading an intermediate frequency signal having a frequency fI by a PN sequence pr '+ whose phase is advanced by one chip and a PN sequence pr'- which is delayed by one chip from r'.
At 6, the levels of both signals after the synchronization acquisition of the PN sequence between the transmitting and receiving sides are close to the autocorrelation value of the PN sequence pr '. Therefore, when both signals are mixed with the local signals of the same frequency in the mixers 35, 39, 44 and 48, respectively, the frequency signal of the difference between them is output, but when both the frequencies are the same, they are transmitted. A data waveform is obtained.

However, when the reception frequency fluctuates due to the Doppler effect and changes to fI + ΔfI, the mixers 35, 39, 44 and 48 output data signals as well as data signals.
The change ΔfI is included.

Therefore, the DLL 30 uses the DFT 38 during the bit period of the information data D, that is, while the bit value keeps the state value of 0, 1 or -1, and the real number component signal I.
1 and the imaginary number component signal Q1 to obtain the real number component value X1 and the imaginary number component value Y1 of each frequency component, and further, the calculator 41
Then, the maximum absolute value M1 of the frequency spectrum component of the despread signal SR1 in the baseband is obtained. or,
Similarly, using the DFT 47, the real number component value X2 of each frequency component is calculated from the real number component signal I2 and the imaginary number component signal Q2.
And the imaginary number component value Y2 are calculated, and the maximum value M2 of the absolute value of the frequency spectrum component of the despread signal SR2 in the baseband is calculated by the calculator 50.

That is, the frequency bandwidth Bx [Hz] of the spectrum that can be calculated by the DFTs 38 and 47 is vmax [m / s] as the maximum relative speed between the transmitting side and the receiving side, and c [m / s] as the light speed. ], The following equation is obtained: Bx ≧ f0 · vmax / 2c [Hz]. Therefore, in the range where the frequency deviation ΔfI due to the Doppler effect or the like satisfies Bx ≧ ΔfI, the maximum value of the absolute value of the frequency spectrum component is obtained. Since the differential signal obtained by subtracting M2 from M1 can be accurately obtained and the frequency bandwidth Bx can be made very wide, the frequency of the output clock signal of the VCO 33 is controlled in proportion to the differential signal. , By varying the shift clock frequency of the PN sequence generator 23, the synchronization state between the PN sequence on the transmitting side and the PN sequence pr 'can be accurately controlled.

In the above embodiment, the PN series pr '+ whose phase is advanced by one chip and the PN series pr- delayed by one chip from the PN series pr' supplied to the data demodulation unit are used. Not limited to this, if a PN sequence whose phase leads the PN sequence pr 'and a PN sequence delayed from it are used,
It will be apparent that the PN sequence synchronization state can be maintained between the transmitting side and the receiving side as described above. Further, in the above-described embodiment, the despreading operation is performed after converting the carrier frequency of the spread spectrum signal transmitted from the transmitting side to the intermediate frequency signal, but the present invention is not limited to this, and the spread spectrum signal is applied to the spread spectrum signal. It is obvious that the despreading operation may be directly performed by using the above method.

[0022]

As described above, according to the present invention, the spread spectrum signal is divided into a PN sequence whose phase is advanced and a PN sequence whose phase is delayed as compared with the PN sequence used when despreading the spread spectrum signal. DF capable of widening the frequency band of the spectral components of both despread signals
It is calculated by T, and P is calculated in proportion to the difference value of each maximum value.
Since the N-series shift clock frequency is controlled, even if the reception frequency or the like changes due to the Doppler effect or the like, S /
This is effective in providing a DLL capable of accurately maintaining synchronization of PN sequences on the transmitting side and the receiving side without deteriorating N.

[0023]

[Brief description of drawings]

FIG. 1 is a configuration diagram showing an embodiment of the present invention.

FIG. 2 is a diagram illustrating a conventional DLL.

FIG. 3 is a diagram illustrating another conventional DLL.

[Explanation of symbols]

23 PN sequence generator, 30 DLL, 31 local oscillator, 32 phase shifter, 33 VCXO, 34 and 43 multiplier, 35, 39, 44 and 48 mixer, 36, 4
0, 45, 49 and 51 low pass filter, 37 and 46 A / D converter, 38 and 47 DFT, 41 and 50 arithmetic unit, 42 adder.

Claims (1)

[Claims] 1. When performing spread spectrum communication, a first signal obtained by despreading a received signal with pseudo noise whose phase is ahead of the pseudo noise on the receiving side and a phase delay from the pseudo noise on the receiving side. The shift clock frequency of the pseudo noise on the receiving side is controlled in proportion to the differential signal based on the second signal obtained by despreading the received signal with the pseudo noise, and the synchronization state of the pseudo noise on the receiving side and the transmitting side is controlled. In the holding means, the shift clock frequency of the pseudo noise on the receiving side is proportional to the difference value between the maximum value of the spectral component of the first signal and the maximum value of the spectral component of the second signal. A delay locked loop device characterized by being controlled so as to maintain a synchronized state of pseudo noise between the receiving side and the transmitting side.