JPH0833314A - Load drive - Google Patents

Abstract

(57) [Summary] [Object] To obtain a control voltage higher than a power supply voltage with a simple and highly reliable circuit configuration and to enable an N-type semiconductor element to be used as a switching means of a high-side switch. . A circuit for controlling an average current by turning on / off a power supply voltage (VB) applied to a load (40) by a switch means (Sw1) at a predetermined ratio, and when the switch means (Sw1) is on. The capacitor (C1) is charged by the potential difference generated between the load (40) and the power supply when turned off, and the charge for charging is applied to the drive circuit (20) to exceed the power supply voltage (VB). .

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a load driving device for driving a load by intermittently applying a voltage to a load or controlling a current, and more particularly to a load driving device having a simple structure and high reliability.

[0002]

2. Description of the Related Art Conventionally, as a load driving device for driving an inductive load, for example, a solenoid plunger or a coil of a stepping motor, a circuit opening / closing means such as a semiconductor switch is inserted between the load and a power source. A device for controlling the average voltage or current applied to the load by opening and closing is used. This device is known as chopper control or PWM (pulse width modulation) control, and its typical conventional circuit configuration is shown in FIGS. 11 and 12.

The circuit shown in FIG. 11 has a circuit configuration called a high side switch for switching current on the power supply side of the load, and the circuit shown in FIG. 12 is a low side switch for switching current on the ground side of the load. This is the called circuit configuration.

FIG. 11 is constructed by including two transistors TR1 and TR2 which are switching means and two resistors R0 and R1. A PWM signal having a predetermined duty ratio is applied to the base of the transistor TR1 and the transistor T1 is added.
The power source is connected to the emitter of R2, and the transistor TR2
A resistor R0 is connected between the emitter and the base of the transistor TR1, a resistor R1 is connected between the collector of the transistor TR1 and the base of the transistor TR2, and the emitter of the transistor TR1 is grounded. Further, a flywheel diode FD is connected in parallel to an inductive load L which is a drive target of this circuit, a cathode of the flywheel diode FD is connected to a collector of the transistor TR2, and an anode of the flywheel diode FD is grounded. .

In such a configuration, when the PWM signal becomes high level and the transistor TR1 is turned on, the transistor TR2 is turned on in response to this, and the power source is applied to the load L via the transistor TR2.
A load current flows to the ground via the transistor TR2 and the load L. At this time, this load current increases with time due to the characteristics of the load L, and finally becomes constant at the saturation point given by the resistance component (not shown) in the inductive load L and the power supply voltage.

By the way, when the chopper control is performed for the purpose of controlling the average current flowing through the load, the average current value required for the load L is lower than the current value at the saturation point, and therefore, the load current flowing through the load L. Before reaching its saturation point, the PWM signal changes from high level to low level, the transistor TR1 changes from on to off, the transistor TR2 also changes from on to off, and the power supply is cut off from the load L.

Here, the on / off timing of the transistor TR2, that is, the high-level and low-level timing of the PWM signal is, for example, on / off ratio control (duty control) of the transistor TR2 based only on time or not shown. Load L by current detector
Is determined by a predetermined reference such as constant current control based on the instantaneous value or the average value of the load current flowing through the load current.

In either case, when the transistor TR2 is turned off and the power supply is cut off from the load L, the load current flowing through the load L is circulated through the flywheel diode FD by the inductive component of the load L, It gradually decreases due to the electric resistance of the circuit and the forward voltage of the flywheel diode FD.

In this state, when the PWM signal changes from the low level to the high level again, the transistor TR1 is turned on, and the transistor TR2 is turned on, the power source is connected to the load L again and flows through the load L as described above. The load current gradually increases.

As described above, in the high side switch, the transistor TR2 is turned on and off by the on / off timing of the PWM signal applied to the base of the transistor TR1.
It is possible to control the load current flowing through the load L by changing the on / off timing of the.

The advantages of this high side switch are as follows: (1) Since the load L is on the ground side of the switch (transistor TR2) and no voltage is constantly applied to the load L, it is safe even if there is a short circuit in the non-driven state. Also, it is on the safe side against electrolytic corrosion due to humidity.

(2) The load L can be cut off by the switch (transistor TR2) even if the wiring of the load is short-circuited while the load is being driven.

(3) The return wiring from the load L can be omitted.

And so on.

On the other hand, a drawback of this high-side switch is that P-type elements such as PNP bipolar transistors, P-channel FETs and P-channel IGBTs must be used due to the nature of the circuit. P-type devices are generally NPN transistors, N-channel FETs, N-channel I
Compared with N-type elements such as GBT, current gain, withstand voltage,
It has inferior characteristics in many aspects such as saturation voltage, and is expensive. Therefore, the efficiency of the switch circuit is poor and the economy is impaired. If an N-type element such as an NPN transistor or an N-channel FET is used instead of a P-type element to form a circuit, the circuit normally becomes an emitter follower or source follower circuit, and the base-emitter voltage or gate is used. -The source voltage directly affects the collector-emitter voltage or the drain-source voltage, resulting in poor efficiency.

As a method of preventing this, in a high side switch circuit using a circuit of an emitter follower and a source follower of an N type element such as an NPN transistor and an N channel FET, is a drive power supply for driving a base or a gate provided independently of the main circuit? Alternatively, a circuit configuration is adopted in which the driving power source is boosted above the main power source voltage by an amount corresponding to the base-emitter voltage or the gate-source voltage.

However, since a large number of circuit elements are required to form such an insulated power supply and a booster circuit, the cost of the apparatus increases accordingly, and the failure rate increases as the number of elements increases, resulting in a circuit failure. The reliability will be reduced.

On the other hand, FIG. 12 shows a circuit configuration called a low side switch for switching current on the ground side of the load.

In this circuit, a transistor TR1 which is a switch means is provided on the ground side of the load, and the transistor T1 is provided.
The circuit can be configured more simply than the high-side switch because it can be configured by adding a PWM signal having a predetermined duty ratio to the base of R1.

When the PWM signal becomes high level and the transistor TR1 is turned on, a load current flows from the power source to the ground via the load L and the transistor TR1. At this time,
This load current increases with time due to the characteristics of the load L,
Finally, it becomes constant at the saturation point given by the resistance component (not shown) in the inductive load L and the power supply voltage. The method of chopper control for holding the average current value required for the load L lower than this saturation point is basically the same as that described for the high side switch.

This method can be realized with a configuration simpler than that of the high-side switch, an N-type semiconductor element can be used as a switch element, and the drive voltage of the N-type semiconductor element is always with reference to the ground. Although it excels because the potential of the source is constant, there is a problem that the current cannot be interrupted when a part of the load is short-circuited with the ground side.

In addition to this, as a conventional example, Japanese Patent Laid-Open No. 5-579.
There are eighteen. In this example, the configuration is as shown in FIG. 13, and the power source E2 is connected to the gate of the N-channel FET Sw1 via the resistor R1 and the diode D1 and the transistor switch element TR1 is provided. Further resistance R
1, the capacitor C1 is provided at the connection point of the diode D1,
The transistor switch element TR1 is charged when it is on and discharged when it is off to give a charge to the gate. This allows N-channel FET on the power supply side of the load.
Can be controlled.

However, as can be seen from FIG. 13, a separate power source E2 is required to give an electric charge to the gate, and when the transistor switch element TR1 is on, this power source E
In addition to the current that charges the capacitor C1 from 2
1 is not always efficient because a current also flows to the ground through the transistor 1 and the transistor switch element TR1. If the resistance value is increased in order to reduce the current flowing through the resistor R1, the charging resistance between the gate and the source when the N-channel FET Sw1 is turned on increases, and F
There is a problem that the switching speed of ET decreases.

In addition to the current control by digital switching as described above, there is a method of directly controlling the current amount in an analog manner.

FIG. 14 is a block diagram of a conventional analog control type constant current circuit.

The circuit includes a current detection circuit 1, a drive control circuit 2 for controlling the current by a method of changing the base voltage of the transistor Tr1 according to the current detection value of the current detection circuit 1, a bipolar transistor Tr1 and a power supply. It consists of 3. Compared with the chopper control, this method has no problems such as unnecessary radiation of electric waves from the output wiring and generation of induced noise. Further, in principle, even if there is a short circuit in the wiring of the load during load driving, constant current control can be performed without special means. However, in actuality, the current control element causes the load current to flow while the potential difference between the power supply voltage and the output terminal voltage is being applied, resulting in a large loss of the element and dissipating the amount of heat generated by the element at this time as a device. Will be required, and problems will occur in terms of cost and reliability.

[0027]

As described above, in the conventional inductive load drive device, in the case of current control by digital switching, a high side switch that is safe against a short circuit during operation of the load is provided with a P-type element. If it is configured by using it, the circuit configuration is simple, but the unit price of the element is high and the element characteristics are poor, and if it is configured using N-type elements with excellent element characteristics, the number of elements will increase and a separate power supply will be required. In any case, there was a problem in terms of price and reliability, such as a decrease in switching speed. Also in the case of analog current control, the element loss during operation is large, and the element having a large allowable loss is required, and the device is required to take heat dissipation measures.

Therefore, the first object of the present invention is to construct an efficient high-side switch by using N-type elements with relatively few circuit components. A second object is to control the voltage applied to the element of the analog control type constant current circuit while maintaining the constant current control so that the heat generation amount of the element is reduced as much as possible.

[0029]

In order to achieve the above object, the present invention provides a load drive device having a current control element provided between a load and a power supply and a drive circuit for driving the current control element, The current control element is switch means for turning on / off the voltage of the power supply applied to the load, and the drive circuit is charged by a potential difference generated between the load and the power supply when the switch means is off. It is characterized by comprising a capacitor and a control power supply circuit for applying a control voltage exceeding the voltage of the power supply to the drive circuit when the switch means is turned on by the charge charged in the capacitor.

Further, in the load driving device, the switch means is charged by a first capacitor charged by a potential difference generated between the load and the power source when the switch means is off, and the charge of the first capacitor. Control voltage that exceeds the voltage of the power supply to the drive circuit when the switch is turned on, and a potential difference generated between the load and the power supply when the switch means is turned off from the charging voltage of the first capacitor. And a second capacitor for charging the first capacitor when the voltage for control by the first capacitor is lowered by a high charging voltage. .

Further, in a load drive device having a current control element provided between the load and the power supply and a drive circuit for driving the current control element, the current control element varies the current applied to the load. A current varying means, wherein the driving circuit obtains a potential difference before and after the current varying means, and a power source for controlling the voltage of the power source so as to reduce the potential difference according to the detection result of the potential difference measuring means. And a voltage control means.

[0032]

In the first aspect of the present invention, it is not necessary to provide a drive power supply for driving the current control element independently of the main circuit, and a control voltage higher than the power supply voltage can be obtained with a simple circuit configuration. . Moreover, since the number of circuit elements is small, the reliability of the circuit is high.

Further, in the second aspect of the invention, the voltage applied to the element of the constant current circuit is controlled to be as small as possible while maintaining the constant current control, and the heat generation amount of the element is reduced.

[0034]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS An inductive load drive device according to the present invention will be described in detail below with reference to the accompanying drawings.

FIG. 1 shows a block diagram of the principle of the first invention of the present invention, and FIG. 2 shows a circuit diagram of an embodiment of the first invention.

The outline of the first invention of the present invention will be described with reference to FIG.

In FIG. 1, Sw1 is a switch element, C1 is a capacitor, 10 is a charging circuit, 20 is a drive circuit, 30 is a drive signal, 40 is a load, and VB is a power supply.

The switch element Sw1 is turned on / off by the drive circuit 20 according to the drive signal 30.
Consider now that the switch element Sw1 is off. Then, a potential difference is generated between the terminals a and b, and the charging circuit 10 charges the capacitor C1 based on the potential difference. The charging circuit 10 includes means for managing the charging voltage of the capacitor C1 and means for blocking the charge stored in the capacitor C1 from flowing to the power source VB side. Here, the voltage due to the electric charge stored in the capacitor C1 is used for the power supply of the drive circuit 20.

Next, the drive signal 30 inputs an ON signal to the drive circuit 20, and the drive circuit 20 causes the switch element Sw1 to be turned on.
Is turned on. Then, the terminals a and b have almost the same potential. On the other hand, since the stored electric charge of the capacitor C1 is blocked from flowing to the power source VB side, the capacitor C1 continues to supply a voltage higher than those of the terminals a and b as the power source of the drive circuit 20 due to the stored electric charge. be able to.

With the configuration shown in FIG. 1, a high-side switch using an N-type element is configured without using a second power source or a complicated circuit and without impairing the reliability of the device with a small number of circuit components. it can.

FIG. 2 shows a circuit diagram of an embodiment of the first invention based on such an idea. This embodiment is an improved version of the high side switch using N type elements.

In FIG. 2, C1 is a capacitor, D1 is a diode, L is a load, Q1 is an inverter circuit, R1 and R2 are resistors, Sw1 is a switch element (here, FET), and Tr.
1, Tr2 is an NPN transistor, and ZD1 is a Zener diode.

Consider the case where the load L is not driven. Terminal a
The power source VB is applied between the terminal and the terminal c (ground).
At this time, since a potential difference is generated between the terminals a and b,
A voltage equal to the Zener voltage of the Zener diode ZD1 is charged in the capacitor C1. This charging voltage becomes the drive power supply voltage of the inverter circuit Q1 and the inverter circuit Q1
Becomes active.

Next, consider the case where the load L is driven. At this time, a drive signal (negative pulse signal) for driving the load is applied to the terminal d or the transistor Tr surrounded by the dotted line
A positive pulse signal is applied to the base of No.2. As a result, the output of the inverter circuit Q1 becomes high, and this high voltage is applied between the gate and source of the FET Sw1. As a result, the FET Sw1 is turned on, the voltage at the point b becomes substantially equal to the voltage VB at the terminal a, and the load L is driven by the power supply VB.

At this time, the charge stored in the capacitor C1 is prevented from flowing out to the power supply side by the diode D1, so that the positive side of the capacitor C1 maintains the same voltage as the Zener voltage of the Zener diode ZD1. Then, the supply of the drive power supply voltage of the inverter circuit Q1 is continued. Therefore, the driving voltage higher than the source potential by the charging voltage of the capacitor C1 is continuously supplied to the gate of the FET Sw1. As a result, FETSw
1 can be sufficiently saturated, and FET Sw1
The loss inside can be reduced.

By configuring the circuit as described above, an NPN transistor, N
Even if an N-type element such as a channel FET or an N-channel IGBT is used, the high side switch can be configured without increasing the number of circuit elements, and an efficient inductive load drive device can be realized.

Since the circuit of this embodiment adopts a configuration in which the power source of the inverter circuit Q1 is charged when the load L is not driven, it is suitable for a circuit application in which the load is repeatedly turned on / off, and a capacitor is used. The capacitance of C1 and the impedance of the drive circuit of the inverter circuit Q1 continuously cause F
The time during which the ETSw1 can be maintained in the ON state is determined, and the load drive time ratio (duty ratio) of the FET Sw1 is determined by the charging current of the capacitor C1 and the impedance of the drive circuit of the inverter circuit Q1.

FIG. 3 shows a circuit diagram of another embodiment of the first invention. This circuit has a configuration in which another set of capacitors and a charging circuit are added to the circuit shown in FIG. With this circuit, the Zener voltage V of the Zener diode ZD1
_ZD1 is the Zener voltage V of the Zener diode ZD2
_It is larger than _ZD2 and V _ZD1 > V _ZD2 .

Now, consider the case where the load L is not driven.
The power supply VB is applied between the terminals a and c (ground). At this time, since a potential difference is generated between the terminals a and b, the Zener voltage V of the Zener diode ZD1 is
_A voltage equal to _ZD1 charges capacitor C1. At the same time, the Zener voltage V of the Zener diode ZD2
Voltage equal to _{ZD 2} is charged in the capacitor C2. The charging voltage of the capacitor C2 becomes the driving power supply voltage of the inverter circuit Q1 and the inverter circuit Q1 is in the operating state.

Next, consider the case where the load L is driven. At this time, a drive signal (negative pulse signal) for driving the load is applied to the terminal d and input to the inverter circuit Q1, or a positive pulse signal is applied to the base of the transistor Tr2 surrounded by the dotted line. As a result, the output of the inverter circuit Q1 becomes high, and this high voltage is applied between the gate and source of the FET Sw1. As a result, the FET Sw1 is turned on, the voltage at the point b becomes substantially equal to the voltage VB at the terminal a, and the load L is driven by the power supply VB.

At this time, the charges stored in the capacitors C1 and C2 are prevented from flowing out to the power source side by the diodes D1 and D2, respectively, so that the positive side of the capacitor C1 is a zener diode. The potential equal to the Zener voltage V _ZD1 of _ZD1 is maintained as it is, and the positive side of the capacitor C2 is maintained to the potential equal to the Zener voltage V _ZD2 of the Zener diode ZD2.

The drive power supply voltage of the inverter circuit Q1 is continuously supplied by the potential of the capacitor C2. That is, the driving voltage higher than the source potential by the charging voltage of the capacitor C2 is continuously supplied to the gate of the FET Sw1. As a result, the FET Sw1 can be sufficiently saturated during driving the load, and the loss in the FET Sw1 can be reduced.

If the potential of the capacitor C2 becomes lower than the zener voltage V _ZD2 due to the power consumption of the inverter circuit Q1 or the like, the electric charge of the capacitor C1 is passed through the transistor Tr2 and the diode D2 to the capacitor C2.
, And the positive side of the capacitor C2 acts to maintain a potential equal to the Zener voltage V _ZD2 of the Zener diode ZD2. If the losses of the transistors Tr1 and Tr2 and the diodes D1 and D2 are ignored, the potential of the capacitor C2 is kept at a value equal to the Zener voltage V _ZD2 until the time when the potential of the capacitor C1 becomes equal to the potential of the capacitor C2. Since the power supply voltage of the inverter circuit Q1 is constant while the potential of the capacitor C2 is maintained, the increase in loss at Sw1 due to the generally known FET gate voltage drop does not occur.

In FIG. 3, the charging circuit and the capacitor are 2
Although the number of sets is two, and the number of sets of the charging circuit and the capacitor is more than one, the power supply voltage of the inverter circuit Q1 can be kept constant for a longer period of time. Needless to say.

Although the inverter circuit is used as the drive circuit in FIGS. 2 and 3, the polarity of the pulse of the drive signal may be inverted to use the buffer circuit. Alternatively, an operational amplifier circuit, a comparator circuit, or the like can be used as the driving circuit.

FIG. 4 is a circuit example of a PWM control circuit using the first embodiment of the first invention shown in FIG. By adding a PWM signal to the base or terminal d of the transistor Tr2 and turning on / off the FET Sw1 at a timing by a current detector (not shown) or the like, the average value of the current flowing through the load L can be controlled. FD is a flywheel diode.

FIG. 5 shows a so-called circuit between the inverter circuit Q1 and the gate of the FET Sw1 in order to lower the impedance between the gate and the source of the FET Sw1 to increase the switching speed in the circuit of the embodiment of the present invention shown in FIG. In this embodiment, a complementary circuit 50 is provided. In the present invention, since the charging circuit including the capacitor and the driving circuit are independent of each other, such a configuration can be easily realized and high-speed switching operation is possible.

As described above, according to the first aspect of the present invention, Since the capacitor is charged by the potential difference generated between the load and the power source when the switch means is off, and the charged electric charge is used as the power source of the drive circuit, a separate power source for driving the switch means is not required.

2. Since the capacitor is automatically charged / discharged according to ON / OFF of the switch means, an element for switching the charging / discharging of the capacitor is not required.

3. Since the capacitor and the drive circuit are independent, it is easy to insert a complementary circuit or the like that realizes low impedance drive of the switch means.

4. Since the charging voltage of the capacitor can be accurately determined by the charging circuit, it is not necessary to set it appropriately higher, and since the charging operation is stopped when the specified voltage is reached, there is no extra charging and Minute efficiency.

It has excellent features such as.

Moreover, the circuit is simple, the loss is small, the operation is stable, and it is economical.

Next, FIG. 6 shows a block diagram of an embodiment of the second invention.

As shown in FIG. 6, the circuit includes a current detection circuit 1, a drive control circuit 2, a drive bipolar transistor Tr1, a power supply 3, and Vce detection for detecting a collector-emitter voltage of the transistor Tr1. It has a power supply voltage control circuit 5 for controlling the output voltage of the power supply 3 by the detection outputs of the circuit 4 and the Vce detection circuit 4.

Now, consider the case where this circuit is performing a constant current control operation. Assuming that the power supply 3 supplies the transistor Tr1 with the minimum voltage required to cause a predetermined current to flow through the load L, the collector-emitter voltage drop Vce of the transistor Tr1 becomes small. If the voltage of the power supply 3 becomes larger than the minimum voltage required to flow a predetermined current through the load L, the large voltage appears in the collector-emitter drop voltage Vce of the transistor Tr1. When the voltage Vce becomes large, this is detected by the Vce detection circuit 4, and the power supply voltage control circuit 5 is activated by the output of the Vce detection circuit 4 to control so as to lower the power supply voltage.
The voltage of the power supply 3 is set to operate at the lowest voltage required to flow a predetermined current through the load L, that is, the voltage Vce is always the lowest value. By operating the circuit in this way, heat generation in the drive transistor Tr1 can be suppressed to the minimum. This can reduce the heat radiation cost of the device and improve the reliability of the circuit.

FIG. 7 shows another embodiment of the second invention of the present invention, in which the Vce detection circuit 4 portion of FIG.
There is a circuit shown in 2.

When this circuit is performing a constant current control operation, when the power supply voltage is equal to the minimum voltage required to supply a predetermined current to the load L, the voltage drop Vce between the collector and the emitter of the transistor Tr1 is small. Therefore, by comparing the voltage Vbe between the base and the emitter for turning on the transistor Tr2 and Vce, Vce of the transistor Tr1 ≦ Vce of the transistor Tr2
If it is be, the transistor Tr2 is not turned on.

However, in this state, when the impedance of the load L changes and decreases (for example, when the temperature of the load is gradually decreased by the solenoid plunger), a predetermined current is applied to the load L. The power supply voltage becomes higher than the minimum voltage required to supply the transistor T
Vce of r1 increases. Then, this voltage causes a voltage Vb between the base and the emitter which turns on the transistor Tr2.
When e is exceeded, the transistor Tr2 is turned on. The power supply voltage control circuit 5 receives the output signal generated when the transistor Tr2 is turned on, and lowers the voltage of the power supply 3.

By this series of operations, the power supply voltage is lowered, the collector-emitter voltage drop Vce of the transistor Tr1 can be lowered again, and the transistor T1 can be lowered.
The heat generation of r1 can be suppressed.

The above is the case where the transistor Tr1 is on. However, in this circuit, the collector-emitter voltage Vce increases even when the transistor Tr1 is off due to an output control signal from the outside. This prevents the circuit from lowering the power supply voltage.
The output control signal sent to the drive control circuit 2 for driving the
When 1 is in the cutoff state, the power supply voltage adjusting function of the power supply voltage control circuit 5 is stopped. In this case, the power supply voltage is output as it is immediately before the transistor Tr1 is turned off by the output control signal, or is fixed to a predetermined voltage. This selection can be determined by the power supply voltage control circuit 5 as needed.

In the embodiments shown in FIGS. 6 and 7, the case where the bipolar transistor is used as the transistor Tr1 has been described, but the same effect can be obtained even if it is a general current control element transistor. . For example, FETs, IGBTs, SITs, etc. can be used for this purpose.

In the embodiments shown in FIGS. 6 and 7, the circuit has been described with the N-type element such as the NPN transistor as the transistor Tr1, but the P-type element such as the PNP transistor may be used as shown in FIG. It goes without saying that the same effect can be obtained.

Further, in the embodiment shown in FIGS.
Although the case where a bipolar transistor is used as r2 has been described, a voltage detection element OP such as an operational amplifier or a comparator may be used as shown in FIG. Also,
As shown in FIG. 10, by providing a voltage dividing resistor between the base and the emitter of the transistor Tr2, the voltage drop Vce between the collector and the emitter of the transistor Tr1 in operation can be arbitrarily set at a voltage higher than the voltage (Vbe) for turning on the transistor Tr2. It can be set.

[0075]

As described above, in the first invention, in the load drive device having the current control element provided between the load and the power supply and the drive circuit for driving the current control element, the current control element is Turn on / off the power supply voltage applied to the load
This drive circuit is a switch means for turning off, and this drive circuit is for controlling a capacitor charged by a potential difference generated between a load and a power source when the switch means is off, and a voltage exceeding the power source voltage when the switch means is on due to the charge charged in the capacitor. And a control power supply circuit for supplying a voltage to the drive circuit.

As a result, it is not necessary to provide a drive power supply for driving the current control element independently of the main circuit, and a control voltage higher than the power supply voltage can be obtained with a simple circuit configuration. Therefore, the high-side switch can be easily configured by using the N-type semiconductor element which is cheaper and has better performance than the P-type element. Moreover, since the number of circuit elements is small, high reliability can be obtained.

Further, in the second invention, in the above-mentioned load driving device, the current control element is a current varying means for varying the current applied to the load, and the potential difference detecting means for detecting the potential difference generated in the current varying means. And a power supply voltage control means for controlling the voltage of the power supply so as to reduce the potential difference according to the detection result of the potential difference detection means. As a result, the voltage applied to the element of the constant current circuit is controlled to be as small as possible while maintaining the constant current control, so that the heat generation amount of the element can be reduced and the heat dissipation cost of the device can be reduced. The reliability of the circuit can be improved.

[Brief description of drawings]

FIG. 1 is a block diagram showing the principle of the first invention of the present invention.

FIG. 2 is a circuit diagram showing an embodiment of a load driving device according to the first invention of the present invention.

FIG. 3 is a circuit diagram showing another embodiment of the load driving device according to the first aspect of the present invention.

4 is a circuit diagram of a PWM control circuit using the embodiment of the present invention shown in FIG.

FIG. 5 is a circuit diagram showing another embodiment of the load driving device according to the first aspect of the present invention.

FIG. 6 is a block diagram of an embodiment of a load driving device according to a second invention of the present invention.

FIG. 7 is a block diagram of another embodiment of the second invention of the present invention.

FIG. 8 is a block diagram of yet another embodiment of the second invention of the present invention.

FIG. 9 is a block diagram of still another embodiment of the second invention of the present invention.

FIG. 10 is a block diagram of yet another embodiment of the second invention of the present invention.

FIG. 11 is a circuit diagram of a conventional example of a load driving device.

FIG. 12 is a circuit diagram of another conventional example of the load driving device.

FIG. 13 is a circuit diagram of still another conventional example of the load driving device.

FIG. 14 is a block diagram of still another conventional example of the load driving device.

[Explanation of symbols]

C1, C2 capacitor D1, D2 diode E1, E2 power supply FD flywheel diode L, P, 40 load OP voltage detection element Q1 inverter circuit R0, R1, R2, R3 resistance Sw1 switch element Tr1, Tr2, Tr3, Tr4, TR1, TR2 transistor ZD1, ZD2 Zener diode VB power supply voltage 1 current detection circuit 2 drive control circuit 3 power supply 4 Vce detection circuit 5 power supply voltage control circuit 10 charging circuit 20 drive circuit 30 drive signal 50 complementary circuit

─────────────────────────────────────────────────── ─── Continuation of the front page (51) Int.Cl. ⁶ Identification number Internal reference number FI Technical location H03K 17/567 17/695 9184-5K H03K 17/687 B (72) Inventor Yasushi Kawaji Hiratsuka, Kanagawa Prefecture 2597 Ichinomiya, Komatsu Ltd. Electronics Division

Claims (4)

[Claims] 1. A load drive device having a current control element provided between a load and a power supply, and a drive circuit for driving the current control element, wherein the current control element is a voltage of the power supply applied to the load. Is a switch means for turning on / off, wherein the drive circuit has a capacitor charged by a potential difference generated between the load and the power source when the switch means is off, and a charge charge of the capacitor when the switch means is on. A load drive device, comprising: a control power supply circuit that supplies a control voltage exceeding the voltage of the power supply to the drive circuit. 2. A load drive device comprising a current control element provided between a load and a power supply and a drive circuit for driving the current control element, wherein the load is generated between the load and the power supply when the switch means is off. A first capacitor that is charged with a potential difference that controls the voltage, a control power supply circuit that provides the drive circuit with a control voltage that exceeds the voltage of the power supply when the switch device is turned on by the charge stored in the first capacitor, and the switch device. Is charged at a charging voltage higher than the charging voltage of the first capacitor due to a potential difference generated between the load and the power supply when the power is turned off, and the control voltage by the first capacitor decreases A load driving device comprising one or a plurality of second capacitors for charging one capacitor. 3. A load drive device having a current control element provided between a load and a power supply and a drive circuit for driving the current control element, wherein the current control element varies a current applied to the load. A current varying means, wherein the drive circuit obtains a potential difference before and after the current varying means, and a power source that controls the voltage of the power source so as to reduce the potential difference according to the detection result of the potential difference measuring means. A load drive device comprising: a voltage control unit. 4. The load driving device according to claim 3, wherein the operation of the power supply voltage control means is stopped and the voltage of the power supply is fixed while the current varying means is turning off the current.