JPH09224399A - Induction motor control device - Google Patents

Abstract

(57) 【Abstract】 PROBLEM TO BE SOLVED: To correct using a rotation matrix corresponding to a primary frequency, or to correspond to a phase delay equivalent to a primary current in the past 1
The purpose is to use the next magnetic flux estimated value at the time of torque estimation to correct the torque estimation error and perform the control calculation accurately. SOLUTION: For the primary currents i1d and i1q which are dq converted by the 3-phase / 2-phase conversion means 52b, the frequency converter 11 is provided.
To obtain the primary frequency f, and the phase compensator 10 to calculate the phase delay angle to obtain the primary current i1d, i1q and the phase-compensated primary current detection value i1d ', i1q' based on the rotation matrix. Or obtain the primary magnetic flux φ from the integrators 54a and 54b in the phase compensation unit 30.
Input 1d, φ1q, phase-compensated primary magnetic flux φ1d 'and φ
1q 'is input to the multiplier for obtaining the vector product, and the primary current values i1d and i1q that are not phase-compensated are input to the multiplier as they are to compensate the phase delay of the primary current value.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a control device for an induction motor, and more particularly, it detects a primary current of the motor, compensates the phase of the primary current, and feeds back the compensated primary current detection value. The present invention relates to a control device for an induction motor that controls a motor.

[0002]

2. Description of the Related Art Normally, when the control of an induction motor is configured by a digital control device, various control quantities such as magnetic flux and torque are estimated from the taken-in value of primary current and the like, and the estimated value and the control target value are The switching command signal to the inverter is determined and output from the comparison. By the way, in recent years, since the performance of digital arithmetic elements has improved and high-speed arithmetic processing has become possible,
The digitization of various proposed control methods is progressing,
An all-digital induction motor controller has also been reported.

FIG. 6 is a system block diagram incorporating an analog filter, and FIG. 7 is a system block diagram incorporating a digital filter. In the figure, 41 is a current detector, 42 is an analog filter, 43 is an A / D converter,
Reference numeral 44 is a control calculation means, 45 is an inverter, 46 is an induction motor (IM), and 47 is a digital filter. In order to detect the primary current flowing through the induction motor 46 by the current detector 41 and digitize the control method of the control calculation in the control calculation means 44, the primary current detection value which is an analog quantity is converted into a digital quantity. A / D converter 43 is required.

Normally, the filter is the current detector 41 from the A /
It is provided in order to remove the noise component superimposed on the current detection waveform in the wire harness portion up to the D converter 43, and as shown in FIG. Low-pass filter 42 by wear
Is generally used, or a low-pass filter 47 is incorporated in the digital value after A / D conversion as shown in FIG.

In the above system, the primary current detection value fetched by the A / D converter 43 is used in the control calculation means 44 in the subsequent stage for the control calculation based on each control algorithm. For example, in vector control widely applied to control of induction motors, torque control is realized by comparing a command value of a primary current with a detected value to form a PWM signal. In addition, as a conventionally known document, for example, "instantaneous magnetic flux vector control method" proposed by Isao Takahashi of Nagaoka University of Technology
(Research Report of Nagaoka University of Technology, No. 8 1986, No. 4
See pages 3 to 50). In this control method, the output torque is estimated from the vector product of the estimated primary magnetic flux value and the detected primary current value, and the command value of the output torque and the estimated value are compared to form a voltage vector to realize torque control. doing.

In this instantaneous space magnetic flux vector control method, the primary magnetic flux of the induction motor is estimated from the primary current detected value of the induction motor and the like, and the estimated primary magnetic flux value and the detected primary current are obtained as follows. This is a magnetic flux calculation type control method that estimates the output torque from, and is characterized in that the torque control can be performed at high speed.
However, as will be described in detail below, the present control method does not mention compensation of an error due to a phase delay of the primary current after passing through the filter means.

In this control method, in a three-phase induction motor,
When the primary magnetic flux vector is φ1 and the output torque is T, φ1 = ∫ (V1−R1 × i1) dt (1) T = φ1 × i1 (2) Here, V1: primary voltage vector i1: primary current vector R1: primary winding resistance. The control principle of the instantaneous space magnetic flux vector control method will be schematically described below with reference to the drawings.

FIG. 8 is a system configuration diagram of a conventional induction motor control device to which the instantaneous space magnetic flux vector control method is applied. As shown, the system includes a three-phase inverter 50,
Primary voltage detection means 50a, primary current detection means 50b, 3
3-phase induction motors 51, 1 controlled by a phase inverter
Three-phase / two-phase conversion means 52a for the secondary voltage detection value, three-phase / two-phase conversion means 52b for the primary current detection value, primary current detection value and 1
It is composed of an instantaneous space magnetic flux vector forming means 60 which forms a voltage vector for controlling the three-phase inverter by performing a control calculation process on the value after the three-phase / two-phase conversion of the next voltage detection value.

The three-phase / two-phase conversion means 52a for the primary voltage has three
Each phase voltage V1 of the phase induction motor is input and converted into voltages V1d and V1q in order to be expressed by the dq axis variables. on the other hand,
The primary current three-phase / two-phase conversion means 52b inputs each phase current i1 of the three-phase induction motor and converts it into currents i1d and i1q in order to represent it by variables on the dq axis. The operation of the instantaneous space magnetic flux vector forming means 60 is roughly as follows. That is,
After inputting the primary voltage detection value and the primary current detection value, d-
q-converted voltages V1d and V1q and currents i1d and i1
Based on q, the d-axis component integrator 54a and the q-axis component integrator 54b with respect to the voltage, absolute value calculation means 55 for obtaining the magnitude of the primary magnetic flux, and a magnetic flux (φ) using a multiplier or the like.
And a predetermined calculation is performed on the torque (T).

This calculation result and the input primary magnetic flux (φ
The difference between the command value of 1) and the command value of the torque (T) is binarized by a binary hysteresis comparator 57 and a ternary hysteresis comparator 56 for judging excess and deficiency of two levels of excess and deficiency. And tri-valued, these values (τ),
The optimum switching table is referred to based on (φ) and the flux linkage vector angle (θ) that has been hexadecimalized by the magnetic flux angle comparator 58, and the output voltage vectors UU, VU, WU are output to the inverter 50.

Further, the voltage drop calculators 53a and 53b have three units.
1 using the primary current after the conversion of the phase / two-phase conversion means 52b
Calculate the voltage drop in the secondary winding resistance. Integrator 54a,
More specifically, 54b performs time integration of various amounts obtained by subtracting the voltage drop amount obtained by the voltage drop calculation unit from the converted primary voltage of the three-phase / two-phase conversion unit 52a, and then the expression (1) is obtained. The primary magnetic fluxes φ1d and φ1q are obtained.

FIG. 9 is a diagram showing a switching state of the three-phase inverter shown in FIG. The illustrated three-phase inverter 50 is
Positive bus 71a, negative bus 71b, 72a and 72b, 7
3a and 73b, and U and 74a and 74b, which form a pair,
Three sets of changeover switches for V and W phases (for example, semiconductor switching elements) and the positive bus 71 are connected to these changeover switches.
a, a DC power source 70 connected via a negative bus 71b
And

With such a structure, the changeover switches are changed over to apply the voltage on the positive bus side or the negative bus side to the three-phase induction motor. In this case, the pair of change-over switches do not conduct vertically and simultaneously. Here, the case where the upper changeover switches 72a, 73a, 74a are conducted is set to "1", and the case where the lower changeover switches 72b, 73b, 74b are conducted is set to "0". As defined in this way, U,
When the conduction state is expressed in order of the V and W phases, in the state example of the changeover switch shown in FIG. 9, 72a is "0", 7 due to the lower conduction.
3a is "1" in the upper conduction and 74a is "0" in the lower conduction, and as a result, it is expressed as (010).

FIG. 10 shows a three-phase inverter 50 in FIG.
7 is an explanatory diagram of a voltage vector applied to the phase induction motor 51. FIG. As shown in the figure, voltage vectors having the same magnitude but different directions are applied to the three-phase induction motor 51 depending on the conduction state of each changeover switch. That is, there are 8 types (10
0,110,010,011,001,101,11
A voltage vector of 1,000) can be applied.
Six of these voltage vectors (100, 110,
010,011,001,101) are equal in size,
Voltage vectors with different directions. For the other two types, the voltage on the negative bus 71b side of the three-phase induction motor is (000), and the positive bus 71a is (111).
The voltage on the side is applied, and the voltage vector has no magnitude and direction.

FIG. 11 shows an instantaneous space magnetic flux vector forming means 6
It is explanatory drawing of the state of the primary magnetic flux vector of 0. here,
When the voltage drop due to the primary winding resistance is omitted, the primary magnetic flux can be represented by the primary voltage, that is, the vector sum of the voltage vectors, as shown in the figure. Absolute value calculation means 5 described above
5 is (φ1d ² + φ1q ² ) ^1/2 (3) based on the primary magnetic fluxes φ1d and φ1q obtained by the integrators 54a and 54b, that is, the magnitude of the estimated primary magnetic flux is obtained. .

The magnetic flux determination means 57 corresponding to the above-mentioned binary hysteresis comparator takes the difference between the magnitude of the estimated primary magnetic flux value calculated by the equation (3) in the absolute value calculation means 55 and the primary magnetic flux command value. , Determine the excess or deficiency. Then, using the determination result of excess or deficiency, when the primary magnetic flux is to be increased, the voltage vector in the outer circumferential direction of the magnetic flux circle (see the outward arrow) is to be decreased, and in the inner circumferential direction (toward the inward direction). By outputting the voltage vector of (see arrow) every moment,
The primary magnetic flux can be controlled by approximating it to a magnetic flux circle having a constant hysteresis width.

FIG. 12 is an explanatory diagram of the magnetic flux quadrant of the primary magnetic flux vector. As shown in the figure, the dq coordinate system of the primary magnetic flux is divided into six equal parts to determine the magnetic flux quadrant. The magnetic flux quadrant determining means 58 corresponding to the magnetic flux angle comparator of FIG.
Based on the primary magnetic fluxes φ1d and φ1q obtained in 4b, the phase θ of the primary magnetic flux vector is determined, and it is determined in which quadrant the primary magnetic flux vector exists in the above 6 divisions. Is output to the switching table 59.

The torque determination means 56 corresponding to the three-valued hysteresis comparator of FIG. 8 is calculated by the vector product of the primary magnetic fluxes φ1d and φ1q and the primary currents i1d and i1q as shown in the equation (2). Excess or deficiency is determined by taking the difference between the estimated value of the output torque of the induction motor and the command value of the input output torque. In the case of a two-level inverter, since the size of the voltage vector that can be output is one type, excess / deficiency determination is performed at three levels: excess, appropriate, and insufficient, and output to the switching table.

Since the torque of the induction motor is approximately proportional to the slip, if it is desired to increase the output torque, the voltage vector in the rotation direction of the primary magnetic flux (the voltage vector for rotating the induction motor in the positive rotation direction) is decreased. If desired, the output torque can be controlled by outputting a voltage vector that rotates in the reverse rotation direction. In the present control, from the three determination results of the torque determining means 56, the magnetic flux determining means 57, and the magnetic flux quadrant determining means 58 described above, finally 3 is obtained.
The voltage vector output to the phase inverter 50 is selected.

FIG. 13 shows the switching table 59 shown in FIG.
It is a figure which shows the table of the voltage vector selected by.
As shown in the figure, in the switching table, the voltage vector to be selected in each magnetic flux quadrant is preset for the determination results of the torque excess / deficiency determination and the magnetic flux excess / deficiency determination,
This voltage vector (UU, VU, WU) is selected by the respective judgment results of the torque judging means 56, the magnetic flux judging means 57, and the magnetic flux quadrant judging means 58, and the three-phase inverter 50 is selected.
Is output to

The instantaneous space magnetic flux vector forming means 60
The means for realizing
You may process by SP (Digital Signal Processor) etc. Alternatively, the primary voltage may be uniquely calculated from the DC voltage of the three-phase inverter and the output voltage vector without directly detecting the primary voltage. Further, as another known document of the control device for the induction motor, there is, for example, Japanese Patent Laid-Open No. 64-1497. This document discloses a method of correcting the phase delay after passing through a low-pass filter. In this method, the frequency characteristic of the transmission delay angle of the filter is stored in advance, and the stored data and the filter output are waveform-shaped. The PWM signal is controlled based on the above.

However, in this method, the phase difference between the primary current and the primary voltage is estimated by detecting and determining the polarity of the primary voltage at the zero cross point of the primary current, and the optimum phase difference ( It is a method to control the induction motor with a high power factor)
1 corresponding to the phase delay of the primary current due to the filter
The polarity detection timing of the secondary voltage is shifted to eliminate the primary voltage discrimination error due to the phase delay of the primary current.

[0023]

As described above, in the conventional motor control device in which the primary current flowing in the induction motor is applied to the control calculation via the A / D converter, the detection error of the primary current is detected. Causes a decrease in estimation accuracy of magnetic flux, torque, and the like, and as a result, problems such as occurrence of an unstable phenomenon and deterioration of torque control accuracy occur.

By the way, the cause of the detection error in the primary current is due to the error generated in the primary current detector and in the wire harness section from the primary current detector to the A / D converter. It may be caused by noise superimposed on the next current detection waveform. Among these factors, with respect to the noise that is superimposed on the wire harness portion, as described in the related art of FIGS. 6 and 7, a countermeasure using a low-pass filter is generally taken. On the other hand, since the low-pass filter is for removing high frequency components, the influence of the phase delay is small in the low frequency region and large in the high frequency region. Since the low-pass filter has a phase characteristic as described above, a phase difference (phase delay) occurs between the primary current actually flowing in the induction motor and the detected primary current value. Since the control calculation is performed using the primary current detection value in which the phase difference has occurred,
The estimation accuracy of the control quantities such as the secondary magnetic flux and the torque decreases, and as a result, the above-mentioned problem occurs.

FIG. 14 is an explanatory view (part 1) of the relationship between the primary magnetic flux vector and the primary current vector, and FIG. 15 is an explanatory view (part 2) of the relationship between the primary magnetic flux vector and the primary current vector. . Here, a decrease in torque estimation accuracy in the instantaneous space magnetic flux vector control method will be specifically described below with reference to the drawings. In this control method, the output torque is estimated from the vector product of the primary magnetic flux estimated value and the primary current detected value, as shown in the equation (2).

In these figures, the shaded area (| T |
And | T | ') represent the magnitude of the vector product, that is, the magnitude of the estimated value of the output torque. Here, FIG.
When a phase difference (phase lag) occurs in the primary current (i1) 81a of FIG. 15, it is expressed as the primary current (i1 ′) 81b of FIG. 15, and the shaded area of FIG. 14 and the shaded area of FIG. As is clear from the comparison of the above, the estimated value of the output torque is calculated smaller than that when there is no phase delay.

As described above, in this control, the estimated value of the output torque is compared with the command value to form the voltage vector. Therefore, when a phase delay occurs in the primary current detection value, the output torque is estimated to be smaller than the actual output torque, and a voltage vector that increases the output torque is selected and output. Become. As a result, an excessive output torque is generated with respect to the command value of the output torque, and the torque control accuracy is reduced. Then, as described above, the excessive degree of the output torque becomes more remarkable as the frequency becomes higher.

The object of the present invention is to provide a low pass filter
When compensating for the phase difference (phase delay) of the secondary current detection value, correction is performed using the rotation matrix corresponding to the primary frequency, or the past primary magnetic flux estimation value corresponding to the phase delay equivalent to the primary current is calculated. It is an object of the present invention to provide a control device capable of accurately performing control calculation and improving controllability by correcting a torque estimation error due to a phase delay used during torque estimation.

[0029]

According to the invention described in claim 1, a rotation matrix (rotation vector) corresponding to a primary frequency of a primary current including a phase difference (phase delay) by a filter.
Is calculated and multiplied by the primary current that is not phase-compensated, or the past primary magnetic flux estimation value corresponding to the phase delay of the primary current is used at the time of torque estimation to obtain the current primary current value. The error in the torque estimation value due to the phase delay of the primary current is eliminated.

FIGS. 5A, 5B and 5C are waveform charts for explaining the relationship between the primary current detection waveforms in each means for explaining the effect of the present invention. As shown in (A) and (B), the primary current value 91 flowing in the induction motor and the primary current detection waveform 92 detected by the primary current detection means are represented by a fundamental wave for convenience. The phase delay of the primary current detection waveform 92 as shown in (B) is for removing the noise component superimposed on the primary current detection waveform in the wire harness section from the primary current detection means to the A / D conversion means. Due to the filter means (low-pass filter).

As described above, the low-pass filter cuts high-frequency components because it aims to remove noise components. However, depending on the frequency band used, there is a phase delay of the fundamental wave component. Primary current detection waveform 92 with this phase delay
On the other hand, as shown in (C), in the present invention, the phase characteristic of the filter is grasped in advance by the primary current phase compensating means, and the phase angle corresponding to the calculated delay time is compensated.
By the operation of each of the above means, the phase of the primary current detection waveform 93 after the phase compensation and the phase of the primary current flowing through the motor are made to coincide with each other, and as a result, accurate control calculation can be performed.

[0032]

BEST MODE FOR CARRYING OUT THE INVENTION A first embodiment of the present invention will be described below with reference to the drawings. FIG. 1 is a basic configuration block diagram of the first embodiment of the present invention. First, the frequency calculation means 5
Calculates the primary frequency for the current value i1 obtained by A / D converting the primary current detection value detected by the primary current detection means 1 through the filter 2. In this embodiment, the time of one cycle of the primary current detection value is calculated by counting with a timer. In addition, 1
The primary current detection value is not used for calculating the secondary frequency, and other quantities such as the primary voltage value or the primary magnetic flux value that can calculate the primary frequency may be used. Further, the frequency calculating means 5 can be realized by other methods such as deriving from a modified arithmetic expression of the voltage-current equation instead of using a timer. Since the control calculation means 6 and the inverter 7 are the same as the conventional ones described above,
Description is omitted.

The phase compensating means 4 firstly comprises the frequency calculating means 5
The delay phase angle of the filter means at the present time for the primary current is calculated from the primary frequency calculated in step 1 and the phase characteristic of the filter means 2 which is known in advance. Then, the calculated delay phase angle is substituted into the rotation matrix shown in the equation (4) below to calculate the current phase compensation matrix. In this embodiment, the rotation matrix for the primary current detection value after the 3-phase / 2-phase conversion is applied.

[0034]

[Equation 1]

Next, the rotation matrix calculated by the equation (4) is multiplied by the determinant of the detected primary current as shown in the following equation (5) to obtain the delay phase of the detected primary current. Compensate by advancing the angle θ. Here, i1 ′ is a phase-compensated primary electric value, and i1d ′ and i1q ′ in FIG.
Corresponding to

[0036]

[Equation 2]

FIG. 2 is a system configuration diagram of FIG. 1, and is a system configuration diagram when the above-described first embodiment is applied to the instantaneous space magnetic flux vector control method. In the figure, 10 is a phase compensator, which corresponds to the phase compensator in FIG. 1, and 11 is a frequency calculator, which corresponds to the frequency calculator in FIG. The other components are the same as those in FIG.

As shown, the three-phase / two-phase conversion means 52b
With respect to the primary currents i1d and i1q that are dq converted by the above, the primary frequency f is obtained by the frequency converter 11, and the phase delay angle is calculated by the phase compensating unit 10 to obtain the primary current i1.
Phase-compensated primary current detection values i1d ′ and i1q ′ are obtained as shown in equation (5) based on d, i1q and the rotation matrix of equation (4).

The phase-compensated primary current detection values i1d 'and i1q' from the phase compensator 10 are the voltage drop calculator 53.
a, 53b, a multiplier, etc., and thereafter, FIG.
Operates in the same manner as described above. As described above, instead of the primary current detection values i1d and i1q with phase delay, the phase-compensated primary current detection values i1d ′ and i1q ′ are used.
Is used for the subsequent control calculation, it is possible to perform a control calculation that does not cause an error due to the phase delay of the primary current detection value even when the primary current detection value has a phase delay.

FIG. 3 is a block diagram of the basic configuration of the second embodiment of the present invention. In the above-described first embodiment 1, an example in which the phase compensation is performed by the primary current detection value itself is shown, but in the second embodiment, for various amounts other than the primary current (for example, output torque). This is an example in which it is possible to obtain the same effect as in the case where the primary current detection value is phase-compensated by compensating the phase.

That is, as described above, in the instantaneous space magnetic flux vector control method, the output torque is the vector product of the estimated primary magnetic flux value (φ1) and the detected primary current value (i1) as shown in the equation (2). Therefore, if the primary current detection value is phase-delayed as described above, the included angle between the primary magnetic flux vector and the primary current vector will be underestimated. As a result, after the vector multiplication, An error occurs in the magnitude of the estimated value of the output torque (see FIGS. 14 and 15).

In the second embodiment, the primary current is not compensated for the phase delay, but the phase of the primary magnetic flux estimated value is retarded to relatively compensate the angle between the primary current detected value and the detected primary current. It is possible to obtain the same effect as when the lead angle is compensated for the phase of the primary current detection value. That is, in the first embodiment, the phase delay of the primary current due to the filter is corrected by using the rotation matrix corresponding to the primary frequency, but in the second embodiment,
Of the two values of the primary current value and the primary magnetic flux estimated value that form the torque estimation equation of the equation (2), the past primary magnetic flux estimated value corresponding to the phase delay of the primary current is estimated by the torque. By using it at any time, the torque estimation error due to the phase delay of the primary current is eliminated.

In FIG. 3, the filter means and the A / D converting means after the primary current detecting means 20 and the primary voltage detecting means 21 are omitted for the sake of simplifying the description. First, the phase compensating means 23 for the primary magnetic flux estimated value stores the past torque estimated value for the time corresponding to the delay time generated by the filter means in the memory in advance in the past calculation cycle. Read from inside. In the digital control, since each control value is a discrete value, the nearest primary magnetic flux estimation value is read at the past time corresponding to the delay time and input to the phase compensating means 23. In order to secure the control accuracy, it may be calculated using various interpolations from the primary magnetic flux estimation values before and after the past time corresponding to the delay time. And
The estimated primary magnetic flux value in the current calculation cycle is stored for use by the primary magnetic flux phase compensating means in the subsequent calculation cycle.

In the control calculation means 24, the primary magnetic flux estimation value (φ1 in FIG. 4) which has been retarded by the primary magnetic flux phase compensation means 23 is compensated.
d ′, φ1q ′) and the phase-compensated primary current detection value (i1d, i1q in FIG. 4) are used as the vector product to estimate the magnitude of the output torque, and the result is estimated as the output torque command value (Fig. (4 T command value of 4), and the magnetic flux judgment and the magnetic flux quadrant judgment result are combined to select a voltage vector from the switching table and output to the three-phase inverter.

FIG. 4 is a system configuration diagram of FIG. Reference numeral 30 in the figure denotes a phase compensating unit, which corresponds to the phase compensating means 23 in FIG. The primary magnetic fluxes φ1d and φ1q from the integrators 54a and 54b are input to the phase compensation unit 30, and the phase-compensated primary magnetic fluxes φ1d ′ and φ1q ′ are input to a multiplier for obtaining a vector product. On the other hand, the primary current values i1d and i1q that are not phase-compensated are input to the multiplier as they are and the vector product with the phase-compensated ones is obtained. The result and the inputted torque command value (T command value) are compared and judged by the torque judgment means 56, and the voltage vector is selected from the switching table by combining the magnetic flux judgment and the magnetic flux quadrant judgment result to the three-phase inverter. Output. Further integrator 54
The primary magnetic fluxes φ1d and φ1q from a and 54b are directly input to the absolute value calculating means 55 without phase compensation. The other components are the same as those in the conventional FIG.

[Brief description of drawings]

FIG. 1 is a basic configuration block diagram of a first embodiment of the present invention.

FIG. 2 is a system configuration diagram of FIG.

FIG. 3 is a basic configuration block of a second embodiment of the present invention.

FIG. 4 is a system configuration diagram of FIG.

FIG. 5 is an explanatory diagram of a relationship between a primary current waveform (A) flowing through a motor, a primary current waveform (B) after passing through a filter, and a primary current waveform (C) after phase compensation.

FIG. 6 is a system block diagram incorporating an analog filter.

FIG. 7 is a system block diagram incorporating a digital filter.

FIG. 8 is a system configuration diagram of an induction motor control device to which a conventional instantaneous space magnetic flux vector control method is applied.

9 is an explanatory diagram of a switching state of the three-phase inverter of FIG.

FIG. 8 is an explanatory diagram of voltage vectors applied from a three-phase inverter to a three-phase induction motor.

FIG. 11 is an explanatory diagram of a state of a primary magnetic flux vector.

FIG. 12 is an explanatory diagram of a magnetic flux quadrant of a primary magnetic flux vector.

FIG. 13 is an explanatory diagram of a switching table that selects a voltage vector.

FIG. 14 is an explanatory diagram (part 1) of the relationship between the primary magnetic flux vector and the primary current vector.

FIG. 15 is an explanatory diagram (Part 2) of the relationship between the primary magnetic flux vector and the primary current vector.

[Explanation of symbols]

 1, 20 ... Primary current detection means 21 ... Primary voltage detection means 2 ... Filter means 3 ... A / D conversion means 4, 23 ... Phase compensation means 5 ... Frequency calculation means 6, 24 ... Control calculation means 7 ... Inverter 8 … Induction motor

 ─────────────────────────────────────────────────── ─── Continuation of the front page (72) Inventor Masaaki Tsuzuki, 14 Iwatani, Shimohakaku-cho, Nishio-shi, Aichi Japan Auto Parts Research Institute, Inc. (72) Inventor Yasushi Kusaka 1-cho, Toyota-cho, Aichi Prefecture Toyota Auto Car Co., Ltd. (72) Inventor Katsumi Kondo 1 Toyota Town, Toyota City, Aichi Prefecture Toyota Motor Co., Ltd.

Claims (2)

[Claims] 1. An induction motor control device for controlling an output torque of an induction motor by an inverter, comprising: a primary current detection means for detecting a primary current flowing through the induction motor; Filter means for removing a noise component superimposed on the detected value of the next current, A / D conversion means for converting the detected value of the analog primary current from the filter means into a digital value, and detection of the primary current of the digital quantity A frequency calculating means for calculating a primary frequency from various electrical quantities on the primary side including a value; and an output of the A / D converting means and an output of the frequency calculating means according to the calculated value of the primary frequency. The phase-compensated primary current is obtained by multiplying the rotation vector whose rotation angle changes by the primary current detection value that has passed through the filter means, and the primary current detection value of the filter means is obtained. Control for an induction motor, characterized by comprising: a phase compensating means for compensating a phase delay, a. 2. An induction motor control device for controlling an output torque of an induction motor by an inverter, comprising primary current detection means for detecting a primary current flowing through the induction motor, and a primary voltage flowing through the induction motor. Primary voltage detection means for detecting a noise, a filter means for removing a noise component superimposed on the primary current detection value by the primary current detection means, and a digital value for converting the analog primary current detection value from the filter means. A / D conversion means for converting the primary magnetic flux estimated value obtained from the detection values of the primary current detection means and the primary voltage detection means is temporarily stored, and 1 is stored from the stored estimated primary magnetic flux values. By reading the past primary magnetic flux estimation value corresponding to the phase delay time of the secondary current detection value and estimating the output torque by using the read past primary magnetic flux estimation value and the current primary current detection value. The control device of an induction motor, characterized by comprising a phase compensation means for compensating a phase delay of the primary current detection value in said filter means.