JPH09262000A - Induction motor control device - Google Patents

Abstract

(57) [Abstract] [PROBLEMS] To enable maximum efficiency operation even when driving an induction motor in which iron loss is not negligible or even when the motor constant fluctuates due to temperature changes during operation. SOLUTION: A calculation result of a predetermined function having a constant value of an induction motor including a constant related to iron loss as a coefficient value is stored, and a variable related to the predetermined function is input to input a torque current command and an exciting current. Outputs the command amplitude ratio. From the torque command of the induction motor and the above ratio, the current component command calculating means is used as the d-axis component command of the primary current on the orthogonal rotation coordinate axis (referred to as dq axis) that rotates the exciting current command at the primary frequency. Then, the torque current command of the induction motor is output as the q-axis component command of the primary current, the slip frequency of the induction motor is calculated from these outputs, and the primary frequency is obtained by adding it to the primary current. A primary current command in which the product value of the outputs of the current component command calculation means is proportional to the torque command and the amplitude ratio is equal to the predetermined function value is output.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to an induction motor vector control device, and particularly to an induction motor in which iron loss is not negligible, and good vector control performance even when driving an induction motor whose temperature changes greatly during operation. And an induction motor controller that enables maximum efficiency operation, and
The present invention relates to a control device for an induction motor that eliminates saturation of output voltage and prevents deterioration of torque control characteristics.

[0002]

2. Description of the Related Art As is well known, an induction motor vector control system separates a primary current of an induction motor into an exciting current component and a torque current component, and can freely control a generated torque and a secondary magnetic flux. It is a method. Here, when the exciting current component is Io and the torque current component is It, the torque Tm generated by the induction motor is expressed by the following equation.

[0003]

[Equation 1] Here, Pm is the number of pole pairs, and M is the primary and secondary mutual inductance of the induction motor.

The primary frequency ω of the induction motor is given by the following equation.

[0005]

[Equation 2] Here, ωr is the rotation frequency of the induction motor, and ωs is the slip frequency, which is given by the following equation.

[0006]

(Equation 3)

As can be seen from the above equation (1), there are innumerable combinations of Io and It for obtaining the generated torque of a certain value. By adjusting the combination of Io and It, it becomes possible to operate the induction motor with high efficiency. As a first conventional technique related to this, as a reference [Showa 57th National Conference of the Institute of Electrical Engineers of Japan, No. 574] is known. In this method, the induction motor can be operated with the maximum efficiency by controlling Io and It so as to satisfy the following expressions in accordance with the torque command.

[0008]

(Equation 4)

At this time, it can be seen from equation (3) that ωs has a constant value as in the following equation.

[0010]

(Equation 5)

FIG. 23 shows a conventional example of a control device for an induction motor that performs maximum efficiency operation of the induction motor based on the first conventional technique. This control device is disclosed in Japanese Patent Laid-Open No. 5-38181. In the figure, 1 is an induction motor, and the primary currents Ius, Ivs, and Iws for each phase supplied from the PWM inverter 10 to the induction motor 1 are detected by the current detector 4. The actual rotation speed of the induction motor 1, that is, the rotation frequency ωr is detected by the speed detector 7 and input to the subtractor 121.

The subtractor 121 outputs a rotation frequency ωr and a speed command ωr * output from a speed command generator (not shown).
And the deviation is obtained and input to the amplifier 122. Amplifier 12
2 outputs the deviation result as the torque current component command It *. The multiplier 123 multiplies the torque current component command It * by the exciting current component command Io * obtained by the calculation described later, and outputs a signal T proportional to the torque to the function generator 124.

When the function generator 124 receives the signal T, it calculates the exciting current component command Io * and outputs it through the compensation circuit 125. The exciting current component command Io * is a multiplier 123 and a divider 126 to which the torque current component command It * is input.
And the vector calculator 129.

The divider 126 outputs the torque current component command It
* Is divided by the exciting current component command Io *, and the division result is input to the coefficient unit 127 for calculating the slip frequency ωs. Then, the adder 128 determines the slip frequency ωs and the rotation frequency ωr.
Are added to calculate the primary frequency ω of the induction motor 1 and input to the vector calculator 129. Further, the vector calculator 129 inputs the primary current commands Ius *, Ivs * for each phase to be supplied to the induction motor 1 from the input torque current component command It *, exciting current component command Io * and primary frequency ω. Calculate and output Iws *.

Deviations between these primary current commands and the primary currents Ius, Ivs and Iws detected by the current detector 4 are obtained by subtractors 130, 131 and 132, respectively, and these deviations are amplified by amplifiers 133 and 134. And 135, and outputs to the PWM inverter 10 as primary voltage commands Vus *, Vvs * and Vws * for each phase of the induction motor 1.

The operation of the conventional device will be described. First,
The subtracter 121 obtains the deviation between the speed command ωr * output from the speed command generator (not shown) and the rotation frequency ωr output from the speed detector 7, and the amplifier 12
2 is input. Then, the amplifier 122 outputs the torque current component command It *.

Next, the multiplier 123 multiplies the torque current component command It * by the exciting current component command Io * obtained by the calculation described later, and outputs a signal T proportional to the torque. Here, from the equation (1), the signal T
It can be seen that is expressed by the following equation.

[0018]

(Equation 6)

When the signal T is input to the function generator 124, the exciting current component command Io * is obtained according to the following equation.

[0020]

(Equation 7)

However, K is a constant and is given by the following equation from the equation (1).

[0022]

(Equation 8)

The compensating circuit 125 is provided for the purpose of correcting the secondary magnetic flux of the induction motor 1 because it has a response characteristic of a primary delay with respect to the exciting current component. Then, the exciting current component command Io * is output.

Next, when the torque current component command It * and the exciting current component command Io * are input to the divider 126, It *
The value of / Io * is determined. Therefore, when the output of the divider 126 is input to the coefficient unit 127 whose coefficient value is equal to Rr / M, the slip frequency ωs is output by the calculation of the equation (3). Then, the slip frequency ωs and the velocity detector 7
When the rotation frequency ωr output from the adder 128 is added by the adder 128, the calculation of Expression (2) is performed, and the primary frequency ωr
Is required.

Further, the above torque current component command It *,
When the exciting current component command Io * and the primary frequency ω are input to the vector calculator 129, the primary current commands Ius *, Ivs * and Iws * for each phase to be supplied to the induction motor 1 are output. Subsequently, these primary current commands and the primary currents Ius, Ivs and I detected by the current detector 4 are detected.
The deviations from ws are subtractors 130, 131 and 1 respectively.
32. These deviations are calculated by the amplifier 13
When input to 3, 134 and 135, the primary voltage commands Vus *, Vvs * and Vws * for each phase of the induction motor 1 are output respectively.

By the PWM inverter 10, the primary voltages Vus, Vvs and Vws of the induction motor 1 are respectively
It is controlled so as to follow these primary voltage commands. As a result, the primary currents Ius, Ivs and I of the induction motor 1
ws follows each command. As a result, the exciting current component and the torque current component of the induction motor 1 also follow the respective commands.

Further, in the conventional vector control method for the induction motor, when the output voltage is saturated due to the limitation of the DC voltage on the power source side, the output waveform of the inverter is disturbed, causing vibration and deterioration of torque control performance. As a second related art related to this, the vector control device shown in Japanese Utility Model Laid-Open No. 6-21394 is known. In the vector control device according to this conventional technique, the limit value of the primary voltage that can be output obtained from the DC voltage on the power supply side is compared with the primary voltage command, and the correction component of the magnetic flux command is calculated by proportional integral (PI) calculation. When the voltage saturation occurs, the voltage saturation is canceled by correcting the magnetic flux command with the magnetic flux command correction component.

FIG. 24 shows a vector control device according to this conventional technique. In the figure, the same reference numerals as those in FIG. 23 denote the same or corresponding parts. In FIG. 24, 141 is a three-phase power supply,
Reference numeral 142 is a converter for converting the three-phase alternating current from the three-phase power supply 141 into direct current, 143 is a direct current voltage detecting portion for detecting the direct current voltage converted by the converter 142 and outputting it as an output signal Vdc, 149 is the current detector 4 Inputting the three-phase currents Ius, Ivs, and Iws detected by the d-axis component Ids of the primary current on the orthogonal rotation coordinate axes (referred to as dq axes)
And a coordinate transformation unit (A) that outputs the q-axis component Iqs, 14
Reference numeral 8 denotes a primary current q-axis component command I used for the torque current command.
As qs *, ASR amplifier (speed command ωr * and speed ω
The switching unit 144, which selects either the output of the (r deviation amplifier) 122 or the externally set torque command, selects the DC voltage Vdc and the primary voltage command Vs * which is the output of the coordinate conversion unit (B) described later. A field weakening calculation unit that inputs and outputs an excitation command correction component ΔIds at the time of voltage saturation, 145 is a limiter processing unit that inputs a field command setting value and performs limiter processing, and 146 is an excitation command that has been subjected to limiter processing. 1 is used for the excitation current command by subtracting the excitation command correction component ΔIds from
A subtractor that forms the d-axis component command Ids * of the next current, 150 is a subtracter that inputs the q-axis component command Iqs * of the primary current and the q-axis component Iqs, and obtains the deviation thereof. 151 is the d of the primary current.
A subtractor for inputting the axis component command Ids * and the d-axis component Ids to obtain the deviation thereof, 152 and 153 amplify the deviation obtained by the subtractors 150 and 151, respectively, and amplify the q-axis correction components ΔVqs and d of the primary voltage. An ACR amplifier (amplifier) 147 that outputs the axis correction component ΔVds is a d-axis component command Ids * and a q-axis component command Iqs * of the primary current and a speed detector 7.
The input of the rotation frequency ωr, which is the output of the induction machine, performs a vector operation simulating the induction machine 1, and outputs the d-axis component Vds and the q-axis component Vqs of the primary voltage.
An adder that adds the q-axis component Vqs of the next voltage and the q-axis correction component ΔVqs to form a q-axis component command Vqs * of the primary voltage,
155 is the d-axis component Vds of the primary voltage and the d-axis correction component ΔV
An adder for adding ds to form a d-axis component command Vds * of the primary voltage, 156 is a q-axis component command Vqs * of the primary voltage and d
A coordinate conversion unit (B) that inputs the axis component command Vds *, converts it into polar coordinates, and outputs the primary voltage command Vs * and the phase ψ, 1
57 inputs the primary voltage command Vs * and the phase ψ and inputs the primary voltage commands Vus *, Vvs * and Vw for each phase of the induction motor 1.
A PWM calculation unit for obtaining s * and performing PWM calculation, 158 is P
It is a base circuit that receives the output of the WM calculator 157 and generates switching signals for each phase in the PWM inverter.

FIG. 25 is a block diagram showing the detailed structure of the field weakening calculator 144 described above. Field weakening calculator 1
Reference numeral 44 denotes a limit output calculation unit 161 that inputs the DC voltage Vdc, calculates and outputs the limit Vsmax of the primary voltage that can be output by the PWM inverter 10, and the primary voltage command Vs *.
Is input to obtain the first-order lag component Vs * F, and Vs * F <
In the case of Vsmax, the saturation voltage command Vs * '= Vsm
ax, and when Vs * F ≧ Vsmax, a saturation voltage command generator 162 that outputs Vs * ′ as a saturation voltage command Vs * ′ = Vs * F, and a limit Vsmax of the primary voltage that can be output. It is composed of a subtracter 163 for obtaining a deviation of the saturation voltage command Vs * ', and a PI calculator 164 for amplifying the deviation and outputting an excitation command correction component ΔIds.

The operation of this conventional device will be described. First, the switching unit 148 selects whether the induction motor 1 is operated by the speed control system or the torque control system. When the ASR amplifier 122 side is switched, the speed control system is selected, and when the external set torque command side is switched, the torque control system is selected. When the speed control system is selected, the subtractor 121 obtains the deviation between the speed command ωr * and the rotation frequency ωr output from the speed detector 7, and the output of the ASR amplifier 122, which inputs this deviation, becomes the torque current command. Q-axis component command I of primary current to be used
It becomes qs *. When the torque control system is selected, the externally set torque command becomes the q-axis component command Iqs * of the primary current used for the torque current command.

Further, the excitation command set value that has passed through the limiter processing section 145 is processed by the subtracter 146 in the PI calculation section 1 which will be described later.
Excitation command correction component ΔIds at the time of voltage saturation output by 64
And the d-axis component command Ids * of the primary current used for the excitation current command is output.

When the primary currents Ius, Ivs, and Iws detected by the current detector 4 are input to the coordinate conversion unit (A) 149, the d-axis component Ids and the q-axis component Iq of the primary current are input.
s is output.

Next, the subtracter 150 obtains the deviation between the q-axis component command Iqs * of the primary current and the q-axis component Iqs. When this deviation is input to the ACR amplifier 152, the q-axis correction component ΔVqs of the primary voltage is output. Further, the subtracter 151 obtains the deviation between the d-axis component command Ids * of the primary current and the d-axis component Ids. When this deviation is input to the ACR amplifier 153, the d-axis correction component ΔVd of the primary voltage
s is output.

Further, the induction machine model 147 has a d of the primary current.
When the axis component command Ids *, the q axis component command Iqs *, and the rotation frequency ωr are input, the d axis component Vds and the q axis component Vqs of the primary voltage are calculated based on the voltage-current equation of the induction motor.
Is required. Subsequently, the q-axis component Vqs of the primary voltage and the q-axis correction component ΔVqs are added by the adder 154, and the q-axis component command Vqs * of the primary voltage is output. Further, the adder 155 adds the d-axis component Vds of the primary voltage and the d-axis correction component ΔVds, and outputs the d-axis component command Vds * of the primary voltage.

When the q-axis component command Vqs * and the d-axis component command Vds * of the primary voltage are input to the coordinate conversion section (B) 156, conversion into polar coordinates is performed according to the following equation, and the primary voltage command Vs *. And the phase ψ is output.

[0036]

[Equation 9]

PW the primary voltage command Vs * and the phase ψ.
When input to the M calculation unit 157, the primary voltage command V for each phase
Us *, Vvs *, and Vws * are obtained, and the PWM operation is performed. The base circuit 158 generates a switching signal for each phase of U, V, and W based on the PWM calculation result, and P
The WM inverter 10 is driven. Therefore, the induction motor 1
The primary voltages Vus, Vvs, and Vws of the induction motor 1 are controlled so as to follow these primary voltage commands, and as a result, the primary currents Ius, Ivs, and Iw of the induction motor 1 are increased.
s follows each command. As a result, the exciting current component and the torque current component of the induction motor 1 also follow the respective commands.

Here, the DC voltage Vdc output from the DC voltage detection unit 143 and the primary voltage command Vs * output from the coordinate conversion unit (B) 156 are input to the field weakening calculation unit 144. In the field weakening calculator 144, the limit output calculator 161 calculates the limit Vsmax of the primary voltage that can be output from the DC voltage Vdc based on the following equation, and outputs it.

[0039]

(Equation 10)

Further, the saturation voltage command generator 162 obtains the primary delay component Vs * F of the primary voltage command Vs *, and outputs the saturation voltage command Vs * 'according to the following equation.

[0041]

[Equation 11]

When Vs * F <Vsmax, it means that voltage saturation has not occurred, and Vs * F ≧ Vsmax.
In the case of, it indicates that voltage saturation has occurred. Subtractor 1
Output limit voltage Vsmax and saturation voltage command Vs at 63
Deviation of * 'is required. When voltage saturation has not occurred, the deviation is zero (0), and when voltage saturation has occurred, the deviation depends on the degree of saturation. This deviation is input to the PI calculation unit 164, and the above-described excitation command correction component ΔIds is obtained as an output. That is, the saturation voltage command generation unit 162 detects whether or not the voltage saturation occurs, and when the voltage saturation occurs, the excitation command correction component ΔIds is adjusted according to the degree, and the primary command which is the excitation command. The d-axis component command Ids * of the current is suppressed. Suppressing the d-axis component command Ids * of the primary current suppresses the primary voltage command, so that voltage saturation is eliminated.

[0043]

In the above-mentioned first conventional control device, the exciting current component and the torque current component are controlled according to the equations (4) and (5) in order to perform highly efficient vector control. . By the way, these equations are relational equations obtained by ignoring the iron loss of the induction motor. Therefore, if the exciting current component and the torque current component are controlled according to the equations (4) and (5), the copper loss of the induction motor, that is, the loss caused by the primary resistance and the secondary resistance can be minimized, but the iron loss can be reduced. Is not the minimum. Here, the iron loss of the induction motor is proportional to the 1.6th power of the primary frequency,
Reference [Symposium S. 10
-6] and the like. Therefore, for example, in a high-speed rotating induction motor used for driving an electric railway or an electric vehicle, iron loss cannot be ignored with respect to copper loss.

As a result, the conventional control device cannot operate at maximum efficiency when driving these induction motors. In particular, in the case of an electric vehicle, since the energy source is a battery, realization of maximum efficiency operation of the driving induction motor is an important issue for reducing the size and weight of the battery.

Further, the induction motor used for such an application is being downsized and the overload 2 is exceeded in excess of the rating.
It is often operated at a power output of more than 00%,
The fluctuation of the motor constant due to the change of the motor temperature during operation is large, and the deterioration of the torque control performance caused by this is a problem.

The above-mentioned second conventional control device detects this when voltage saturation occurs, corrects and reduces the magnetic flux command according to the degree of voltage saturation, thereby suppressing the primary voltage command and suppressing the voltage saturation state. It will be resolved. However, since the magnetic flux command is immediately operated upon occurrence of voltage saturation, when operating as a torque control system, there is a problem that the output torque becomes insufficient with respect to the command torque immediately when voltage saturation occurs.

Therefore, a first object of the present invention is to solve the problems of the first prior art described above, and to drive an induction motor in which iron loss is not negligible, and also due to temperature change during operation. Maximum efficiency operation is possible even when the motor constant fluctuates, and vector control with high-accuracy and high-speed response can be realized. Also, in such maximum high-efficiency control, it is also possible to realize with non-high-speed control calculation capability. Another object of the present invention is to provide a control device for an induction motor that saves computation. A second object of the present invention is to solve the problems of the second prior art described above, and even when voltage saturation occurs, the output torque does not become insufficient with respect to the command torque as much as possible, and the voltage saturation can be eliminated. It is to provide a control device for such an induction motor.

[0048]

According to another aspect of the present invention, there is provided an induction motor control device for detecting an induction motor and a speed (rotation frequency) for detecting a rotation frequency of the induction motor. The result of the calculation of a predetermined function whose coefficient value is the constant of the induction motor including constants related to iron loss is stored as data, and the variables related to the predetermined function are input to input the torque current command and the excitation current. A current ratio storage arithmetic circuit that outputs a ratio of command amplitudes (referred to as a current ratio);
Orthogonal rotation coordinate axis (d- that rotates the exciting current command at the primary frequency by inputting the torque command of the induction motor and the current ratio
(referred to as the q-axis) as a d-axis component command of the primary current on the above, and a torque current command of the induction motor as a q-axis component command of the primary current, and current component command calculation means.
A current detector for detecting the primary current of the induction motor;
An integrator that integrates the next frequency and outputs a phase; a current component calculation means that inputs the output of the current detector and the phase to calculate the d-axis component and the q-axis component of the primary current;
Slip frequency for calculating the slip frequency of the induction motor by inputting at least one of the d-axis component command and d-axis component of the primary current and at least one of the q-axis component command and q-axis component of the primary current A calculating means, an adder for adding the slip frequency output from the slip frequency calculating means and the output of the rotation frequency detector to form the primary frequency, and a d-axis component and a q-axis of the primary current. And a current component control circuit for controlling the primary current of the induction motor so that the components follow the d-axis component command and the q-axis component command of the primary current, respectively. The primary current command is output such that the product value is proportional to the torque command and the amplitude ratio is equal to a predetermined function value using the constant of the induction motor including the constant related to iron loss. thing A.

The induction motor controller according to the invention of claim 2 relating to the first object relates to an induction motor, a speed (rotation frequency) detector for detecting a rotation frequency of the induction motor, and iron loss. The calculation result of a predetermined function having a constant value of the induction motor including the constant as a coefficient value is stored as data, and a variable related to the predetermined function is input to input the ratio of the amplitude of the torque current command and the excitation current command (current Called ratio)
On the orthogonal rotation coordinate axes (referred to as dq axes) for inputting the torque command of the induction motor and the current ratio and rotating the exciting current command at the primary frequency.
A current component command calculating means for outputting a d-axis component command of the next current and a torque current command of the induction motor as a q-axis component command of the primary current, and a current detection for detecting the primary current of the induction motor. Component, an integrator that integrates a primary frequency and outputs a phase, and an output that outputs the phase of the current detector and the phase that are input to calculate a d-axis component and a q-axis component of the primary current. The operation means, at least one of the d-axis component command and the d-axis component of the primary current, and at least one of the q-axis component command and the q-axis component of the primary current are input to determine the slip frequency of the induction motor. The slip frequency calculating means for calculating, the slip frequency output from the slip frequency calculating means (referred to as slip frequency before correction) and the rotation frequency are input, and the induction motor includes a constant related to iron loss. It is approximated from the ratio of the slip frequency obtained from the result of the predetermined function operation with the number as the coefficient value and the slip frequency obtained from the result of the predetermined function operation with the constant of the induction motor not related to iron loss as the coefficient value. The slip frequency iron loss component correcting means for interpolating the slip frequency before correction and outputting the slip frequency after correction by the predetermined linear function obtained by the above, and the slip frequency iron loss component correcting means for outputting An adder that adds the corrected slip frequency and the output of the rotation frequency detector to form the primary frequency;
And a current component control circuit for controlling the primary current of the induction motor so that the d-axis component and the q-axis component of the primary current follow the d-axis component command and the q-axis component command of the primary current, respectively. The value of the product of the output of the current component command computing means is proportional to the torque command, and the amplitude ratio is equal to a predetermined function value using the constant of the induction motor including the constant related to iron loss. A primary current command is output.

The control device for an induction motor according to the invention of claim 3 relating to the above-mentioned first object is claim 1 or claim 2.
In, further comprising a motor temperature detector for detecting the temperature of the stator windings of the induction motor, the predetermined function having a constant value of the induction motor including a constant related to iron loss as a coefficient value is at least a function of the temperature of the motor. In addition, by setting at least one of the variables related to the predetermined function input to the current ratio storage arithmetic circuit to the temperature of the electric motor, high-efficiency operation corresponding to the temperature change of the induction motor is achieved. Is to maintain.

The control device for an induction motor according to the invention of claim 4 relating to the above-mentioned first object, claims 1 to 3.
In any of the above, the d-axis component and the rotation frequency of the primary current on the orthogonal rotation coordinate axes (called dq axes) that rotate at the primary frequency are input, and the excitation current and the rotation frequency and the excitation that are obtained in advance are input. Inductance correction means for obtaining and outputting the exciting inductance from the correspondence data of the inductance is further provided, and the current component command calculation means is a torque command of the induction motor and the current ratio output from the current ratio storage calculation circuit and the inductance correction means. An exciting current command calculation circuit that inputs the exciting inductance to be output and outputs an exciting current command as a d-axis component command of the primary current, and a limit value at which the magnetic flux saturates the maximum value of the d-axis component command of the primary current. And a component proportional to the torque command of the induction motor is divided by the output of the limit circuit to obtain a primary current that is a torque current command. The q-axis component command is used to prevent the torque control performance from degrading even when the magnetic flux is saturated and the exciting inductance fluctuates. .

The induction motor control device according to the invention of claim 5 relating to the above-mentioned first object, claims 1 to 4.
In any of the above, the current ratio data stored in the current ratio storage arithmetic circuit is at least a linear function relating to the rotation frequency.

The control device for an induction motor according to the invention of claim 6 relating to the above-mentioned first object, claims 1 to 4.
In any one of the above, the current ratio data stored in the current ratio storage arithmetic circuit is at least a linear function relating to the primary frequency.

The control device for an induction motor according to the invention of claim 7 relating to the above-mentioned first object is, in claim 2, characterized in that the predetermined linear function in the slip frequency iron loss component correcting means is a function of the primary frequency. It is designed to be

The control device for an induction motor according to the invention of claim 8 relating to the second object uses a voltage type inverter to rotate a torque current and an exciting current at a primary frequency in an orthogonal rotary coordinate axis (d-). In the device for controlling the induction motor by the vector control method in which the q-axis component and the d-axis component of the primary current on the above (q-axis) are separately controlled, a DC for detecting the DC voltage on the power source side of the voltage type inverter is used. 1 which is the torque current command of the voltage detection unit and the induction motor
Input the q-axis component command of the next current (called the uncorrected primary current q-axis component command) and the d-axis component command of the primary current that is the exciting current command (called the uncorrected primary current d-axis component command). A current ratio calculation means (A) for outputting a ratio of the amplitudes (referred to as a current ratio FA), and a maximum output voltage calculation circuit for inputting the power supply side DC voltage and calculating a maximum primary voltage value that can be output. , The primary voltage command on the three-phase coordinate axes or the dq axes and the maximum primary voltage value that can be output are input, and the presence or absence of voltage saturation is detected to determine the degree of voltage saturation as the first saturation amount component. The q-axis component of the primary current that minimizes the primary voltage required to output the torque according to the torque command by inputting the voltage saturation detection means that outputs Amplitude ratio of command and d-axis component command (current ratio FB) , The current ratio calculation means (B) for calculating and outputting the calculation or the result of the calculation from map data, and the first saturation amount component, the current ratio FA and the current ratio FB are input to determine the degree of saturation. 1 after correction accordingly
The amplitude ratio (called current ratio FC) of the q-axis component command of the next current (called the corrected primary current q-axis component command) and the d-axis component command (called the corrected primary current d-axis component command) A current ratio adjusting means for outputting, a torque command for the induction motor, and the current ratio FC are input, and a corrected current for outputting the corrected primary current q-axis component command and the corrected primary current d-axis component command. And a component command calculation means.

In the control device for an induction motor according to the invention of claim 9 related to the above-mentioned second object, in claim 8, even if the current ratio adjusting means adjusts the current ratio FC, the voltage saturation still occurs. When it occurs, the second saturation amount component corresponding to the degree of saturation is detected and output, and the second saturation amount component and the torque command are input to correct the torque command. A torque command correction means is further provided.

[0057]

BEST MODE FOR CARRYING OUT THE INVENTION Before proceeding to a detailed description of the operation of an embodiment of the present invention, first, a vector control system of an induction motor considering iron loss, which is the basis of the first invention, will be described.

First, the voltage-current equation of the induction motor on the dq coordinate axes rotating at the primary frequency is given by the following equation.

[0059]

(Equation 12) However, Rs, Rr, and Rm are 1 of the induction motor, respectively.
Secondary resistance, secondary resistance and iron loss resistance. Ls, Lr and M are the primary self-inductance of the induction motor,
The secondary self-inductance and the mutual inductance. Further, Vds and Vqs are respectively the d of the primary voltage.
The axis component and the q-axis component. Ids and Iqs are the d-axis component and the q-axis component of the primary current, respectively, and Idr and Iqr are the d-axis component and the q-axis component of the secondary current, respectively.

Ω and ωs are the primary frequency and the slip frequency of the induction motor, respectively, and P is the differential operator (= d / dt). In the above equation, iron loss resistance Rm
When the value of is set to zero, it agrees with the voltage-current equation on the dq coordinate axes of the induction motor when iron loss is ignored.

Next, the d-axis component Φdr and the q-axis component Φqr of the secondary magnetic flux of the induction motor when the iron loss is taken into consideration are given by the following equations.

[0062]

(Equation 13)

Further, the generated torque Tm is given by the following equation.

[0064]

[Equation 14]

The equation (20) is established regardless of the presence or absence of iron loss. From the expressions (18) and (19), the d-axis component Φdr of the secondary magnetic flux is affected by the q-axis components Iqs and Iqr of the primary current and the secondary current due to the presence of the iron loss resistance Rm,
On the contrary, the q-axis component Φqr of the secondary magnetic flux is affected by the d-axis components Ids and Idr of the primary current and the secondary current. As a result, it can be seen from the equation (20) that the generated torque Tm is also affected by the iron loss.

Next, the equations (18) and (19) are changed to (16) and
Substituting into equation (17) and eliminating Ids and Iqs, the following equation is obtained.

[0067]

(Equation 15)

Further, by modifying the equations (18) and (19), I
When dr and Iqr are obtained, the following equation is obtained.

[0069]

(Equation 16)

Next, when the condition for Φqr = 0 is obtained,
From equation (22), it is understood that the slip frequency ωs should be given by the following equation.

[0071]

[Equation 17]

Since it is difficult to directly detect Iqr, the following formula is obtained by substituting the formula (24) into the formula (26) and erasing the Iqr.

[0073]

(Equation 18)

Further, by substituting equation (23) into equation (21) to eliminate Idr and set Φqr = 0, the following equation is obtained.

[0075]

[Equation 19]

Therefore, if the induction motor is controlled at the slip frequency ωs that satisfies the expression (27), Φqr is made zero as in the steady state even in the transient state in which the secondary magnetic flux changes with time. It is possible to keep, that is, the direction of the secondary magnetic flux vector can be made to coincide with the d axis. Here, the d-axis component Φdr of the secondary magnetic flux is 1 by the calculation of the equation (28).
It can be obtained from the d-axis component Ids and the q-axis component Iqs of the next current.

D-axis component Ids and q-axis component I of the primary current
If qs is detected, the slip frequency ωs can be obtained by the calculation of equation (27). In equations (27) and (28), the value of k is a function of the primary frequency ω, as is apparent from equation (25). Further, as is well known, the value of the iron loss resistance Rm also changes depending on the primary frequency ω.

For this reason, the primary frequency ω is required when the equations (27) and (28) are calculated. Ω is the rotational frequency ωr and the slip frequency of the induction motor as shown in the equation (2). It is obtained by adding ωs.

Next, since Φqr = 0 always holds,
From equation (20), it can be seen that the following equation holds true not only in the steady state but also in the transient state.

[0080]

(Equation 20)

Using this equation, the d-axis component command Φdr * of the secondary magnetic flux and the q-axis component command Iqr * of the secondary current are obtained according to the torque command input from the outside, and the secondary of the induction motor is obtained. D-axis component Φdr of magnetic flux and q-axis component I of secondary current
If qr is controlled so as to follow these command values, respectively, the generated torque of the induction motor can be controlled as a result.

However, since the d-axis component Φdr of the secondary magnetic flux and the q-axis component Iqr of the secondary current of the induction motor cannot be directly controlled, here, the d-axis component command Φdr * of the secondary magnetic flux and the q-axis of the secondary current. The d-axis component command Ids * and the q-axis component command Iqs * of the primary current are obtained from the component command Iqr *, and the d-axis component Ids and the q-axis component Iq of the primary current of the induction motor are obtained.
s is controlled so as to follow each command value. Therefore, first, the equations (18) and (19) are modified to obtain Ids and Iqs, and the following equations are obtained.

[0083]

(Equation 21)

Equation (21) is substituted into the above equation to eliminate Idr, and

[0085]

(Equation 22) Then, the following equation is obtained.

[0086]

(Equation 23)

Therefore, the d-axis component command Φdr * of the secondary magnetic flux and the q-axis component command Iqr * of the secondary current are converted from the d-axis component command Id of the primary current by the calculation of the equations (34) and (35).
The s * and q-axis component command Iqs * can be obtained.
In both equations, the value of k and the value of Rm change depending on the value of the primary frequency ω, so the value of ω is necessary in the calculation.

In the equation (36), (Lr-M)
Since the value of is equal to the value of the secondary leakage inductance of the induction motor, the value of the time constant Tr3 is much smaller than the value of Tr2. Therefore, in the calculation of equation (35), Tr3
Even if = 0, there is almost no difference in control.

Since the d-axis component command Ids * and the q-axis component command Iqs * of the primary current are obtained as described above, the d-axis component Ids and the q-axis component Iqs of the primary current of the induction motor are calculated as follows. Control may be performed so as to follow the command values.

Here, as is well known, the d-axis component Ids and the q-axis component Iqs of the primary current are the primary currents Ius and Iv.
It is calculated from s and Iws and the primary frequency ω by the following calculation.

[0091]

(Equation 24)

Next,

(Equation 25) When Iws is eliminated from the equations (37) and (38) using the relationship of, the following equation is obtained.

[0093]

(Equation 26)

Therefore, the d-axis component Ids and the q-axis component Iqs of the primary current obtained by the calculation of the above equation may be controlled so as to follow the respective command values.

The above is the vector control method of the induction motor considering the iron loss which is the basis of the first invention, and the d-axis component Φdr of the secondary magnetic flux of the induction motor and the q-axis component Iq of the secondary current.
By operating r, good torque control performance can be realized even when iron loss cannot be ignored.

Next, from the equation (30), it is understood that there are innumerable combinations of Φdr and Iqr for obtaining a generated torque of a certain value. Then, how to select the values of Φdr and Iqr to realize the maximum efficiency operation of the induction motor will be described. First, the differential term is ignored in the equations (14) to (17). Then, the following equation is obtained from the condition of Φqr = 0 and the equation (21).

[0097]

[Equation 27]

The following expressions are obtained by ignoring the differential term in the expressions (14) and (15) and substituting the expression (43).

[0099]

[Equation 28]

Further, in equations (31) and (32), Φ
When qr = 0 and Idr = 0 are set, the following equation is obtained.

[0101]

(Equation 29)

Now, the active power P input to the induction motor
in is given by the following equation as is known.

[0103]

[Equation 30]

Therefore, equations (44) to (47) are converted into equation (48).
Substituting into the equation yields the following equation:

[0105]

(Equation 31)

Next, the mechanical output Pout of the induction motor
Is given by the following equation as is known.

[0107]

(Equation 32)

Here, ωrm is the mechanical rotation frequency of the induction motor, and the electrical rotation frequency ωr has the following relationship.

[0109]

[Equation 33]

By substituting the equation (26) into the equation (51) and eliminating ωs, the following equation is obtained.

[0111]

(Equation 34)

Therefore, the equations (30) and (52) are changed to (5
By substituting into equation (0) and eliminating Tm and ωrm, the following equation is obtained.

[0113]

(Equation 35)

Then, from equations (49) and (53), the loss Ploss generated in the induction motor is given by the following equation.

[0115]

[Equation 36]

Under the condition that the generated torque is constant, (5
Φ such that the loss Ploss shown in the equation 4) is minimized
It can be seen that the maximum efficiency operation can be realized by obtaining the combination of dr and Iqr. Therefore, next, the loss Plos
A combination of Φdr and Iqr that minimizes s is found. First, the condition that the generated torque is constant is expressed by the following equation from the relation of the equation (30).

[0117]

(37)

Then, by substituting the equation (55) into the equation (54) and eliminating Φdr, the following equation is obtained.

[0119]

(38)

From the equation (56), it can be seen that the loss Ploss becomes the minimum when the condition that the value obtained by differentiating the right side of the equation (56) by Iqr ² is zero is satisfied. Therefore, the relationship between Φdr and Iqr when the loss Ploss is the minimum is given by the following equation from this condition and the equation (55).

[0121]

[Equation 39]

That is, the ratio of the amplitudes of Φdr and Iqr is (5
Maximum efficiency operation can be realized by controlling Φdr and Iqr according to the torque command so as to satisfy the expression (7). still,
The expression (57) is derived by ignoring the differential term in the expressions (14) to (17), but the expressions (14) to (17) express the voltage / current equation of the induction motor on the rotating coordinate axis as described above. It represents.

Therefore, when the induction motor is accelerated or decelerated with a constant torque, the differential terms of the equations (14) to (17) are zero, and therefore the maximum efficiency operation can be realized by the equation (57).

As a procedure for realizing the maximum efficiency operation as described above, the d-axis component command Φdr * of the secondary magnetic flux and the q-axis component command Iqr * of the secondary current satisfying the expression (57) according to the torque command are set. It may be determined so that the d-axis component Φdr of the secondary magnetic flux and the q-axis component Iqr of the secondary current follow the respective commands. However, as described above, Φdr, Iq
Since r cannot be directly controlled, the d-axis component command Ids * and the q-axis component command Iq of the primary current are calculated from Φdr * and Iqr *.
s * is calculated, and d-axis component Ids and q-axis component I of the primary current
qs is controlled so as to follow each command value.

That is, Φdr * and Iqr * corresponding to the torque command Tm * and the primary frequency ω are obtained using the equations (30) and (57), and this is obtained using the equations (34) and (35). Φdr * and Iqr * are converted into Ids * and Iqs *, and Ids and Iqs are controlled so as to follow their respective command values.

However, the d-axis component command Ids * and the q-axis component command Iqs * of the primary current are based on the above procedure.
The calculation process is complicated until the above is obtained, and a high calculation capacity is required to realize it. Therefore, this calculation process is simplified to reduce the calculation of Ids * and Iqs * generation.

First, the absolute value of the ratio of the amplitudes of Iqs * and Ids * is expressed by the following equation from the equations (34) and (35).

[0128]

(Equation 40)

If the absolute value | ω | of the primary frequency is determined (5
Equation (7) is determined, and therefore equation (58) is determined. Therefore, | Iqs * / Ids * | corresponding to | ω |
Can be obtained in advance and converted into map data.

The generated torque Tm is expressed by the following equation using the d-axis component Ids and the q-axis component Iqs of the primary current.

[0131]

[Equation 41]

Here, by using the fact that the signs of Tm * and Iqs * match and using | Iqs * / Ids * | and the equation (59), Tm = Tm *, Ids = Ids *, Iqs = Iq.
Substituting for s *, we get:

[0133]

(Equation 42)

As described above, with respect to the primary current command generation, | Iqs * / Iqs * | that satisfies the maximum efficiency condition (57) is calculated from the map data using the primary frequency ω, and this and the torque command Tm are calculated. * From to (60), (61)
The calculation can be reduced by using the procedure of obtaining Ids * and Iqs * using the formula.

Next, the slip frequency ωs is calculated from the equation (27) using the d-axis component Φdr of the secondary magnetic flux, the d-axis component Ids of the primary current and the q-axis component Iqs. Φ
dr is calculated by the equation (28) using Ids and Iqs. However, the process of calculating the slip frequency in consideration of these iron losses becomes complicated and requires a high computing ability. Therefore, for the purpose of effectively utilizing the limited computing power, the computations of the equations (27) and (28) are simplified to reduce the computation of ωs.

First, when iron loss is not taken into consideration, Φdr is given by the following equation by setting k = 0 in equation (28).

[0137]

[Equation 43]

Therefore, the slip frequency ωs 'in the case where the iron loss is not taken into consideration (expressed as ωs' in order to distinguish it from the slip frequency taking the iron loss into consideration) is k = 0 in the equation (27).
When the equation (62) is used, it is expressed as the following equation.

[0139]

[Equation 44]

Here, when the ratio of the slip frequency ωs when the iron loss is taken into consideration and the slip frequency ωs ′ when the iron loss is not taken into account is obtained, as shown in FIG. Take the form of the following function. Therefore, the ratio of ωs and ωs ′ is approximated to a linear function as in the following equation.

[0141]

[Equation 45]

Here, N and O are constants of the induction motor and constants determined by operating conditions. Using the equation (64), the slip frequency ωs considering the iron loss is calculated by the following equation from the slip frequency ωs ′ not considering the iron loss and the rotation frequency ωr.

[0143]

[Equation 46]

As described above, by using the formulas (63) to (65), it is possible to reduce the calculation required for calculating the slip frequency ωs.

The conditional expression for the maximum efficiency control expressed by the expression (57) includes the primary resistance Rs and the secondary resistance Rr. As is known, the values of Rs and Rr depend on temperature.
For example, when the stator winding is made of copper, the relationship between the temperature and the value of the primary resistance Rs is as follows.

[0146]

[Equation 47]

Similarly, Rr is also expressed as a function of temperature. Therefore, Rs and R are changed by the temperature change of the motor.
The r value fluctuates, and the value on the right side of the expression (57) changes. As a result, the right side of the equation (58) also changes, so it can be said that the d-axis component command Ids * and the q-axis component command Iqs * of the primary current for performing maximum efficiency control are variables that depend on the temperature of the electric motor.

Embodiments of the present invention will be described below with reference to the accompanying drawings. Embodiment 1. 1 is a block diagram showing the configuration of an induction motor control device according to a first embodiment of the present invention. still,
23, the same reference numerals as those in FIG. 23 indicate the same or corresponding portions.
In FIG. 1, 2a is a q-axis component of a primary current on an orthogonal rotation coordinate axis (referred to as d-q axes) that rotates at a primary frequency that is a torque current command based on the rotation frequency ωr input from the speed detector 7. It is a current ratio storage arithmetic circuit that outputs the amplitude ratio (referred to as current ratio) between the command Iqs * and the d-axis component command Ids * of the primary current that is the exciting current command. Reference numeral 3a is a torque command Tm * of the induction motor 1 and a current ratio storage arithmetic circuit 2a.
The current component command calculation means outputs the d-axis component command Ids * and the q-axis component command Iqs * of the primary current based on the current ratio input from. 12 is the output I of the current detector 4.
It is a current component calculating means for calculating the d-axis component Ids and the q-axis component Iqs of the primary current by inputting us and Ivs and a phase θ obtained by integrating a primary frequency ω described later. Further, 8 is a slip frequency calculating circuit for inputting the d-axis component Ids and the q-axis component Iqs of the primary current and calculating a slip frequency ωs ′ when the iron loss of the induction motor 1 is not taken into consideration, and 9 is a slip frequency calculating circuit. An adder that adds the slip frequency ωs ′ when the iron loss output from 8 is not taken into consideration and the rotation frequency ωr that is the output of the speed detector 7 to form a primary frequency ω, and 11 integrates the primary frequency ω Then, the integrator 5 for outputting the phase θ to the current component calculating means 12 and the voltage command calculating circuit 6 to be described later is provided with the d-axis component Ids and the q-axis component I of the primary current.
The primary current of the induction motor 1 is controlled so that qs follows the d-axis component command Ids * and the q-axis component command Iqs *, and the d-axis component command Vds * and the q-axis component command Vqs * of the primary voltage are controlled. An output current component control circuit 6 is a primary voltage command V based on Vds * and Vqs * output from the current component control circuit 5 and the phase θ output from the integrator 11.
us * (U phase), Vvs * (V phase), Vws * (W phase)
Is a voltage command calculation circuit for outputting to the PWM inverter 10.

FIG. 2 is a block diagram showing a detailed structure of the current ratio storage operation circuit 2a. As shown in this figure, the current ratio storage operation circuit 2a receives the rotation frequency ωr and calculates the absolute value | ωr | of the absolute value circuit 23 and | ωr |
Is input, and the q-axis component command Iqs * and the d-axis component command Ids * of the primary current are input based on the stored map data.
And a map data section 24a for outputting the ratio of the amplitudes (current ratio | Iqs * / Ids * |).

FIG. 3 is a block diagram showing a detailed structure of the current component command calculating means 3a. As shown in this figure, the current component command calculating means 3a includes a coefficient unit 31 that multiplies the torque command Tm * by a predetermined coefficient, and outputs it, and an absolute value circuit 32 that obtains the absolute value of the output of the coefficient unit 31. The output of the absolute value circuit 32 is used as a numerator, and the current ratio storage operation circuit 2a is used.
The current ratio | Iqs * / Ids * |, which is the output of I, is input as a denominator, and a divider 33 that divides to obtain Ids * ²
The square root of ds * ² is calculated and the d-axis component command I of the primary current is calculated.
The square root calculation circuit 34 that outputs ds * and the torque command T
A code discriminator 35 for inputting m * and outputting 1 when Tm *> 0, −1 when Tm * <0, and 0 when Tm * = 0, and its code discriminator. Multiplier 36 for multiplying the output of 35 by the current ratio | Iqs * / Ids * |
And Ids *, which is the output of the square root calculation circuit 34, and the output of the multiplier 36, are multiplied to obtain the q-axis component command I of the primary current.
It is composed of a multiplier 37 that outputs qs *.

FIG. 4 is a block diagram showing a detailed structure of the slip frequency calculation circuit 8. As shown in this figure, the slip frequency calculation circuit 8 includes a coefficient unit 41 for multiplying the q-axis component Iqs of the primary current by a predetermined coefficient, and the d-axis component I of the primary current I.
A first-order lag circuit 42 that inputs ds and uses Tr as a time constant to output a first-order lag component, a coefficient unit 43 that multiplies the output of the first-order lag circuit 42 by a predetermined coefficient, and outputs the result, and a coefficient unit 41 thereof. Is used as the numerator, the output of the coefficient unit 43 is used as the denominator, and a divider 44 that outputs the slip frequency ωs ′ without considering the iron loss is configured.

Next, the operation of the first embodiment will be described. The current ratio storage arithmetic circuit 2a inputs the rotation frequency ωr and satisfies the ratio of the amplitude of the q-axis component command Iqs * and the d-axis component command Ids * of the primary current that satisfies the condition (57) for maximum efficiency control (current ratio | Iqs * / Ids * |) is output.
The inside of the current ratio storage operation circuit 2a has the configuration shown in FIG. 2, and the absolute value circuit 23 uses the absolute value | ωr of the rotation frequency ωr.
| Is obtained and the corresponding current ratio is extracted based on the map data of the current ratio (equation (58)) satisfying the expression (57) stored in advance in the map data unit 24a.

Next, the torque command Tm * and the current ratio | Iqs
* / Ids * | is input to the current component command calculation means 3a,
The d-axis component command Ids * and the q-axis component command Iqs * of the primary current are output using the relationships of the equations (60) and (61).

The specific operation of the current component command calculating means 3a will be described below with reference to FIG. Torque command Tm *
Is input to the absolute value circuit 32 through the coefficient unit 31. Ids * ² is obtained by inputting the obtained absolute value to the divider 33 with the numerator and the current ratio | Iqs * / Ids * | as the denominator. When the Ids * ² is input to the square root calculation circuit 34, Ids * represented by the equation (60) is output. When the torque command Tm * is input to the sign discriminator 35,
1 when Tm *> 0, -1 when Tm * <0,
When Tm * = 0, 0 is output. Code discriminator 35
Of the output of the power supply and the current ratio | Iqs * / Ids * |
When the output of the multiplier 36 and the output Ids * of the square root calculation circuit 34 are multiplied by the multiplier 37, Iqs * represented by the equation (61) is obtained.

Next, 1 detected by the current detector 4
Next current Ius (U phase), Ivs (V phase) and integrator 11
When the phase θ calculated by is input to the current component calculation means 12, the calculation of equations (41) and (42) is performed, and the d-axis component Ids and the q-axis component Iqs of the primary current are output.

Subsequently, when the d-axis component Ids and the q-axis component Iqs of the primary current output from the current component calculating means 12 are input to the slip frequency calculating circuit 8, the equation (63) is calculated and the iron loss is considered. The slip frequency ωs ′ that does not occur is output. Specifically, as shown in FIG. 4, a product obtained by multiplying the q-axis component Iqs of the primary current by the coefficient of the coefficient unit 41 is obtained by passing the d-axis component Ids of the primary current through the primary delay circuit 42. When the output obtained by multiplying the coefficient of the coefficient unit 43 is input to the divider 44 as the denominator, the operation of the equation (63) is performed and ω
s' is output.

Next, the rotation frequency ωr output from the speed detector 7 and ω output from the slip frequency calculation circuit 8
The primary frequency ω is obtained by adding s ′ by the adder 9. Further, when the primary frequency ω is integrated by the integrator 11, (3
The phase θ is obtained as shown in the equation (9).

Subsequently, the d-axis component command Ids * and the q-axis component command Iqs * of the primary current output from the current component command calculating means 3a, and the d-axis component of the primary current output from the current component calculating means 12 Ids, the q-axis component Iqs, and the primary frequency ω are input to the current component control circuit 5, and Ids is Id.
Amplify the deviation between Ids * and Ids so as to follow s * (P
(I control), the d-axis component command Vds * of the primary voltage is output. Similarly, when the deviation between Iqs * and Iqs is amplified (PI control) so that Iqs follows Iqs *, the q-axis component command Vqs * of the primary voltage is output.

Next, the voltage command calculation circuit 6 outputs the d-axis component command Vds of the primary voltage output from the current component control circuit 5.
Input * and the q-axis component command Vqs *, and enter (6
7), (68), (69) according to the primary voltage command Vu
It outputs s * (U phase), Vvs * (V phase), and Vws * (W phase). Here, since the equations (37), (38), and (40) are similarly established for the voltage, Ius, Ivs, Iws, Ids, and Iqs in these equations are Vus *, Vvs *, Vws *, and Vds, respectively. After replacing with *, Vqs * and solving for Vus *, Vvs *, and Vws *, the following equation is obtained.

[Equation 48]

Then, by the PWM inverter 10,
The primary voltage applied to the induction motor 1 for each phase is dq
On-axis control is performed so as to follow the primary voltage command output from the current component control circuit 5. As a result, the d-axis component Ids and the q-axis component Iqs of the primary current of the induction motor 1 follow the respective commands, and the maximum efficiency condition (5
It becomes possible to operate the induction motor 1 based on the equation (7).

Embodiment 2. FIG. 5 is a block diagram showing a configuration of an induction motor control device according to a second embodiment as an embodiment related to the first invention. In FIG. 1, FIG.
The same reference numerals indicate the same or corresponding parts. In FIG. 5, in the second embodiment, the slip frequency (pre-correction slip frequency) ωs ′ that does not consider the iron loss output from the frequency calculation circuit 8 is changed to the slip frequency (correction slip frequency) ωs that takes the iron loss into consideration. Embodiment 1 of FIG. 1 except that a slip frequency iron loss component correcting means 13 for correcting
Is the same as

FIG. 6 shows slip frequency iron loss component correction means 13
FIG. 3 is a block diagram showing a detailed configuration of FIG. As shown in this figure, the slip frequency iron loss component correction means 13 receives the rotation frequency ωr and calculates an absolute value | ωr | of the absolute value circuit 51, and inputs | ωr | to rotate the induction motor 1. A linear function calculator 52 that outputs the calculation result of a predetermined linear function to the frequency ωr as a slip frequency ratio J, and a slip frequency ratio J output from the linear function calculator 52 and a slip frequency calculation circuit 8 And a multiplier 53 that outputs the slip frequency ωs considering the iron loss by multiplying the slip frequency ωs ′ not considering the iron loss.

A specific operation of the slip frequency iron loss component correction means 13 will be described with reference to FIG. Rotation frequency ωr
Is input to the absolute value circuit 51 to obtain | ωr |
The r | is input to the linear function calculator 52. Then, (6
Slip frequency ωs' that does not consider iron loss and slip frequency ω that considers iron loss, which is approximated by a linear function expressed by equation (4).
The ratio J to s (slip frequency ratio) J is output. When the output J of the linear function calculator 52 and the slip frequency ωs ′ output from the slip frequency calculation circuit 8 which does not consider the iron loss are multiplied by the multiplier 53, the iron loss is considered based on the equation (65). The slip frequency ωs is obtained. The slip frequency ratio J approximated by a linear function is expressed as shown in FIG.

Next, the rotation frequency ωr output from the speed detector 7 and the slip frequency ωs considering the iron loss output from the slip frequency iron loss component correcting means 13 are added by the adder 9 to obtain 1 The next frequency ω is obtained.

Embodiment 3 FIG. 8 is a block diagram showing a configuration of an induction motor control device according to a third embodiment as an embodiment related to the first invention. In the figure, the same reference numerals as those in FIG. 1 indicate the same or corresponding portions. In this embodiment, as shown in FIG. 8, an electric motor temperature detector 14 for detecting the temperature of the stator winding of the induction motor 1 is provided, and the output of the electric motor temperature detector 14 is stored in the current ratio storage arithmetic circuit 2b. It is input and reflected in the calculation of the current ratio.

That is, as described above, when the stator winding is copper, the value of the primary resistance Rs fluctuates according to the equation (66) due to changes in the motor temperature. Therefore, (5
Since the value on the right side of equation (7) changes with temperature,
The right side of equation (8) also changes. From this, the ratio of the amplitudes of the q-axis component command Iqs * of the primary current and the d-axis component command Ids * that realize the maximum efficiency control (current ratio | Iqs * / Ids *
| Is a function of motor temperature.

Therefore, the current ratio | Iqs * / Ids * stored in the map data section 24b of the current ratio storage operation circuit 2b.
The data of │ (57), considering the variation with temperature,
It is assumed that the results calculated according to the equations (58) and (66) are converted into map data. FIG. 9 shows the current ratio storage arithmetic circuit 2b.
FIG. 3 is a block diagram showing a detailed configuration of FIG. That is, the current ratio storage calculation circuit 2b is configured to detect the rotation frequency ωr and the motor temperature detector 1
The electric motor temperature, which is the output of No. 4, is input, and the current ratio | Iqs * / Ids * | corresponding to the state of each input is output.

Fourth Embodiment FIG. 10 is a block diagram showing a configuration of an induction motor control device according to a fourth embodiment as an embodiment related to the first invention. In the figure, the same reference numerals as those in FIG. 1 indicate the same or corresponding portions. In FIG. 10, the fourth embodiment is provided with an inductance correction means 15 which inputs the d-axis component Ids of the primary current and the rotation frequency ωr and outputs the exciting inductance M ′ according to the state of each input. The output of the inductance correction means 15 is input to the current component command calculation means 3b and reflected in the calculation. That is, the current component command calculation means 3b has the torque command Tm * of the induction motor 1, the current ratio output from the current ratio storage calculation circuit 2a, and the exciting inductance M ′ output from the inductance correction means 15.
To output the primary current d-axis component command Ids * and the q-axis component command Iqs *.

FIG. 11 is a block diagram showing a detailed structure of the inductance correction means 15. As shown in this figure, the inductance correction means 15 receives the rotation frequency ωr and calculates the absolute value | ωr | of the absolute value circuit 61.
And | ωr | and the d-axis component Ids of the primary current are input, and the map data unit 62 that outputs the exciting inductance M ′ based on the map data is configured.

FIG. 12 is a block diagram showing the detailed structure of the current component command calculating means 3b. The current component command calculation means 3b inputs the torque command Tm *, the current ratio | Iqs * / Ids * | output from the current ratio storage calculation circuit 2a, and the exciting inductance M ′ output from the inductance correction means 15 and receives the primary Current d-axis component command Ids * and (Tm *
-Lr) / (Pm-M'2) -exciting current command calculating circuit 70 and a limiting circuit for inputting Ids * and limiting the maximum value to the limit value at which the magnetic flux is saturated and outputting 71 and (Tm * · Lr) / (Pm · M′2) as the numerator, and the Ids * output from the limiting circuit 71 is input as a denominator and divided to obtain the q-axis component command Iqs * of the primary current. And a divider 72. Further, the exciting current command arithmetic circuit 70 multiplies the torque command Tm * by a predetermined coefficient, and outputs the coefficient unit 73 and the inductance correction means 15.
The output of the multiplier 74, which squares the exciting inductance M ′, which is the output of the above, and the output of the coefficient unit 73 are used as the numerator, the output of the multiplier 74 is input as the denominator, and the division (Tm * · L
r) / (Pm · M′2), a divider 75, an absolute value circuit 76 for obtaining the absolute value of the output of the divider 75, and an output of the absolute value circuit 76 as a numerator The divider 77 which inputs the current ratio | Iqs * / Ids * | which is the output of 2a as a denominator and divides to obtain Ids * ²
And the square root of Ids * ² is calculated to calculate the d-axis component command Ids
And a square root calculation circuit 78 that outputs *.

Next, the operation of the fourth embodiment will be described. When the rotation frequency ωr and the d-axis component Ids of the primary current are input to the inductance correction means 15, the absolute value circuit 61 determines the absolute value | ωr | of the rotation frequency inside the inductance correction means 15, and this | ωr The excitation inductance M ′ associated with | and Ids is output from the map data unit 62. This is the output of the inductance correction means 15. Next, the torque command Tm
*, Exciting inductance M ', and current ratio | Iqs * /
When Ids * | is input to the current component command calculation means 3b,
Inside thereof, the torque command Tm * multiplied by the coefficient of the coefficient multiplier 73 is used as a numerator, and the exciting inductance M ′ is multiplied by the multiplier 7
The value obtained by squaring by 4 is used as the denominator, and the divider 75 obtains (Tm * · Lr) / (Pm · M ′ ² ). Further, (Tm * · Lr) / (Pm · M ′ ² ) is passed through the absolute value circuit 76 to obtain | (Tm * · Lr) / (Pm ·
M ^'2) | was used as a molecule, the current ratio | Iqs * / Ids * |
With I as the denominator, Ids * ² is obtained by the divider 77, and Ids * is obtained by the square root operation circuit 78. The above calculation process is based on the equation (60). Next, Ids
The maximum value of * is limited by the limiting circuit 71 to a limit value at which the magnetic flux is saturated. Furthermore, (Tm * ・ Lr) / (Pm ・
Ids * that has passed through the limiting circuit 78 with M ′ ² ) as the numerator
When the denominator is used as the denominator, the divider 72 divides it to obtain Iqs * by the operation based on the equation (59).

Embodiment 5. In the above-described first to fourth embodiments, the map data unit 24 of the current ratio storage arithmetic circuit 2a.
Alternatively, the current ratio map data stored in the map data unit 24b of the current ratio storage / calculation circuit 2b may be data that is linearly approximated at least with respect to the rotation frequency ωr. An example of the relationship of the current ratio | Iqs * / Ids * | with respect to | ωr | in this case is shown in FIG.

Sixth Embodiment In the current ratio storage arithmetic circuit 2a or 2b of the fifth embodiment, the primary frequency ω output from the adder 9 may be used instead of inputting the rotation frequency ωr output from the speed detector 7.

Seventh Embodiment In the slip frequency iron loss component correction means 13 of the second embodiment, the primary frequency ω output from the adder 9 may be used instead of inputting the rotation frequency ωr output from the speed detector 7.

Before describing the embodiment of the second invention, a vector control method for generating a predetermined torque, which is the basis of the second invention, with a minimum primary voltage will be described.
The voltage of the induction motor on the dq axes that rotates at the primary frequency
The current equation is expressed by the following equation from equations (14) to (17) as is well known when the steady state is considered without assuming the iron loss.

[0176]

[Equation 49] However, Rs and Rr are the primary resistance and secondary resistance of the induction motor, respectively. Ls, Lr and M are the primary self-inductance, secondary self-inductance and mutual inductance of the induction motor, respectively. Further, Vds and Vqs are the d-axis component and the q-axis component of the primary voltage, respectively. Ids and Iqs are the d-axis component and the q-axis component of the primary current, respectively, and Idr and Iqr are 2 respectively.
These are the d-axis component and the q-axis component of the next current. ω and ωs ′ are the primary frequency of the induction motor and the slip frequency that does not assume iron loss, respectively.

Here, in order to match the direction of the secondary magnetic flux vector with the d-axis, if Φqr is set to zero among the d-axis component Φdr and the q-axis component Φqr of the secondary magnetic flux, the slip frequency in the known vector control is arranged. ωs ′ is expressed by the equation (63), and the generated torque Tm is expressed by the equation (59).
The d-axis component Φdr of the next magnetic flux and the voltage / current equation are respectively expressed by the following equations.

[0178]

[Equation 50]

From the equations (75) and (76), the primary voltage Vs
Becomes as follows.

[0180]

(Equation 51)

Here, consider the case where the generated torque Tm = TA and the primary voltage Vs = VsA. From equation (59), Ids
Iqs can be expressed as the following equation.

[0182]

[Equation 52]

The expression (78) is substituted into the expression (77), and the expression is transformed into the following expression.

[0184]

(Equation 53)

Here, the ellipse expressed by the equation (79) and (7
When the curve represented by the equation (8) touches at one point, the combination of the d-axis component Ids and the q-axis component Iqs of the primary current at the contact point generates the torque TA and the primary voltage is V.
It is the only combination that is sA. This pattern is shown in FIG.
Shown in Other than this, Ids and Iq that satisfy the expression (78)
In the combination of s, a value larger than VsA is required as the primary voltage in order to generate the torque TA. In other words, if the combination of Ids and Iqs in the case where the ellipse represented by the equation (79) and the curve represented by the equation (78) are in contact at one point, the required torque is generated at the minimum primary voltage, or Maximum torque can be generated with a given primary voltage.

The contact point between the ellipse represented by the equation (79) and the curve represented by the equation (78) is obtained as follows.
When Iqs is expressed using Ids from the expression (78), the following expression is obtained.

[0187]

(Equation 54)

Substituting equation (80) into equation (79) and multiplying both sides by Ids ² and rearranging yields the following equation.

[0189]

[Equation 55]

Here, the condition that equations (79) and (78) contact at one point is

[0191]

[Equation 56] Than,

[0192]

[Equation 57]

Therefore, it is expressed by the following equation.

[0194]

[Equation 58]

At this time, Ids is

[0196]

[Equation 59] Therefore, the combination of Ids and Iqs is represented by the following expression when expressed by the ratio of the amplitudes of Iqs and Ids.

[0197]

[Equation 60]

From the above, based on the equation (86), 1
By controlling the d-axis component Ids and the q-axis component Iqs of the next current, it is possible to generate a required torque with a minimum primary voltage or generate a maximum torque with a predetermined primary voltage. Hereinafter, the control method based on this method will be referred to as the minimum primary voltage control method.

[Embodiment 8] FIG. 16 is a block diagram showing a configuration of an induction motor control device according to an eighth embodiment as an embodiment related to the second invention. In the figure, the same reference numerals as those in FIG. 1 indicate the same or corresponding portions. 16, the induction motor control device according to the embodiment has the following components in place of the component 5-12 of the first embodiment shown in FIG. 1 instead of the current component command calculation means 3a and the current ratio storage calculation circuit 2a. It is provided with components. That is, the induction motor control device includes a direct current power supply 25 that supplies forward-converted alternating current or a direct current voltage, a direct current voltage detector 16 that detects the voltage value of the direct current voltage that the direct current power supply 25 supplies, and a direct current voltage detector. A maximum output voltage calculation circuit 17 that calculates and outputs the maximum value Vsmax of the primary voltage that can be output based on the DC voltage value input from the unit 16, and the maximum output voltage calculation circuit 17
Whether voltage saturation occurs based on the maximum primary voltage value Vsmax that can be output from the current component control circuit 5, the d-axis component command Vds * of the primary voltage input from the current component control circuit 5, and the q-axis component command Vqs * Is detected and the degree of voltage saturation is output as the first saturation amount component Vovr1, and the rotation frequency ω output by the speed detector 7 is detected.
A current that outputs a ratio (current ratio) FB = | IqsB * / IdsB * | of the amplitudes of the q-axis component command IqsB * and the d-axis component command IdsB * of the primary current in the minimum primary voltage control based on r A ratio calculation means (B) 19, a q-axis component command Iqs * of the primary current, which is a torque current command output from a current command generator of a conventional device (not shown) (called a pre-correction primary current q-axis component command), and A d-axis component command Ids * of the primary current, which is an exciting current command (called a pre-correction primary current d-axis component command), is input and the amplitude ratio (current ratio) FA = | Iqs * /
Current ratio calculation means (A) 20 that outputs Ids * | and the first saturation amount component Vov that the voltage saturation detection means 18 outputs
r1 and the current ratio FB output by the current ratio calculating means (B) and the current ratio FA output by the current ratio calculating means (A) are input, and the corrected primary current q is corrected according to the magnitude of Vovr1.
Axis component command Iqs * '(called corrected primary current q-axis component command) and d-axis component command Ids *' (corrected primary current d)
Amplitude ratio (current ratio) FC = | Iq
Current ratio adjusting means 21a for outputting s * '/ Ids *' |
And 1 after correction based on the torque command Tm * of the induction motor 1 and the current ratio FC input from the current ratio adjusting means 21a.
The post-correction current component command calculation means 30a for outputting the next current d-axis component command Ids * 'and the corrected primary current q-axis component command Iqs *'.

FIG. 17 is a block diagram showing the detailed structure of the current ratio calculating means (A) 20. As shown in this figure,
The current ratio calculation means (A) 20 is an uncorrected primary current q-axis component command I output from a current command generator of a conventional device (not shown).
With qs * as the numerator, the uncorrected primary current d-axis component command Ids
It is composed of a divider 81 which inputs * as a denominator and performs division, and an absolute value circuit 82 which obtains and outputs an absolute value FA = | Iqs * / Ids * | of the output of the divider 81.

FIG. 18 is a block diagram showing the detailed structure of the voltage saturation detecting means 18. As shown in this figure, the voltage saturation detection means 18 uses the d-axis component Vds * of the primary voltage command.
Multiplier 91 for finding the squared Vds * ² of Vqs * ² and multiplier 9 for finding the squared Vqs * ² of the q-axis component Vqs * of the primary voltage command
2, an adder 93 that adds and outputs Vds * ² and Vqs * ² , a square root calculation circuit 94 that obtains a square root of the output of the adder 93, and a maximum primary voltage value Vsmax that can be output.
And the output of the square root calculation circuit 94 are compared and compared to obtain the deviation thereof, and a limiting circuit 96 for limiting the minimum value of the output of the subtractor 95 to zero.

FIG. 19 is a block diagram showing the detailed structure of the current ratio adjusting means 21a. As shown in this figure, the current ratio adjusting means 21 a multiplies the first saturation amount component Vovr 1 which is the output of the voltage saturation detecting means 18 by a predetermined coefficient K 1 and outputs the coefficient element 101. A limit circuit 102 that limits the maximum output value to 1.0, a current ratio FA that is the output of the current ratio calculating means (A), and a current ratio FB that is the output of the current ratio calculating means (B) are input. Subtractor 103 for obtaining deviation, output of limiting circuit 102 and subtractor 10
3 and the output of the multiplier 104, and the output of the multiplier 104 is subtracted from the current ratio FA to obtain the current ratio FC.
And a subtractor 105 that outputs

Next, the operation of the eighth embodiment will be described. When the uncorrected primary current q-axis component command Iqs * and the uncorrected primary current d-axis component command Ids * output from the current command generator of the conventional device (not shown) are input to the current ratio calculation means (A) 20, The current ratio FA = | Iqs * / Ids * | is output through the divider 81 and the absolute value circuit 82. Further, when the rotation frequency ωr output from the speed detector 7 is input to the current ratio calculating means (B) 19, the q-axis component command Iq of the primary current in the minimum primary voltage control based on the equation (86).
The amplitude ratio (current ratio) FB = | IqsB * / IdsB * | of sB * and the d-axis component command IdsB * is calculated and output from map data of the calculation result.

Subsequently, the DC voltage detection unit 16 changes the DC power supply 2
The maximum output voltage calculation circuit 17 detects the voltage value Vdc of 5
Output to In the maximum output voltage calculation circuit 17, (12)
The maximum primary voltage value Vsmax that can be output is calculated and output based on the equation.

Next, the maximum primary voltage value Vsmax that can be output to the voltage saturation detection means 18 and the primary axis d-axis component command V
When ds * and the q-axis component command Vqs * are input, the presence or absence of voltage saturation is detected, and the degree of voltage saturation is first determined.
Is output as the saturation amount component Vovr1. Below, FIG.
The specific operation of the voltage saturation detection means 18 will be described using.

The d-axis component command Vds * and the q-axis component command Vqs * of the primary voltage are squared by the multiplier 91 and the multiplier 92, respectively, and are added by the adder 93 to Vds * ² + Vqs * ²
Is calculated. Then, the square root calculation circuit 94 ((1
The primary voltage command Vs * is output by the calculation based on the equation (0). Next, the maximum primary voltage values Vsmax and Vs that can be output
The subtractor 95 matches * and the deviation is input to the limiting circuit 96. The limiting circuit 96 limits the minimum value of the deviation to 0 and outputs it as the first saturation amount component Vovr1. When Vs * <Vsmax, voltage saturation has not occurred and the deviation is negative, so Vovr1 = 0 is output. When Vs * ≧ Vsmax, voltage saturation has occurred and the deviation is positive, so an amount proportional to the deviation is output as Vovr1.

Next, when the first saturation amount component Vovr1 and the current ratio FB and the current ratio FA are input to the current ratio adjusting means 21a, when voltage saturation occurs, Vovr1
The current ratio FA of the uncorrected primary current command is shifted so as to be closer to the current ratio FB in the minimum primary voltage control, and the correction calculation is performed, and the corrected current ratio FC is output. Hereinafter, a specific operation of the current ratio adjusting means 21a will be described with reference to FIG.

The current ratio FA and the current ratio FB are subtracted by the subtractor 103.
And the deviation (FA-FB) is obtained. Further, the first saturation amount component Vovr1 is multiplied by K1 by the coefficient unit 101, then the maximum value is limited to 1.0 by the limiting circuit 102, and (FA-FB) is multiplied by the multiplier 104. As a result, the adjustment amount ΔF = (FA−FB) · K1 · Vovr1 when the current ratio is shifted from FA to FB.
Is calculated. The adjustment amount ΔF is subtracted from the current ratio FA by the subtractor 105, and the corrected current ratio FC (= FA−Δ
F) is output. Here, the limiting value of 1.0 in the limiting circuit 102 is a value resulting from adjusting the current ratio FC to shift to FB, and finally controlling to prevent further shifting when FC = FB. And is obtained as follows. When FC = FB, the following equation is obtained.

[0209]

[Equation 61]

The above equation can be summarized as follows for the current ratio FA and the current ratio FB.

[0211]

(Equation 62)

Therefore, when FA ≠ FB, FC and FB coincide with each other when 1−K1 · Vovr1 = 0, that is,

[0213]

[Equation 63] In the case of. Therefore, by limiting the maximum value of K1 · Vovr1 to 1.0, the FC shift is stopped when FC = FB. When FA = FB, ΔF = 0 is obtained from FA-FB = 0.

Next, when the torque command Tm * and the current ratio FC are input to the corrected current component command calculating means 30a, (6
0), the corrected primary current d-axis component command Ids * 'and the corrected primary current q-axis component command Iqs using the relationship of equations (0) and (61).
* 'Is output. The configuration and operation of the corrected current component command calculation means 30a are the same as those of the current component command calculation means 3a in the first embodiment.

Subsequently, the d-axis component Ids and the q-axis component Iqs of the primary current are converted into the corrected primary current d-axis component command Ids * 'and the corrected primary current q-axis component based on the known vector control method. The commands Iqs * 'are controlled so as to follow each.

By the above procedure, the presence / absence of voltage saturation is detected, and the d-axis component command of the primary current and the d-axis component command of the primary current in the minimum primary voltage control are determined according to the degree of saturation. By approaching the q-axis component command and the q-axis component command, the output torque according to the command torque can be obtained while eliminating the voltage saturation state. This pattern is shown in FIG. In addition, the voltage saturation detecting means 18 causes the d-axis component command Vds * of the primary voltage.
Also, instead of calculating the primary voltage command Vs * from the q-axis component command Vqs *, the primary voltage phase command Vus * (U
Phase), Vvs * (V phase), Vws * (W phase) to Vs *
May be calculated. Further, Vs * matched inside the voltage saturation detection means 18 and the maximum primary voltage value V that can be output
smax may be a Vs * ² and Vsmax ^2.

Embodiment 9 FIG. FIG. 20 is a block diagram showing a configuration of an induction motor control device according to a ninth embodiment as an embodiment related to the second invention, and FIG.
The same reference numerals indicate the same or corresponding parts. 20, the ninth embodiment is the same as the eighth embodiment of FIG. 16 except that the current ratio adjusting means 21b and the torque command correcting means 22 are provided. The current ratio adjusting means 21b outputs the first saturation amount component Vov output by the voltage saturation detecting means 18.
r1 and the current ratio FB output by the current ratio calculation means (B) and the current ratio FA output by the current ratio calculation means (A) are input and V
A q-axis component command Iqs * ′ of the primary current after correction according to the magnitude of ovr1 (called a corrected primary current q-axis component command)
And the amplitude ratio (current ratio) of the d-axis component command Ids * '(called the corrected primary current d-axis component command) FC = | Iqs *'
/ Ids * '| and the second saturation amount component Vovr2 indicating the degree of voltage saturation when voltage saturation still occurs even after using the corrected primary current command. The torque command correction means 22 includes a torque command Tm * and a current ratio adjustment means 2.
The second saturation amount component Vovr2 output by 1b is input, the torque command is corrected according to the magnitude of Vovr2, and the corrected torque command Tm * ′ is output.

FIG. 21 is a block diagram showing the detailed structure of the current ratio adjusting means 21b. As shown in this figure, the current ratio adjusting means 21b outputs 0 when the output of the coefficient unit 101 is 1.0 or less with respect to the current ratio adjusting means 21a shown in FIG. Has a configuration in which a saturation amount calculation circuit 106 that outputs an amount exceeding 1.0 as the second saturation amount component Vovr2 is added.

FIG. 22 is a block diagram showing a detailed structure of the torque command correction means 22. As shown in this figure, the torque command correcting means 22 inputs 1 when the torque command Tm * is input and Tm *> 0, and -1 when Tm * <0.
A code discriminator 111 that outputs 0 when Tm * = 0
And a coefficient unit 112 that outputs the second saturation amount component Vovr2, which is the output of the current ratio adjusting unit 21b, by multiplying it by a predetermined coefficient K2, and the output of the sign discriminator 111 and the output of the coefficient unit 112. It is composed of a multiplier 113 which forms the correction amount ΔTm * of the command, and a subtracter 114 which subtracts the correction amount ΔTm * from the torque command Tm * and outputs the corrected torque command Tm * ′.

Next, the operation of the ninth embodiment will be described. First, when the first saturation amount component Vovr1 and the current ratio FB and the current ratio FA are input to the current ratio adjusting means 21b, when voltage saturation occurs, the value of 1 before correction is obtained according to the magnitude of Vovr1. The current ratio FA of the next current command is shifted so as to approach the current ratio FB in the minimum primary voltage control, and correction calculation is performed, and the corrected current ratio FC is output. If voltage saturation occurs even after using the corrected primary current command, the degree of voltage saturation is determined by the second saturation amount component Vo.
Output as vr2. Vovr2 is specifically generated as follows. When the output K1 · Vovr1 of the coefficient unit 101 in FIG. 21 is input to the saturation amount calculation circuit 106, when K1 · Vovr1 ≦ 1.0, Vovr2
= 0.0, V1 if K1 · Vovr1> 1.0
r2 = K1 · Vovr1-1.0 is output respectively.
Vovr2 is a component that represents the degree of voltage saturation when voltage saturation still occurs even if the current ratio FC is adjusted to match the current ratio FB during the minimum primary voltage control.

Subsequently, when the torque command Tm * and the second saturation amount component Vovr2 are input to the torque command correction means 22, a corrected torque command Tm * 'corrected according to the magnitude of Vovr2 is output. It Hereinafter, a specific operation of the torque command correction means 22 will be described.

The second saturation component Vovr2 is calculated by the coefficient unit 11
2 is multiplied by a predetermined coefficient K2, and the torque command Tm
Output of sign discriminator 111 for discriminating the sign of * and multiplier 1
The correction amount ΔT of the torque command is multiplied by 13
m * is generated. Next, the subtracter 114 outputs the torque command T
The torque command correction amount ΔTm * is subtracted from m *, and the corrected torque command Tm * ′ is output.

Next, when the corrected torque command Tm * 'and the current ratio FC are input to the corrected current component command calculating means 3a, the corrected primary current d-axis component command Ids *' and the corrected 1
The next current q-axis component command Iqs * 'is output. Then, based on a known vector control method, the d-axis component Ids and the q-axis component Iqs of the primary current are corrected primary current d-axis component command Ids * ′ and corrected primary current q-axis component command, respectively. It is controlled to follow Iqs * '.

According to the above procedure, in the eighth embodiment, when the voltage saturation occurs, the d-axis component and the q-axis component of the primary current command are changed to the d-axis component and the q-axis component of the primary current command in the minimum primary voltage control. When voltage saturation still occurs even if the voltage components are matched with the axial components, it is possible to prevent the occurrence of voltage saturation by reducing the torque command.

It goes without saying that the configurations described in the embodiments relating to the first invention and the second invention can be used in combination with each other. Further, the hardware configuration in the above embodiment may be realized by software processing using a microcomputer.

[0226]

As described above, according to the first aspect of the present invention, the iron loss cannot be ignored by operating the primary current of the induction motor under the condition that the loss including the iron loss is minimized. Even when the induction motor is driven, it is possible to obtain a control device capable of operating the induction motor with high efficiency. Furthermore, considering the iron loss also in the slip frequency, detecting the stator winding temperature of the motor and operating the primary current in consideration of the temperature fluctuation, and considering the fluctuation of the exciting inductance such as magnetic flux saturation. By operating the secondary current, a control device for an induction motor that prevents a decrease in torque control performance can be obtained. According to the second aspect of the invention, even if voltage saturation occurs during operation of the induction motor, by operating the primary current of the induction motor as the first step, while maintaining the output torque as instructed, If the induction motor is operated to eliminate the voltage saturation and the voltage saturation is not eliminated even in the first stage, the induction motor can be operated to eliminate the voltage saturation by reducing the torque command in the second stage. . Therefore, the output voltage and the output current are not distorted due to the occurrence of the voltage saturation, and it is possible to prevent the vibration of the induction motor and the deterioration of the torque control accuracy due to them.

[Brief description of drawings]

FIG. 1 is a block diagram showing an entire first embodiment of the present invention.

FIG. 2 is a block diagram showing a configuration of a current ratio storage arithmetic circuit according to the first embodiment of the present invention.

FIG. 3 is a block diagram showing a configuration of a current component command calculation means according to the first embodiment of the present invention.

FIG. 4 is a block diagram showing a configuration of a slip frequency calculation circuit according to the first embodiment of the present invention.

FIG. 5 is a block diagram showing an entire second embodiment of the present invention.

FIG. 6 is a block diagram showing a configuration of slip frequency iron loss component correction means according to a second embodiment of the present invention.

FIG. 7 is an explanatory diagram of the operating principle of the second embodiment of the present invention.

FIG. 8 is a block diagram showing an entire third embodiment of the present invention.

FIG. 9 is a block diagram showing a configuration of a current ratio storage arithmetic circuit according to a third embodiment of the present invention.

FIG. 10 is a block diagram showing an entire embodiment 4 of the present invention.

FIG. 11 is a block diagram showing a configuration of an inductance correction unit according to a fourth embodiment of the present invention.

FIG. 12 is a block diagram showing a configuration of a current component command calculation means according to a fourth embodiment of the present invention.

FIG. 13 is an explanatory diagram of the operating principle of the fifth embodiment of the present invention.

FIG. 14 is an explanatory diagram of the operating principle of the eighth embodiment of the present invention.

FIG. 15 is an explanatory diagram of the operating principle of the eighth embodiment of the present invention.

FIG. 16 is a block diagram showing an entire embodiment 8 of the present invention.

FIG. 17 is a block diagram showing a configuration of a current ratio calculating means (A) according to an eighth embodiment of the present invention.

FIG. 18 is a block diagram showing a configuration of voltage saturation detection means according to an eighth embodiment of the present invention.

FIG. 19 is a block diagram showing a configuration of a current ratio adjusting means according to an eighth embodiment of the present invention.

FIG. 20 is a block diagram showing an entire embodiment 9 of the present invention.

FIG. 21 is a block diagram showing a configuration of a current ratio adjusting means according to a ninth embodiment of the present invention.

FIG. 22 is a block diagram showing a configuration of torque command correction means according to a ninth embodiment of the present invention.

FIG. 23 is a block diagram showing an entire control device for an induction motor based on a first conventional technique.

FIG. 24 is a block diagram showing an entire control device for an induction motor based on a second conventional technique.

FIG. 25 is a block diagram showing a configuration of a field weakening calculator of an induction motor controller according to a second conventional technique.

[Explanation of symbols]

1 induction motor, 2a, 2b current ratio storage arithmetic circuit, 3
a, 3b Current component command calculation means, 4 Current detector, 5
Current component control circuit, 6 voltage command calculation circuit, 7 speed (rotation frequency) detector, 8 slip frequency calculation circuit,
9, 93, 128, 154, 155 Adder, 10 P
WM inverter, 11 integrator, 12 current component calculating means, 13 slip frequency iron loss component correcting means, 14 electric motor temperature detector, 15 inductance correcting means, 16,
143 DC voltage detecting section, 17 maximum output voltage calculating circuit, 18 voltage saturation detecting means, 19 current ratio calculating means (B), 20 current ratio calculating means (A), 21a, 21b
Current ratio adjusting means, 22 torque command correcting means, 23,
32, 51, 61, 76, 82 absolute value circuit, 24a,
24b, 62 map data section, 25 DC power supply, 3
1, 41, 43, 73, 101, 112, 127 Coefficient unit, 33, 44, 72, 75, 77, 81, 126 divider, 34, 78, 94 Square root arithmetic circuit, 35, 111
Code discriminator, 36, 37, 53, 74, 91, 92,
104, 113, 123 Multiplier, 42 First-order delay circuit, 52 First-order function calculator, 70 Excitation current command calculation circuit, 71, 96, 102 Limiting circuit, 95, 103, 1
05, 114, 121, 130-132, 146, 15
0, 151, 163 subtractor, 106 saturation amount calculation circuit, 122, 133 to 135 amplifier, 124 function generator, 125 compensation circuit, 129 vector calculation circuit, 1
41 three-phase power supply, 142 converter, 144 field weakening calculation unit, 145 limiter processing unit, 147 induction machine model, 148 switching unit, 149 coordinate conversion unit (A), 1
52,153 ACR amplifier, 156 coordinate conversion unit (B), 157 PWM calculation unit, 158 base circuit,
161 limit output calculation unit, 162 saturation voltage command generation unit, 164 PI calculation unit.

Continuation of the front page (51) Int.Cl. ⁶ Identification code Internal reference number FI Technical display area H02P 5/408 H

Claims (9)

[Claims] 1. An induction motor, a speed (rotation frequency) detector for detecting a rotation frequency of the induction motor, and calculation of a predetermined function having a constant value of the induction motor including a constant related to iron loss as a coefficient value. A current ratio storage arithmetic circuit that stores the result as data and outputs the ratio of the amplitude of the torque current command and the excitation current command (called current ratio) by inputting the variables related to the above-mentioned predetermined function, and the torque of the induction motor. Input the command and the above current ratio, and rotate the exciting current command at the primary frequency in the orthogonal rotation coordinate axis (d-
(referred to as q-axis) as a d-axis component command of the primary current on the above, and a torque current command of the induction motor as a q-axis component command of the primary current, and a current component command calculation means, and the induction motor of the induction motor. A current detector that detects a primary current, an integrator that integrates a primary frequency and outputs a phase, an output of the current detector and the phase are input, and
A current component calculating means for calculating a d-axis component and a q-axis component of the primary current, at least one of the d-axis component command and the d-axis component of the primary current, the q-axis component command and the q-axis component of the primary current The slip frequency of the induction motor, and the slip frequency output from the slip frequency calculating means and the output of the rotation frequency detector are added to each other to obtain the above-mentioned 1
The primary current of the induction motor is adjusted so that the d-axis component and the q-axis component of the primary current follow the d-axis component command and the q-axis component command of the primary current, respectively. And a current component control circuit for controlling, wherein the product value of the output of the current component command computing means is proportional to the torque command, and the amplitude ratio includes a constant of the induction motor including a constant related to iron loss. A control device for an induction motor, which outputs a primary current command that is equal to a predetermined function value. 2. An induction motor, a speed (rotation frequency) detector for detecting a rotation frequency of the induction motor, and calculation of a predetermined function having a constant value of the induction motor including a constant related to iron loss as a coefficient value. A current ratio storage arithmetic circuit that stores the result as data and outputs the ratio of the amplitude of the torque current command and the excitation current command (called current ratio) by inputting the variables related to the above-mentioned predetermined function, and the torque of the induction motor. Input the command and the above current ratio, and rotate the exciting current command at the primary frequency in the orthogonal rotation coordinate axis (d-
(referred to as q-axis) as a d-axis component command of the primary current on the above, and a torque current command of the induction motor as a q-axis component command of the primary current, and current component command calculation means, and the induction motor of the induction motor. A current detector that detects a primary current, an integrator that integrates a primary frequency and outputs a phase, an output of the current detector and the phase are input, and
A current component calculating means for calculating a d-axis component and a q-axis component of the primary current, at least one of the d-axis component command and the d-axis component of the primary current, the q-axis component command and the q-axis component of the primary current And at least one of the slip frequency of the induction motor, a slip frequency calculating means for calculating the slip frequency of the induction motor, the slip frequency output from the slip frequency calculating means (referred to as a slip frequency before correction) and the rotation frequency. However, the slip frequency obtained from a predetermined function calculation result having a constant value of the induction motor including a constant related to iron loss as a coefficient value, and a predetermined value having a constant value of the induction motor not related to iron loss as a coefficient value. A slip that outputs the slip frequency after correction by interpolating the slip frequency before correction by a predetermined linear function obtained by approximating from the slip frequency ratio obtained from the function calculation result A wave number iron loss component correcting means, an adder that adds the corrected slip frequency output from the slip frequency iron loss component correcting means and the output of the rotation frequency detector to form the primary frequency, and And a current component control circuit for controlling the primary current of the induction motor so that the d-axis component and the q-axis component of the next current follow the d-axis component command and the q-axis component command of the primary current, respectively. The value of the product of the output of the current component command calculating means is proportional to the torque command, and the amplitude ratio is equal to a predetermined function value using the constant of the induction motor including the constant related to iron loss. A control device for an induction motor, which outputs a next current command. 3. The induction motor controller according to claim 1, further comprising a motor temperature detector for detecting a temperature of a stator winding of the induction motor, the constant relating to the iron loss. A predetermined function having a constant value of the induction motor including a coefficient is a function of at least the electric motor temperature, and at least one of the inputs of the current ratio storage arithmetic circuit is the electric motor temperature. Control device. 4. The control device for an induction motor according to claim 1, wherein d of the primary current on an orthogonal rotation coordinate axis (called dq axis) that rotates at a primary frequency. Enter the axis component and rotation frequency,
Further, the current component command calculating means is further provided with an inductance correcting means for calculating and outputting the exciting inductance from the correspondence data of the exciting current, the rotation frequency and the exciting inductance which are obtained in advance, and the current component command calculating means is provided with the torque command and the current of the induction motor. An exciting current command calculating circuit for inputting the current ratio output by the ratio memory calculating circuit and the exciting inductance output by the inductance correcting means and outputting an exciting current command as a d-axis component command of the primary current, and the above-mentioned primary current A limiting circuit that limits the maximum value of the d-axis component command to a limit value at which the magnetic flux saturates, and a component proportional to the torque command of the induction motor is divided by the output of the limiting circuit to obtain a primary current that is a torque current command. A control device for an induction motor, comprising: a q-axis component command divider. 5. The control device for an induction motor according to claim 1, 2, or 4, wherein the current ratio data stored in the current ratio storage arithmetic circuit is at least the rotation frequency. A control device for an induction motor, which is a linear function of 6. The control device for an induction motor according to claim 1, 2, or 4, wherein the current ratio data stored in the current ratio storage arithmetic circuit is at least the primary order. A control device for an induction motor, which is a linear function of frequency. 7. The control device for an induction motor according to claim 2, wherein the predetermined linear function of the slip frequency iron loss component correction means is a function of the primary frequency. apparatus. 8. A voltage type inverter is used to separate a torque current and an exciting current into a q-axis component and a d-axis component of a primary current on an orthogonal rotation coordinate axis (referred to as dq axes) that rotates at a primary frequency. In a device for controlling an induction motor by a vector control method for controlling each of them, a DC voltage detection unit for detecting a power supply side DC voltage of the voltage type inverter, and a q-axis of a primary current which is a torque current command of the induction motor. A component command (called the uncorrected primary current q-axis component command) and a d-axis component command of the primary current that is an exciting current command (called the uncorrected primary current d-axis component command) are input, and the amplitude ratio A current ratio calculating means (A) for outputting (current ratio FA), a maximum output voltage calculating circuit for calculating the maximum primary voltage value that can be output by inputting the power supply side DC voltage, and a three-phase coordinate axis Or on dq axis Voltage saturation detection means for inputting the next voltage command and the maximum primary voltage value that can be output, detecting whether or not voltage saturation has occurred, and expressing the degree of voltage saturation as a first saturation amount component and outputting the voltage saturation, The ratio of the amplitude of the q-axis component command and the d-axis component command of the above-mentioned primary current that minimizes the primary voltage required to output the torque according to the torque command by inputting the rotation frequency and other variables of the electric motor (current ratio FB), and a current ratio calculation means (B) for calculating and outputting from the calculation or a map data of the calculation result, and the first saturation amount component, the current ratio FA and the current ratio FB.
Is input, and q of the corrected primary current is adjusted according to the degree of saturation.
A current ratio that outputs the amplitude ratio (called current ratio FC) of the axis component command (called the corrected primary current q-axis component command) and the d-axis component command (called the corrected primary current d-axis component command) Adjusting means, corrected current component command calculating means for inputting the torque command of the induction motor and the current ratio FC, and outputting the corrected primary current q-axis component command and the corrected primary current d-axis component command An induction motor control device comprising: 9. The control device for an induction motor according to claim 8, wherein the current ratio adjusting means detects the voltage saturation even when the current ratio FC is adjusted, and detects the voltage saturation to determine the saturation level. A second saturation amount component corresponding thereto is also output, and the second saturation amount component and the torque command are input, and a torque command correction means for correcting the torque command is further provided. Electric motor controller.