JPH1118433A - Pulse width modulation power converter - Google Patents

Abstract

(57) [Abstract] [Problem] Without controlling AC current and DC voltage,
Provided is a PWM power conversion device capable of maintaining a continuous operation at the time of power supply fluctuation or load fluctuation, and sufficiently obtaining high efficiency by reducing loss due to a switching element (circuit). SOLUTION: Switching elements (circuits) 2a to 2c of a converter unit 2 are detected by an AC voltage detected by a voltage detection circuit 14.
2f, a switching element of an arm connected to the phase of the three-phase AC power supply in which the absolute value of the phase voltage is minimized
(Circuit) performs pulse width modulation control, and for the arms connected to the remaining two phases,
(Circuit) Controlled so that the current flows through 2g to 2l, and the switching element (circuit) flows during the reverse conversion operation.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a power converter for converting AC power into DC power by pulse width modulation of a switching circuit, and more particularly to a power converter suitable as a converter in an inverter device.

[0002]

2. Description of the Related Art A typical example of a converter for converting AC power into DC power is a known diode bridge circuit. However, this diode bridge circuit has a problem that the power supply current includes many harmonics.
Since there is no reverse conversion function, there is a problem that power regeneration cannot be performed.

On the other hand, as a converter circuit which can cope with the problem of the diode bridge circuit, for example, a PWM circuit
There is a (pulse width modulation) power converter. However, this PW
The M power conversion device requires a control device that is not required in the diode bridge circuit, and also has a problem in that a switching loss occurs in the diode bridge circuit due to a switching element that does not exist in principle.

Therefore, as one method for reducing the problems of such a PWM power converter, a simple control method in which instantaneous current value control is omitted is disclosed in, for example, the following document. 1990, IEEJ Transactions on Journal D, 110, 7
No., "Power factor control type three-phase voltage type PWM control power converter"

[0005] The prior art disclosed in this document will be described below. This prior art relates to an inverter device for converting DC power converted by a converter device into AC power of a variable frequency, and FIG. The circuit configuration is shown, 1 is a three-phase AC power supply, 2 is a converter unit for converting three-phase AC to DC, 3 is a capacitor for smoothing a DC voltage, and 4 is an inverter unit.

Here, the converter section 2 is composed of self-extinguishing type switching elements (or switching circuits) 2a to 2f connected in Grenz and diodes 2g to 2l connected in anti-parallel to these switching elements. Have been. Similarly, the inverter unit 4 also includes self-extinguishing type switching elements (or switching circuits) 4a to 4f and diodes 4g to 4l connected in anti-parallel to the respective switching elements. The output of the inverter 4 on the AC side is connected to an induction motor 5.

[0007] Next, reference numeral 6 denotes a reactor, which functions to suppress a PWM frequency component in harmonics included in the AC current, and 7 denotes an AC voltage / current detector, which includes a current transformer 7a and an instrument transformer. The detector 7b functions to detect an AC voltage and an AC current, respectively, 8 is a DC voltage detector, 9 is a power factor detector, and based on the AC voltage and the AC current detected by the AC voltage / current detector 7, It functions to detect the power factor.

Reference numeral 10 denotes a PLL control circuit which outputs a phase command α for keeping the power factor constant. Reference numeral 11 denotes a DC voltage control circuit which converts the DC voltage detected by the DC voltage detector 8 to a set value. A sine wave PWM control signal generator 12 outputs a modulation degree command M for adjustment, and is a firing (ON) / extinguishing (OFF) command 12 for the switching elements 2a to 2f.
a is a sine wave PWM control signal generator on the inverter side, which outputs switching elements 4a to 4f.
It outputs the firing (ON) / extinguishing (OFF) commands 13a to 13f of f.

FIG. 23 shows a sine wave PWM control signal generator 1
2 is a diagram for explaining the operation of FIG. 2, in which a modulated wave having an amplitude modulation rate M is synthesized at a phase advanced by α from the AC voltage signal ER,
The ignition / extinguishing commands 12a and 12d of the switching elements 2a and 2d are output according to the magnitude of the triangular waveform carrier.

FIG. 24 shows an AC power supply voltage ER, a current IR,
A one-phase vector diagram showing the relationship between the AC reactor voltage drop EX and the converter voltage EC. If the modulation factor M is equal to or greater than the critical modulation factor MC determined by the load state, the power factor is always adjusted by adjusting the phase command α. It is said that control to set to 1 is possible. When the load fluctuation on the inverter side is small, the modulation factor M may be fixed to a value close to 1, and in this case, the DC voltage control circuit 11 for adjusting the modulation factor M is not necessarily required. It has been.

Therefore, according to the converter section 2 of the inverter device shown in FIG. 22, the harmonic current can be suppressed as compared with the diode bridge circuit, and the AC current control required in the ordinary PWM power converter is omitted. can do.

[0012]

In the above prior art, it cannot be said that sufficient consideration is given to maintaining operation stability and improving efficiency, and as described below, when the operation state fluctuates. There is a problem in continuity of operation and reduction in loss in the switching element or the switching circuit. According to the above prior art, higher harmonic currents can be suppressed than in the diode bridge circuit, and the AC current control required in the ordinary PWM power converter can be omitted.

However, when it is necessary to continue the operation even when the AC power supply voltage drops or becomes unbalanced, the inductance value of the AC reactor 6 is increased to suppress the overcurrent, or the converter unit 2 is constructed. There is a problem that it is necessary to increase the capacity of the elements and components that are used so as to sufficiently withstand overcurrent.

Further, in the above-mentioned prior art, the DC voltage control circuit can be omitted. In this case, even though the load fluctuation on the inverter side is normally small, the DC voltage control circuit is transiently directly connected to, for example, an AC motor having an inverter load. For example, when the load torque of the used machine suddenly increases, there is a problem that the DC voltage drop becomes large and the operation cannot be continued.

An object of the present invention is to maintain continuous operation in the event of power supply fluctuations and load fluctuations without controlling the AC current and DC voltage, and to improve the efficiency by reducing losses due to switching elements or switching circuits. An object of the present invention is to provide a PWM power conversion device that can be obtained sufficiently.

[0016]

An object of the present invention is to provide a pulse width having a self-extinguishing type switching circuit and a diode circuit connected in an anti-parallel manner between a three-phase AC power supply and a DC smoothing circuit having a capacitor. In the modulated power converter, for the arm connected to the phase of the three-phase AC power supply having the smallest phase voltage amplitude, the switching circuit performs pulse width modulation control, and the arm connected to the remaining two phases. Is achieved by controlling the diode circuit to flow during the forward conversion operation and controlling the switching circuit to flow during the reverse conversion operation. Further, at this time, it is also possible to detect the abnormality on the AC power supply side and prevent the ignition command to all the switching circuits, and release the inhibition after the abnormality detection is completed.

Next, the reason why the object of the present invention is achieved by the above means will be described below in comparison with a well-known diode bridge converter circuit shown in FIG. First, in a converter circuit using a diode bridge as shown in FIG. 25, which is also well-known, as shown in FIG. 26, the waveform of the AC power supply current IR becomes a distorted waveform that is largely separated from the sine waveform. As a result, a large harmonic current is included, which causes various troubles.

Therefore, as a countermeasure, there is a method of suppressing the harmonic current by increasing the inductance value of the AC reactor. In this case, the operation of the diode bridge converter circuit is expressed by the following equation (1): ,

[0019]

(Equation 1)

When the coordinates are converted into αβ axes and displayed on a plane, the locus of the alternating current corresponding to the current waveform in FIG. 26 is as shown in FIG. 27, and the locus of the converter voltage is shown in FIG.
As shown in FIG. In the equation (1), f represents a general variable.

The current voltages at points A to I in FIG. 26 correspond to FIG. 28 and points A to I in FIG. 28, respectively. At this time, the AC power supply voltage is rotating at a constant angular velocity on the circumference of a constant radius. Therefore, the difference between the AC power supply voltage and the converter voltage locus is an AC reactor voltage, and both the locus of the AC current and the locus of the voltage are It is symmetrical at 60 ° cycle. In the following description, the R-phase current waveform shown in FIG. 26 will be described. However, the other S-phase and T-phase are symmetrical and the same.

First, the current locus in FIG.
As shown in FIG.
Waveform 6 corresponds to a period in which current flows in two of the three phases. Next, on the converter voltage locus of FIG.
During the two-phase commutation period, the rotational angular velocity is small, and the commutation period A
The rotational angular velocity is large between B and GH. When the inductance of the AC reactor is increased, the AC current is suppressed because the commutation period becomes longer and the angular velocity decreases, and as a result, the rotation of the voltage locus approaches the constant angular velocity. Even if the voltage locus remains hexagonal, the harmonic current can be further reduced if the voltage locus can be rotated at a constant angular velocity.

Here, the means of "PWM control of the switching element of the arm connected to the phase having the smallest absolute value among the three-phase AC power supplies in the three-phase AC power supply" means that the voltage locus is changed at a constant angular velocity. Acts to rotate.

On the other hand, in the converter circuit using the diode bridge, the DC voltage is automatically adjusted to be substantially constant even when the load on the DC side is suddenly changed, even though no control is performed. However, it has the advantage of stable operation. This is because the hexagonal size of the converter voltage trajectory shown in FIG. 28 is proportional to the DC voltage of the smoothing capacitor, so that even if the DC load fluctuates, it is automatically adjusted accordingly. This is because the modulation factor is not suppressed to 1 or less unlike the converter, so that the fluctuation rate of the DC voltage is suppressed to a small value.

Therefore, according to the present invention, two phases having a large absolute value among the three-phase AC power supplies are continuously fired,
The magnitude of the voltage locus is proportional to the DC voltage, and DC voltage characteristics close to those of a diode bridge converter circuit can be obtained.

As a result, even when the firing command to the self-extinguishing type switching element is completely blocked and the operation shifts to an operation equivalent to that of the diode bridge converter circuit, or when the firing command is restarted, the fluctuation of the capacitor voltage does not occur. The operation on the DC load side can be continued even when the AC power supply is abnormal.

Further, according to the present invention, since the two-phase switching element having a large absolute value among the three-phase AC power supplies is continuously fired, the switching operation frequency is reduced, and the switching loss is accordingly reduced. Decrease and increase efficiency.

[0028]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS Hereinafter, a pulse width modulation power converter according to the present invention will be described in detail with reference to the illustrated embodiments. First, the operation mode of the switching element in each embodiment will be described after being defined by an equivalent circuit shown in FIG. In FIG. 29, three switches SR, SS, and ST correspond to R, S, and T phases, respectively.
The +1 indicates the conduction of the upper arm, and the -1 indicates the conduction of the lower arm. Then, there are eight types of operation modes of these switches as shown in the following (Table 1).

[0029]

[Table 1]

Here, assuming that the DC voltage VDC is constant, the voltage generated by the converter in each mode is
FIG. 30 shows the coordinates converted to αβ axes by the equation (1). Therefore, in FIG. 30, only the switch SS has a different switch state between the point A in mode 1 and the point B in mode 2, and the switch ST remains at +1 while the switch ST
Remains at the -1 side, and the switch state does not change in any case.

Therefore, now, the switch SR and the switch S
If T is fixed and only the switch SS is PWM-modulated, and if the ratio of the time connected to the -1 side is γ and the ratio of the time connected to the +1 side is (1−γ), the converter The trajectory of the voltage is, as shown in FIG.
= Γ. Therefore, when γ is continuously changed from 1 to 0, the locus moves from the point A to the point B. Further, there is a relationship shown in the following equation (2) between the phases θ and γ of the trajectory, and a condition for the phase θ to have a constant angular velocity is obtained from the relationship.

[0032]

(Equation 2)

The above description relates to the movement between the points AB. Similarly, when the conditions for rotating the voltage locus at a constant angular velocity are obtained, the following equation (3) is obtained.
The switch subjected to M modulation is as shown in FIG.

[0034]

(Equation 3)

For example, in FIG. 31, the phase is θ ₁
In this case, the switch is switched by the switch SR. The conduction ratio on the -1 side is the length AB, and the conduction ratio on the +1 side is the length BC.
It turns out that it is. Therefore, if one of the three switches is PWM-controlled in sequence according to FIG. 31, the converter voltage (R phase) at this time is as shown in FIG. Note that FIG. 32 shows the average current value excluding the pulsation component due to the PWM control.

As a result, of the three-phase AC power supply, the switching element of the arm connected to the phase having the smallest absolute value is subjected to PWM control, and the forward conversion operation is performed for the remaining arm connected to the two phases. Sometimes flowing through the diode,
At the time of the inverse conversion operation, the switching element is controlled so as to flow continuously. In the case of the voltage waveform shown in FIG.
%, And the 11th and 13th harmonics are about 0.8% each.
This is based on the point that the locus of the converter voltage is controlled to be hexagonal as shown in FIG. 30 in the first embodiment of the present invention. Called the mold embodiment.

As described above, the magnitude of the harmonic current is determined by the inductance value of the AC reactor.
In the case of the hexagonal basic type embodiment, as described above, it can be seen that even with the same inductance value, the harmonic current can be suppressed as compared with the diode converter circuit.

Next, FIG. 33 shows a case where the locus of the converter voltage is made to be a dodecagon so that the fifth and seventh harmonic components are theoretically eliminated. The embodiment will be referred to as a dodecagon basic embodiment.
In FIG. 33, point E and point F are angle AOE,
At the position where the angle AOF is 15 °, the point C is located on the line segment AB where the angle AOC has the phase θ. And C in FIG.
The point is the same as point C in FIG.
The conduction ratio γ ₁ of the time connected to the +1 side of S is given by the following (4)
By making the value determined by the equation, the control shown in FIG. 33 can be realized.

[0039]

(Equation 4)

Here, in order to make the voltage locus on a dodecagon, it is necessary to further shift the voltage vector to point D. For this purpose, the conduction period from mode 7 to mode 8 shown in Table 1 is used. Should be provided. In this case, on the trajectory from point A to point E, the period of mode 1 is longer than the period of mode 2, so the voltage vector at point O changes the switch SR from mode A to -1 and switches to mode 7 Here, if the ratio of the period of the mode 7 is γ ₂ , OD / OC
= 1−γ ₂ .

On the other hand, the line segment OD and the line segment OC have the following (5)
There is an expression relationship.

[0042]

(Equation 5)

Therefore, the phase θ and γ ₂ may be set to have a relationship represented by the following equation (6).

[0044]

(Equation 6)

As a result, the point of the voltage vector D is obtained by setting the ratio of the mode 1 period to (1−γ ₂ ) × γ ₁ and the ratio of the mode 2 to
(1−γ ₁ ) × (1−γ ₂ ), and the ratio of mode 7 is γ ₂
Can be realized. By the way, the above is the case where the point C is on the line segment GE. When the point C is on the line segment EH, γ ₂ = 0.
What is necessary is just to control a flow rate similarly to the case of a square.

Next, the case of movement between the line segments AH has been described above. If conditions for rotating the voltage locus at a constant angular velocity are similarly obtained, the result is as shown in FIG. In FIG. 32, for example, when the phase is θ ₂ , the switches are switched by the switch SS and the switch SR, and the conduction ratio on the −1 side of the switch SS is a line segment AB, and the conduction ratio on the +1 side is a line segment B
D, the conduction ratio on the -1 side of the switch SR is a line segment AC, and the +1 side is a line segment CD.

When the three switches are PWM-controlled according to FIG. 34, the arm of the three-phase AC power supply connected to the phase having the smallest absolute value is subjected to PWM control of its switching element, and the remaining elements are PWM-controlled. The arm connected in two phases is controlled so that the current flows through the diode during the forward conversion operation and the switching element continuously flows during the reverse conversion operation. At this time, the converter voltage (R phase) is: As shown in FIG.

In the case of the voltage waveform shown in FIG. 35, the fifth harmonic and the seventh harmonic are theoretically eliminated from the fundamental wave, and the 11th 1st harmonic is eliminated.
The tertiary harmonics are about 0.8% each. Therefore, it can be seen that according to the embodiment of the dodecagonal basic type, the harmonic current can be further suppressed as compared with the case of the hexagonal basic type of FIG. .

Next, specific examples of each embodiment of the present invention will be described. First, FIG. 1 shows a specific example of the above-described hexagonal basic type embodiment. Here, the same elements as those in the prior art shown in FIG. 22 are denoted by the same reference numerals, and detailed description thereof will be omitted. I do. In this embodiment, the inverter unit 4 is shown as an example of the load on the DC side.
Embodiments of the present invention are not limited to inverters, and can be implemented with any DC load.

Reference numeral 14 denotes a voltage detecting device, which is a photocoupler 14a to 14f connected by a Grenz connection, and a resistor 14 for suppressing current.
g, and the voltages ER, E of each phase of the three-phase AC power supply 1
It functions to detect the maximum voltage detection signals 15a to 15f from S and ET. Here, the maximum voltage detection signals 15a to 15
f is level 1 when the photocouplers 14a to 14f are conductive, and level 0 when the photocouplers 14a to 14f are open. Reference numeral 16 denotes a minimum voltage detection circuit, which is a combination of AND circuits as shown in the figure, and serves to output minimum voltage detection signals 17a to 17f.

Reference numeral 18 denotes an AC voltage control circuit which functions to output PWM firing commands 19a to 19f, details of which will be described later. 20 is a driver circuit, which is a PWM firing command 19a-
A firing command 21a for each of the switching elements 2a to 2f of the converter unit 2 is formed by an OR circuit and an amplifier that receive the input of 19f and the maximum voltage detection signals 15a to 15f.
-21f.

Here, as an embodiment of the present invention, a self-extinguishing type switching element (circuit) 2a to 2f and a diode
2g to 2l are shown as examples in which independent elements or circuits are connected in anti-parallel, but there are various other examples, for example, a self-extinguishing type switching circuit and an anti-parallel diode. The circuit was provided on the same wafer,
For example, a GTO thyristor with an anti-parallel diode connected in a Grenz connection has exactly the same operation. This is the same in the following embodiments.

Next, referring to FIG.
Will be described in detail. First, the minimum voltage detection signal 17
a to 17f are input to the rise detection circuits 18a to 18f to generate pulse output signals representing the rise of each pulse, and supply them to the OR circuit 18g. Next, the output of the OR circuit 18a is input to the counter 18h, and the phase signal θ is output.
j.

FIG. 3 shows the operation of the counter 18h. As shown in the drawing, the counter 18h is reset every time an input pulse appears, restarts at the same time, starts counting up, and outputs the phase signal θ. Occurs. At this time, as described above, in this embodiment, θmax
Is set to correspond to the phase of 60 °.
The output of 8h is limited so as not to exceed the set value θmax.

Therefore, these function generators 18i, 18
h outputs a conduction ratio in accordance with the phase signal θ. At this time, as shown in the figure, in the function generator 18i, the conduction ratio increases from 0 in accordance with the phase signal θ, and the input phase 60゜, a duty ratio having a characteristic of reaching and saturating to 1 is generated. In the function generator 18j, when the phase signal θ is 0, the duty ratio is 1
Then, a conduction ratio having a characteristic of decreasing from that and saturating to 0 at a phase corresponding to 60 ° is generated.

Therefore, these two function generators 18
The sum of the outputs of i and 18j is almost always 1 so that the conduction ratio approaches the value of the equation (2), and the setting is made by the broken line approximation in consideration of the idle period necessary for the switching element. Will be.

On the other hand, there are provided two carrier generators 18m and 18n which operate in synchronization with the output of the clock generator 18k, from which carrier signals having a triangular waveform are generated. Generators 18i, 1
The output of 8j is input to comparators 18p and 18q.

Therefore, these comparators 18p and 18q
Are the firing commands 19g, 19h
And these are input to an analog switch 18r that is opened and closed by the minimum voltage detection signals 17a to 17f.
FIG. 4 shows the operation of the comparators 18p and 18q. As shown in FIG. 4, the firing command is output by comparing the conduction ratio command with the carrier wave.

The analog switch 18r is composed of a plurality of switching contact type switch elements,
When 7a to 17f are at level 1, they are switched as shown to output firing command signals 19g and 19h, and when they are at level 0, they are switched in the opposite manner as shown to output a 0 signal (common potential signal). As a result, the output from the OR circuit 18s that receives the output signal of the analog switch 18r as an input is:
PWM firing command 19a to each switching element (circuit)
19f, and the control shown in FIG. 32 is obtained. Here, it is also possible to omit the broken line approximation in the function generator and to use only the limiter of the upper limit 1 and the lower limit 0. However, according to this embodiment, the effect of suppressing harmonics is greater than in such a case. There is an advantage.

Next, FIG. 5 shows another embodiment of the present invention.
In this figure, reference numeral 22 denotes a voltage phase detecting device, 23 denotes an external setting device, and the voltage phase detecting device 22 functions to output the voltage phase signal θ corrected by the external setting device 23. Numeral 24 is a function generator for calculating the duty ratio, which functions to output the duty ratio commands 25a to 25c of each phase of the RST.

Reference numerals 26a to 26d denote comparators for comparing the magnitude of the carrier signal from the carrier generator 27 with the duty ratio command.
Each of the comparators 26a to 26d outputs an ignition command to the switching element (circuit) connected to each phase of the RST.
9a to 29f, each of which is a switching element (circuit) 2a
To 2f.

The other structure is the same as that of the embodiment shown in FIG.

Reference numeral 30 denotes an AC ammeter which is used when adjusting the phase correction value α by the external setting device 23. At this time, the phase correction value α is set so that the current amplitude at the rated load of the converter is minimum and the power Set so that the rate becomes 1.

FIG. 6 shows an example of the configuration of the voltage phase detector 22. In FIG. 6, first, a low-pass filter (low-frequency filter) 22a is provided for the purpose of removing harmonic components of a voltage signal. In this embodiment, when the power supply frequency signal is input, the output signal of the low-pass filter 22a is set to be delayed by about 30 °, so that the output signal has substantially the same phase as the ER phase voltage. Reference numeral 22b denotes a comparator which receives an output signal of the low-pass filter 22a and outputs a signal having a level 1 when the signal is positive and a level 0 when the signal is negative.

Reference numeral 22c denotes a rising detection pulse generator.
22d is a pulse shift circuit, and in this embodiment,
The output pulse of the rising detection pulse generator 22c is 3
Assuming that the phase sequence of the phase AC power supply 1 is ER → ES → ET, the three phases are balanced and the frequency is constant, the phase when the phase voltage of the ER phase is a positive peak is θ = 0, and It is output via a pulse shift circuit 22d that shifts the pulse by the phase correction value α manually set by the setting device 23.

22e is a counter, which is a pulse shift circuit 2
The counter 22e operates by using the pulse output from 2d as an input and outputs the voltage phase signal θ. Therefore, the operation of the counter 22e is the same as that of the counter 18h in the embodiment of FIG. Now, let θmax be the phase 36
0 ° is set.

FIG. 7 shows the components of the voltage phase detecting device 22.
FIG. 8 shows a setting example of the function generator 24. In this embodiment, the characteristic of FIG. 31 is approximated by a polygonal line, and the S-phase conduction ratio command is approximated by a polygonal line passing through the points A and B. As a result, when θ = 30 °, the point C at a conduction ratio of 0.5 is obtained. Pass through.

FIG. 9 shows the operation of the comparators 24d to 24f, which outputs a duty ratio command and a firing command by comparing the magnitude of a carrier. A pause section TD for preventing an arm short circuit is set between the ignition command and the ignition command.

According to this embodiment, the harmonic component is removed by the low-pass filter 22a, so that the detection of the rising edge of the pulse is disturbed by the voltage pulsation that appears due to the influence of the power supply impedance of the three-phase AC power supply 1, and the counter is not controlled. Malfunction appears but is suppressed, so that stable control can be obtained. Further, since the impedance drop due to the AC reactor is phase-compensated and the operating power factor can be maintained at 1 near the rated output, the input current can be suppressed.

Next, FIG. 10 also shows an embodiment of the present invention.
Is a voltage phase detector using a digital arithmetic unit, and serves to output the voltage phases θR, θS, and θT of each phase. As a specific calculation method at this time, a method using a moving Fourier transform is known.
Japanese Patent Application Laid-Open No. 682 and Japanese Patent Application Laid-Open No. 1-283015 disclose voltage phase calculation methods that take into account malfunctions inside the operation device, and these methods may be used.

The phase detection signals representing the voltage phases θR, θS, and θT of the respective phases are corrected by adders 32a to 32c based on the correction value α output from the external setting unit 23. 33
Reference numerals a to 33c denote function generators for calculating the conduction ratio. The function generator 33a for the R phase has the same configuration as the function generator 24 in the embodiment of FIG. 5, that is, the R-phase conduction ratio setting function of FIG. It is the configuration used. As for the S phase and the T phase, the phase signals input to the function generators 33b and 33c are signals in which θ = 0 at a positive peak value, similarly to the R phase, so that the function generator has the same setting as the R phase. good.

The other structure is the same as that of the embodiment shown in FIG.

According to this embodiment, the conduction ratio is controlled in each phase in accordance with the individual voltage phase. Therefore, even when a reverse-phase voltage is generated due to an abnormality in the AC power supply, the converter input current is not disturbed. Can be suppressed.

Next, FIG. 11 shows a specific example of the above-described embodiment of the dodecagonal basic type. In the example of FIG. 11, FIG.
10 shows a case where the digital operation device 31 is used, as in the embodiment of FIG.
The other parts are the same except that the function generators 33a to 33c for calculating the conduction ratio are replaced with function generators 34a to 34c.

FIG. 12 shows the details of the setting characteristics in the function generators 34a to 34c, which are the same for each phase. Here, the difference between this embodiment and the setting of the R phase in FIG.
The setting is based on the same polygonal line approximation only in the section of 30 ° each centered on ゜ and 180 °, and in other sections.

In this embodiment, the value obtained by substituting θ = 0 ° into the above equation (6) is first reduced to 0.866 at point A, and linearly changed to 1 before and after. Next, at the point B, the value is increased to 0.134, and also changes linearly to 1 before and after. Therefore, according to the embodiment of FIG. 11, it is possible to suppress the fifth and seventh harmonics while obtaining the same effect as the embodiment of FIG.

In the embodiment of FIG. 11, FIG.
Similarly to the embodiment shown in FIG. 5, the case where the digital arithmetic unit 31 is used is shown.
In this case, the function setting units 34a to 34c shift and output the pattern of each phase by 120 °.

Next, FIG. 13 shows an embodiment of the present invention in which the driving power factor is automatically controlled. In the drawing, first, a voltage-current transformer 35 functions to detect an AC waveform signal. Next, a reactive power detector 36 detects the reactive power from the AC waveform signal input from the voltage / current transformer 35, and has a function of generating the detected value Qm. 37
Is a reactive power control circuit, and the reactive power detection value Qm is
It serves to output the phase correction command α so as to match ref.

The phase correction command α is supplied to the adders 32a to 32c, as in the embodiment of FIG. 10, so that the phase signal θ of each phase is corrected. , And FIG.

FIG. 14 shows an example of the reactive power control circuit 37.
The difference between the command value Qref and the reactive power detection value Qm is
And a limiter circuit 40, and outputs a phase correction command α via a limiter circuit 40. Here, usually
The operation is performed with the command value Qref set to 0, and the power factor is set to 1.
Therefore, according to the embodiment of FIG. 13, control is performed such that the operating power factor always becomes the command value Qref. As a result, when the DC load 4 changes suddenly, Even in the case of regenerative operation, the driving power factor can be kept at 1.

FIG. 15 shows a specific example of a control circuit of the inverter unit 4 in the embodiment of FIG. 13. The configuration of the inverter unit 4 is the same as that of the prior art shown in FIG. It is. A current transformer 41 is provided on the AC side of the inverter unit 4, and a current signal of the AC motor 5 measured by the current transformer 41 is supplied to a current control circuit 42. A speed detector 43 directly connected to the AC motor 5 is provided.
4 outputs a current command 45, which is also supplied to the current control circuit 42. As a result, the current control circuit 42 outputs the current command 4
5 so that the current signal matches 5.
γW is output to a sine wave PWM control signal generator.

In this case, since the current is controlled on the inverter section 4 side, the inverter side has a variable DC current source characteristic when viewed from the converter section 2 and the smoothing capacitor 3.
In this embodiment, however, the converter section 2 does not fluctuate its output DC voltage in response to DC current fluctuation on the inverter side, similarly to the diode bridge. Can be controlled.

Next, FIG. 16 shows an embodiment of the present invention having a voltage protection function. In the figure, reference numeral 46 denotes an AC voltage detector, which uses a movable Fourier transform like the voltage phase detector 31. It functions to output the positive-phase voltage amplitude VP and the negative-phase voltage amplitude VN of the AC power supply 1. 47 is an AC voltage protection circuit, which is a control command signal C according to the voltage amplitude signals VP and VN.
It functions to output a switching command Mg for the TR and the electromagnetic switch 49. The electromagnetic switch 49 is provided with an auxiliary contact so that a signal for confirming the switching operation can be obtained. Numeral 48 denotes an OR circuit which functions to prevent the outputs of the comparators 26a to 26d from being supplied to the amplifier 28 when the control command signal CTR is at level 0. The other configuration is the same as that of the embodiment of FIG.

FIG. 17 shows a voltage comparison characteristic in the AC voltage protection circuit 47. When the positive-phase voltage amplitude VP becomes equal to or less than the set value VP1, the low-voltage detection signal VP59 becomes level 1. In this embodiment, as shown in the figure, a hysteresis characteristic is provided, and the low-voltage detection signal VP59 does not return to level 0 unless the positive-phase voltage amplitude VP becomes larger than the set value VP2. ing. The negative-phase voltage detection signal VN59 becomes level 1 when the negative-phase voltage amplitude VN becomes larger than the set value VN1,
When the value becomes smaller than the set value VN2, the level returns to level 0.

FIG. 18 shows a specific example of the AC voltage protection circuit 47, which is composed of a relay protection interlocking circuit. Then, first, the relay V30 outputs the voltage amplitude signals VP59 and VN.
It is excited when 59 is at level 1. Also, the relay R1
Is a start command STR and a stop command ST from a manual switch.
P controls the opening / closing command Mg to the electromagnetic switch 49 and the timer T1 by the contact. Then, the electromagnetic switch 49 is closed by the relay R1, and the time during which the smoothing capacitor 3 is charged by the diode circuit of the converter unit 2 is secured by the timer T1, and the control command signal CTR becomes level 1 by the closing operation of the contact T1. Thereby, the inhibition of the ignition command to the switching element (circuit) is released, the converter operation starts, and the normal operation state is entered.

When the voltage is abnormal, the control command signal CTR becomes level 0 by the b-contact operation (opening operation of the normally closed contact) of the relay V30, and the firing command is stopped. Level 1 On the other hand, when the voltage abnormality detected by the relay V30 continues beyond the set time value, the timer T2 excites the relay V86, operates the emergency stop signal X86, and stops the operation. Here, the contact Z86 is opened and closed by another emergency stop protection device (not shown), for example, an overcurrent protection relay.

Next, the operation of the AC voltage protection circuit 47 will be described with reference to the timing chart of FIG. Now
Assuming that the start command STR is energized at time t1, the timer T1 starts operating here, and the electromagnetic switch 49 closes after the operation delay time. When the timer T1 is closed at time t2, the operation command signal CTR becomes level 1 and the ignition control of the switching element (circuit) is started. Then, assuming that the stop command STP is energized at time t3,
Here, the operation returns to the stop state. The above is the normal operation stop operation.

Next, it is assumed that the STP is re-energized at time t4, the CTR goes to level 1 at time t5, the operation starts, and then the voltage abnormality signal V30 is detected from time t6 to t7. I do. Then, at this time t6
During the period from t7 to t7, the CTR becomes level 0 and the ignition of the switching element (circuit) is stopped, but since the electromagnetic switch 49 is kept closed, the diode 2
The operation is continued by the commutation operation by a, 21 (ell).

Then, assuming that the abnormality is continued up to the set time value of the timer T2 after the voltage abnormality is detected at the time t8, the V86 is excited at the time t9, whereby the X86 is further excited and the timer T1 is reset. It is opened. As a result, the electromagnetic switch 49 is opened with a delay time, the auxiliary contact × Mg is opened, the excitation of the timer T2 is released, and an emergency stop operation is performed. During this time, the excitation of V86 is maintained, but thereafter, at time t10, RST is energized and reset.

According to this embodiment, an operation of shifting to the operation by the diode bridge before the overcurrent flows in the converter unit 2 due to the abnormality of the AC voltage can be obtained. As a result, the switching of the converter unit 2 can be performed. element
(Circuit) Capacities of 2a to 2f can be reduced.

FIG. 20 shows an embodiment of the present invention having a current protection function. In the figure, first, 50 and 51 are AC overcurrent detectors, and when the absolute values of the AC currents IR and IS exceed the set values, the output signal CR5 which becomes level 1 is provided.
1. It functions to generate CS51. Next, 51 is an AC current protection circuit, which is an overcurrent detection output signal CR51, CS5.
In accordance with 1, the control signal CTR and the switching command Mg to the electromagnetic switch 49 are output. The other configuration is the same as that of the embodiment shown in FIG.

FIG. 21 shows a specific example of the AC current protection circuit 52. The VP in the AC voltage protection circuit 47 shown in FIG.
59 and VN59 are replaced with CR51 and CS51, respectively. Here, when CR51 and CS51 are at level 1, the relay C30 is excited. When the current is abnormal, the control command signal CTR becomes level 0 due to the b-contact operation of the relay C30, and the ignition command is stopped. However, the CTR becomes level 1 again when the relay C30 returns to level 0.

On the other hand, when the abnormal current detected by C30 continues beyond the set time value, the timer T2 sets the relay C
86 is excited, the emergency stop signal X86 is operated, and the operation is stopped. Here, Z86 is controlled to be opened and closed by another emergency stop protection device not shown. Therefore, according to the embodiment shown in FIG. 20, even when the AC current is in the overcurrent state, the switching operation of the converter unit 2 is automatically stopped by detecting the overcurrent, so that the switching element (circuit) 2a To 2f can be reliably protected.

In the above embodiment, the self-extinguishing type switching elements (circuits) 2a to 2f and the diodes (diode circuits) 2g to 2l are shown as examples in which independent elements or circuits are connected in anti-parallel. However, as described above, there are various other examples. For example, a GTO thyristor with an anti-parallel diode provided with a self-extinguishing switching circuit and an anti-parallel diode circuit on the same wafer is used. Therefore, it goes without saying that the present invention may also be implemented using such a GTO thyristor with an anti-parallel diode, for example.

[0095]

According to the present invention, continuous operation can be maintained even when the power supply or the load fluctuates without controlling the AC current and the DC voltage, so that the configuration can be simplified and the pulse width with high reliability can be maintained. A power converter of a modulation method can be easily provided. Further, according to the present invention, the switching frequency of the switching element (circuit) can be reduced, so that a switching loss is reduced and a high-efficiency pulse width modulation power converter can be easily provided.

[Brief description of the drawings]

FIG. 1 shows a first embodiment of a pulse width modulation power converter according to the present invention.
FIG. 2 is a block circuit diagram showing the embodiment.

FIG. 2 is a circuit diagram illustrating a specific example of an AC voltage control circuit according to the first embodiment.

FIG. 3 is an explanatory diagram of an operation of a counter according to the first embodiment.

FIG. 4 is an explanatory diagram of an operation of the comparator according to the first embodiment.

FIG. 5 shows a second embodiment of the pulse width modulation power converter according to the present invention.
FIG. 2 is a block circuit diagram showing the embodiment.

FIG. 6 is a block diagram illustrating a specific example of a voltage phase detection device according to a second embodiment.

FIG. 7 is an explanatory diagram of an operation of the voltage phase detection device according to the second embodiment.

FIG. 8 is an explanatory diagram of setting characteristics of a function generator according to the second embodiment.

FIG. 9 is an operation explanatory diagram of the comparator according to the second embodiment.

FIG. 10 is a block circuit diagram showing a third embodiment of the pulse width modulation power converter according to the present invention.

FIG. 11 is a block circuit diagram showing a fourth embodiment of the pulse width modulation power converter according to the present invention.

FIG. 12 is an explanatory diagram of setting characteristics of a function generator according to a fourth embodiment.

FIG. 13 is a block circuit diagram showing a fifth embodiment of the pulse width modulated power converter according to the present invention.

FIG. 14 is a block diagram illustrating a specific example of a reactive power control circuit according to a fifth embodiment.

FIG. 15 is a block circuit diagram showing a pulse width modulation power converter according to a sixth embodiment of the present invention.

FIG. 16 is a block circuit diagram showing a pulse width modulation power converter according to a seventh embodiment of the present invention.

FIG. 17 is an explanatory diagram illustrating a characteristic example of the voltage comparator according to the seventh embodiment.

FIG. 18 is a block diagram illustrating a specific example of an AC voltage protection circuit according to a seventh embodiment.

FIG. 19 is an explanatory diagram of an operation of the AC voltage protection circuit according to the seventh embodiment.

FIG. 20 is a block circuit diagram showing an eighth embodiment of the pulse width modulation power converter according to the present invention.

FIG. 21 is an explanatory diagram of the operation of the AC voltage protection circuit according to the eighth embodiment.

FIG. 22 is a block circuit diagram showing an example of a conventional pulse width modulation power converter.

FIG. 23 is an operation control diagram of a sine wave pulse width modulation control signal generator according to the related art.

FIG. 24 is a vector diagram of an AC voltage / current in the related art.

FIG. 25 is a circuit diagram illustrating an example of a diode bridge circuit.

FIG. 26 is a waveform diagram of the diode bridge circuit.

FIG. 27 is a locus diagram of an alternating current of the diode bridge circuit.

FIG. 28 is a locus diagram of an AC voltage of the diode bridge circuit.

FIG. 29 is a circuit diagram that defines an operation mode of a switching element (circuit) in the pulse width modulation power converter.

FIG. 30 is a coordinate conversion diagram for explaining the operation of the pulse width modulation power conversion device according to the present invention.

FIG. 31 is a characteristic diagram showing an example of switching control in the pulse width modulated power converter according to the present invention.

FIG. 32 is a characteristic diagram illustrating an example of voltage control in the pulse width modulated power converter according to the present invention.

FIG. 33 is a coordinate conversion diagram for explaining the operation of the pulse width modulation power conversion device according to the present invention.

FIG. 34 is a characteristic diagram showing another example of the switching control by the pulse width modulation power conversion device of the present invention.

FIG. 35 is a characteristic diagram showing another example of voltage control in the pulse width modulation power converter according to the present invention.

[Explanation of symbols]

 DESCRIPTION OF SYMBOLS 1 3-phase AC power supply 2 Converter part 3 Smoothing capacitor 4 Inverter part 5 AC motor 6 AC reactor 14 Voltage detection device 16 Minimum voltage detection circuit 18 AC voltage control circuit 20 Driver circuit

 ──────────────────────────────────────────────────続 き Continuing on the front page (72) Inventor Seiji Ishida 7-1-1, Higashi-Narashino, Narashino-shi, Chiba Industrial Equipment Division, Hitachi, Ltd. (72) Inventor Hiroshi Fujii 7-1-1, Higashi-Narashino, Narashino-shi, Chiba No. Hitachi, Ltd.Industrial Equipment Division (72) Inventor Kazuhiro Imaya 3-1-1, Sachimachi, Hitachi City, Ibaraki Prefecture Inside Hitachi, Ltd.Hitachi Plant (72) Inventor Yasunori Nomoto Omika-cho, Hitachi City, Ibaraki Prefecture 5-2-1, Hitachi, Ltd., Omika Plant, Hitachi, Ltd. (72) Inventor Kiichi Tokunaga 7-1-1, Omika-cho, Hitachi, Ibaraki Pref. Hitachi, Ltd., Hitachi Research Laboratory, Ltd. (72) Inventor Tanaka Chief Tax Ibaraki (1-1) Katsubu Furukawa, Hitachi, Ltd. Hitachi, Ibaraki, Japan 3-1-1, Machi-cho, Hitachi, Ltd.Hitachi Plant (72) Inventor Takashi Hotta 7-1-1, Omika-cho, Hitachi-shi, Ibaraki Pref. Hitachi, Ltd.Hitachi Research Laboratory, Ltd. (72) Inventor Keijiro Sakai Ibaraki 7-1-1, Omika-cho, Hitachi City, Hitachi, Ltd. Within Hitachi Research Laboratory, Hitachi, Ltd. (72) Katsuhiro Okusawa 7-1-1, Omika-cho, Hitachi City, Ibaraki Prefecture, Hitachi, Ltd.

Claims (7)

[Claims] 1. A pulse width modulation power conversion device having a self-extinguishing type switching circuit and a diode circuit connected in an anti-parallel manner between a three-phase AC power supply and a DC smoothing circuit having a capacitor. For the arm connected to the phase of the phase AC power supply having the minimum phase voltage amplitude, the switching circuit performs pulse width modulation control, and for the arm connected to the remaining two phases, A pulse-width-modulated power conversion device, wherein the power is passed through a diode circuit, and the switching circuit is controlled to flow during an inverse conversion operation. 2. A pulse width modulation power conversion device having a self-extinguishing type switching circuit and a diode circuit connected in an anti-parallel manner between a three-phase AC power supply and a DC smoothing circuit having a capacitor. The phase with the smallest amplitude of the phase voltage of the
A minimum voltage detector for detecting the phase voltage phase, and an arm connected to the positive side of the capacitor and a negative side of the two arms connected to the phase selected by the minimum voltage detector. The self-extinguishing type switching circuit of the arm connected to is alternately ignited and extinguished to adjust the conduction ratio of the arm according to the command value. The remaining two phases are connected to the positive voltage phase.
Among the arms, the duty ratio of the positive side arm is kept at 1, and the duty ratio of the negative side arm of the two arms connected to the negative voltage phase is set at 1.
A gate control device that keeps the phase voltage phase of the first phase as an input, and in a period during which the voltage of the first phase changes from negative to positive, the flow of the positive side arm according to the detected phase voltage phase An AC voltage control device that continuously changes the duty command value from 0 to 1 and continuously changes the duty ratio command value of the negative arm from 0 to 1 during a period from positive to negative A pulse width modulation power converter. 3. A pulse-width-modulated power converter having a self-extinguishing switching circuit and a diode circuit connected in an anti-parallel manner between a three-phase AC power supply and a DC smoothing circuit having a capacitor. The phase with the smallest amplitude of the phase voltage of the
A minimum voltage detector for detecting the phase voltage phase, and an arm connected to the positive side of the capacitor and a negative side of the two arms connected to the phase selected by the minimum voltage detector. The self-extinguishing type switching circuit of the arm connected to is alternately ignited and extinguished, and during a period in which the phase voltage of the first phase changes from negative to positive, the positive side according to the detected phase voltage phase The flow rate of the arm is continuously changed from 0 to 1,
During the period when the phase voltage of the first phase changes from positive to negative, the duty ratio of the negative arm is set to 0 in accordance with the detected phase voltage phase.
A first AC voltage control device that continuously adjusts the voltage from 1 to 1, a positive voltage phase among the remaining two phases of the three-phase AC power source is selected as a second phase, and a positive voltage that detects this phase voltage phase is selected. A voltage detection device, and a self-extinguishing type switching circuit of a positive arm and a negative arm connected to the second phase are alternately fired and extinguished, and a phase at which the voltage of the second phase is maximized In the predetermined positive partial flow period centered on θ2, the negative arm is partially ignited to continuously and variably adjust the flow rate of the positive arm from 1 to the minimum set value at θ2. A second AC voltage control device that maintains the duty ratio of the positive arm at 1 during periods other than the positive partial conduction period; and a negative voltage phase, which is the remaining phase of the three-phase AC power supply, as a third phase. A negative voltage detecting device for detecting the phase voltage phase, and a positive-side arm and a negative-side arm connected to the third phase. The self-extinguishing type switching circuit is alternately fired and extinguished, and the positive side arm is partially moved in a predetermined negative side partial conduction period centered on the phase θ3 at which the voltage of the third phase becomes maximum on the negative side. The firing rate of the negative side arm is continuously variably adjusted from 1 to the minimum set value at θ3, and the flow rate of the negative side arm is maintained at 1 during periods other than the negative side partial flow period. A pulse width modulation power conversion device comprising a third AC voltage control device. 4. A power factor detecting device for detecting a power factor of the three-phase AC power source according to claim 2 or 3, a power factor detected value by the power factor detecting device, and a predetermined power factor setting And a power factor adjusting device for adjusting a voltage phase input to the AC voltage control device according to a deviation from the value. 5. The invention according to claim 1, wherein a voltage type inverter circuit connected to the DC smoothing circuit and an AC output of the inverter circuit are set to match a command of a variable frequency and a variable amplitude. A pulse width modulation power conversion device, comprising: a control device for performing pulse width modulation. 6. The AC voltage detection device according to claim 2, wherein the AC voltage detection device detects that the voltage of the three-phase AC power supply has exceeded a set range and generates a voltage abnormality signal. And an overvoltage protection device for preventing a firing command from the AC voltage control device to the self-extinguishing type switching circuits of all arms in response to the above, and releasing the firing command blocking after abnormal voltage output is stopped. A pulse width modulation power converter. 7. The AC current detection device according to claim 1, wherein the AC current detection device detects that the current of the three-phase AC power supply has exceeded a set range and generates a current abnormality signal. An overcurrent protection device is provided for preventing a firing command from the AC current control device to the self-extinguishing type switching circuits of all arms in response to the above, and releasing the firing command blocking after abnormal voltage output is stopped. A pulse width modulation power converter.