JPH1175134A - Method for detecting when co-channel interference NTSC signal and digital television signal are transmitted together using a special NTSC receiver - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a method for detecting when a digital television signal is accompanied by an NTSC signal interfering with a co-channel having a substantial amplitude. SOLUTION: A digital television signal sometimes accompanied by a co-channel interference analog television signal is received, and a video signal portion of the co-channel interference analog television signal is synchronized to a baseband and includes a first artificial noise. Generating a phase demodulation result and generating a quadrature phase demodulation result including second artificial noise; differentially shifting the in-phase and quadrature phase demodulation results by 90 °; Linearly combining the demodulation results to generate a linear combination result in which the first and second artificial noises are suppressed; detecting whether the amplitude exceeds a predetermined value; Generating a signal indicating a time associated with the co-channel interference analog television signal.

Description

DETAILED DESCRIPTION OF THE INVENTION

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a digital television for transmitting a digital television signal by using a high frequency in a television broadcast signal band, and more particularly, to a digital television signal which interferes with the same channel having a substantial amplitude. The present invention relates to a method of detecting a time accompanying a co-channel interferring (NTSC) signal in a digital television receiver.

[0002]

2. Description of the Related Art ATSC (Advanced Television Subcomm)
Digital Television Standards, published on September 16, 1995 by Ittee, Inc.
l Television Subcommittee) 6MHz such as those currently used for radio broadcasting of analog television signals
Vestigial sideband for transmitting digital television signals on a bandwidth television channel
nd: VSB) Specifies the name of the signal. As long as an NTSC analog television signal is broadcast, a certain NTSC
It would be advantageous to be able to detect in a receiver when an analog television signal would cause co-channel interference in a receiving digital television signal. Reducing the undesirable effects of the co-channel interference by designing the digital television receiver to change its mode of operation in response to the determination that co-channel interference is occurring; Which is what was normally done by comb filters.

A comb filter used in a digital television receiver to suppress NTSC co-channel interference is preferably not used if the co-channel interference does not occur. The reason is that a comb filter must be used. This is because Johnson noise can be prevented from being additionally generated from many paths passing through the comb filter. In general, co-channel interference generated from an NTSC analog television signal causes the digital television signal to be synchronized with the baseband by synchronizing.
Symbol decoding for (synchrodyning)
data slicing performed during (symbol decoding) (da
Co-channel interference is considered real if it has enough energy to cause an error in the operation. U.S. patent application Ser. No. 08 / 821,944, filed Mar. 21, 1997, entitled "Using video signals.
from auxiliary analog TV receivers for detecting N
TSCinterference in digital TV receivers '' determines whether substantial co-channel interference has occurred,
A digital television receiver is described that is designed to change its mode of operation accordingly, thereby reducing unwanted effects such as co-channel interference. After synchronizing the NTSC co-channel interference to the baseband, rather than synchronizing the digital television signal to the baseband from the patent,
It can be seen that it is easier to detect SC co-channel interference.

[0004] US Patent 5,122,879, issued June 16, 1992, entitled "Television syc".
hronous receiver with phase shifter for reducing i
In the `` nterference from a lower adjacent channel '', the received NTSC signal is in-phase and quadrature (q
An analog television receiver that detects synchronously in uadrature-phase) is described. This receiver synchronizes the response of the high-frequency amplitude device directly to the baseband so that adjacent low channels appear in the image. The quadrature phase synchronization detection response is 500 to 750 kHz.
90 ° out of phase at all video frequencies above z,
Linearly combined with the in-phase synchronization detection response to suppress image frequency components converted to a baseband during synchronization detection of the received NTSC signal. No. 5,122,879 describes the process as 750.
No mention is made of the fact that even video frequency components above kHz are lost. However, the loss of brightness of high frequency signals can be tolerated in small screen television receivers such as those used in wristwatches.

[0005] Current digital television receivers include:
Designed to use multiple frequency conversions, but ultra high frequency (UHF) above the channel band
The first conversion to an intermediate frequency in the band is specified in television broadcasting and is performed at very high frequencies below the channel band (very high).
frequency: VHF) Second to intermediate frequency in band
The conversion is specified in the television broadcast. Therefore, suppression of the image poses no further problem. In addition, since the carrier of the VSB digital television signal is located only 310 kHz from the channel edge, there is almost no double sideband component compared to the NTSC signal.

[0006]

SUMMARY OF THE INVENTION The inventor of the present invention has proposed an inverse Hilbert transform of an in-phase synchronous video detection response.
An NTSC receiver that linearly combines quadrature locked video detection responses is an auxiliary receiver for detecting when a digital television signal is accompanied by co-channel interfering NTSC signals having substantial amplitude. It is pointed out that it is important in receiving digital television signals because it is used as a digital television. The inverse Hilbert transform of the quadrature synchronized video detection response is
When adjusted to frequencies below 50 kHz, such auxiliary receivers do not respond to artifacts of co-channel digital television signals. If the artificial image of the digital television signal is suppressed, the size of the co-channel interference NTSC signal can be easily measured.

[0007]

According to one embodiment of the present invention, there is provided:
Digital television signal co-channel interference NTS
A method for detecting a time accompanied by a C signal by a digital television receiver is implemented. The method is performed in the following process. First, the video component of the co-channel interference NTSC signal is synchronized to a baseband to generate an in-phase demodulation result including a first artificial image (artifact) of the digital television signal, and a second demodulation result of the digital television signal. A quadrature demodulation result including an artificial image is generated. Next, the in-phase and quadrature-phase demodulation results are differentially shifted by 90 ° at a frequency of several kHz or more, respectively, and then linearly combined to remove the first and second artificial images of the digital television signal. Generate a linear combination result. Thereafter, detecting whether the amplitude of the linear combination result does not exceed a predetermined value, and indicating a time when the digital television signal is accompanied by a co-channel interference NTSC signal having a substantial amplitude.

[0008] Another embodiment of the present invention provides a digital television receiver that includes circuitry for detecting when an analog television signal having a substantial amplitude occupies a television broadcast channel. The receiver includes an input circuit that selects a vestigial sideband amplitude modulated signal from a television broadcast channel indicating a video signal portion of an analog television signal occupying the television broadcast channel. The converted residual sideband amplitude modulation signal is converted into an intermediate frequency signal, and the intermediate frequency signal is amplified to provide an amplified intermediate frequency signal. The vestigial sideband amplitude modulation signal received by the input circuit comprises a vestigial sideband, a video carrier and a full sideband (f
ull sideband).

A video synchro-dyne circuit is associated with the video carrier signal and synchronously detects the amplified intermediate frequency associated with the quadrature carrier of the video carrier signal to provide an in-phase synchronization detection response and quadrature synchronization. Generate a detection response. A phase shift circuit, referred to herein as an inverse Hilbert transform circuit, shifts all frequency components of the quadrature phase lock detection response above a predetermined frequency by 90 ° to generate a phase shift circuit response. The linear combination circuit linearly combines the response of the in-phase synchronization detection and the response of the phase shift circuit,
Restoring the linear combination circuit according to the video signal portion appearing in all the sidebands and the vestigial sideband of the received vestigial sideband amplitude modulation signal. The response of the linear combination circuit does not respond to the response of a digital television signal occupying the currently received television broadcast channel. A threshold detector is included in the receiver, wherein the threshold detector detects when the response of the first linear combinational circuit exceeds a predetermined threshold and causes the co-channel analog television signal to be substantially reduced. Signal having an appropriate amplitude.

[0010]

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS Hereinafter, the present invention will be described in detail with reference to the accompanying drawings. FIG. 1 shows a television receiver capable of receiving not only digital television signals but also NTSC analog television signals. A radio television broadcast signal received by an antenna 1 is amplified by an appropriately tuned radio frequency amplifier 2 to a first detector 3
Sent to The radio frequency amplifier 2 and the first detector 3
Functions as a tuner for selecting a digital television signal from any one of the channels that are appropriately tuned and located at different positions in the frequency band. The first detector 3 has a first tunable over a frequency range equal to or higher than an ultra-high frequency (UHF0 television broadcast signal).
A first local oscillator providing local oscillation, and mixing the first local oscillation with a television signal selected by the appropriately tuned radio frequency amplifier 2 to up-convert the selected television signal A first mixer for generating a UHF intermediate frequency signal in a 6 MHz wide UHF intermediate frequency band located at a frequency equal to or higher than the assigned channel frequency in the UHF television broadcast band.

The first detector 3 transmits the U-band signal to a UHF band intermediate frequency amplifier 6 used for NTSC audio reception.
Provides an HF intermediate frequency signal. The output of the UHF intermediate frequency amplifier 6 is the second output used for NTSC audio reception.
Applied to the detector 9. A second local oscillator for providing a second local oscillation of a predetermined frequency equal to or higher than a UHF television broadcast band frequency;
A second mixer for mixing the output of the HF intermediate frequency amplifier 6 to generate a VHF intermediate frequency signal located at a frequency equal to or lower than the assigned channel frequency in the VHF television broadcast band. This VHF intermediate frequency signal is VHF
Applied to the intermediate frequency amplifier 12.

The output of the VHF intermediate frequency amplifier 12 is applied to an intercarrier sound detector 34, which outputs a 4.5 MHz intercarrier sound intermediate frequency signal to an intercarrier sound intermediate frequency amplifier 35. Supply. The inter-carrier sound intermediate frequency amplifier 35 amplifies the inter-carrier sound intermediate frequency signal, limits the amplified signal symmetrically, and applies the symmetrically limited signal to the FM detector 36. FM detector 36 regenerates a baseband composite signal which is provided to the remaining components of the analog television receiver portion of the digital television receiver. The remaining components generally include stereophonic sound (stereopho
nic) Decoder circuit is included. If the NTSC audio signal is selected by narrow band filtering at the intermediate frequency amplifiers 6 and 12 that pass only the FM audio carrier and convert it to an intermediate frequency, the intercarrier sound detector 34 is a multiplier. The multiplier may be an intermediate frequency amplifier (10 or 11).
The intermediate frequency amplifier 1 as a video carrier selected by the multiplier by a narrow band filter responsive to the output of
2 is multiplied. If the NTSC audio signal is filtered and selected by said intermediate frequency amplifiers 6, 12 which pass both NTS audio and video carriers and convert them to an intermediate frequency in order to produce a "quasi-parallel"sound; , The intercarrier sound detector 34 is a simple rectifier or squarer.

The first detector 3 receives the NTSC video signal and A
A high-frequency band signal is supplied to a UHF band intermediate frequency amplifier 37 used for TSC reception. A surface-acoustic-wave (SAW) filter in UHF IF amplifier 37 determines all IF outputs for ATSC digital television signals and NTSC video signals,
SC audio signals are rejected. SAW filter is UH
It has a flat amplitude output over the remaining 6 MHz wide television broadcast channels converted to the F intermediate frequency band and a linear phase output over its passband. SAW in the UHF intermediate frequency amplifier 37
A preceding stage of the filter is provided with a transistor amplifier designed to drive the SAW filter from a predetermined source impedance that minimizes multiple reflections. In order to maintain the predetermined source impedance, while the gain of the transistor amplifier has a fixed value, the SAW
One that is sufficient to overcome the insertion loss in the filter is preferred. The output of the UHF intermediate frequency amplifier 37 is applied to a second detector 38 used for ATSC digital television reception and NTSC video reception. The second detector 38 mixes the output of the second local oscillator and the UHF intermediate frequency amplifier 37 with a second local oscillator that provides a second local oscillation of a predetermined frequency equal to or higher than the UHF television broadcast band frequency, and
A second mixer for generating a VHF intermediate frequency signal located at a frequency below the assigned channel in the HF television broadcast band.

The V output from the second detector 38
The HF intermediate frequency signal is applied to the VHF intermediate frequency amplifier 41.
Adjustment gain (cotrolled to provide amplification of B or more)
-gain) including a transistor amplifier stage. The VHF intermediate frequency amplifier 41 includes a reverse automatic gain control (AGC) responsive to its output signal level. This inverse AGC is for the linearity of the gain that it provides. The RF amplifier 2 is an intermediate frequency amplifier 4
7 having a delayed inverse automatic gain adjustment stage according to the output signal level.

The output signal of the VHF intermediate frequency amplifier 41 is applied to an ATSC symbol code detector 13 for detecting a baseband symbol code. The ATSC symbol code detector 13 uses an in-phase synchronous detector for detecting the vestigial sideband amplitude modulation of the data carrier and provides automatic frequency / phase control for a control oscillator that supplies a synchronous signal to the synchronous detector. automaitc frequency and pha
A quadrature phase locked detector is used to perform se control (AFPC). The in-phase synchronization detector operates in the analog domain, and its output is digitized to 10 bits by the analog-to-digital converter 14. The symbol code detector 13 and the subsequent analog-to-digital converter 14 convert the VHF band output of the intermediate frequency amplifier 47 into a final intermediate frequency band equal to or higher than the base band. The output of the second detector can be replaced with a digital synchro-dyne circuit that sinks in the baseband.

Such an alternative circuit is known from December 2, 1995.
US Patent C.B. Patel issued on December 5, "Digital
VSB detector with bandpass phase tracker, as for in
clusion in an HDTV receiver ", August 2, 1995
U.S. Pat. No. 5,548,618, issued on Dec. 0, entitled "Digital VSB detector with bandpass phase."
tracker using rader filters, as for use in an HDTV
receiver ". When a digital television signal is received, a direct signal derived from the reception of the pilot signal is detected by pilot carrier detector 15 along with a symbol code reproduced in the baseband, and a digital television enable signal is generated. The digital television enable signal adjusts the display portion of the digital television receiver to display a digital television image rather than an NTSC television image. The pilot carrier detector 15 is of a type corresponding to a digital input signal as shown in FIG. 1 or a type corresponding to an analog input signal provided directly from the symbol code detector 13.

FIG. 1 shows digitized baseband symbol codes supplied from the analog-to-digital converter 14 to the symbol decoder 20. The symbol decoder 20 has been disclosed by the inventor of the U.S. patent application Ser. No. 08 / 746,520 filed Jan. 12, 1996, entitled "Digital
levision receiver with adaptive filter circuitry fo
r suppressing NTSC co-channel interference ". The symbol decoder 20 performs data slicing on an input signal of the symbol decoder 20 to generate an output of the first symbol decoder.
An NTSC removal comb filter 22 for providing a response signal to suppress the NTSC co-channel interference signal to the symbol decoder 20, a data slicer 24 for data slicing the output of the comb filter 22 and correcting the output of the wrong symbol decoder, A matched comb filter 24 that corrects the output of the wrong symbol decoder to generate the output of the second symbol decoder, and one of the outputs of the first and second symbol decoders is supplied to the digital television receiver by the signal decoder 20. And a multiplexer 25 for selecting as the final symbol decoder output provided to the trellis decoder 16. NT at the time excluding the symbol decoder initialization period
If there is no received indication of the SC co-channel interference signal, multiplexer 25 selects the output of the second symbol decoder from matched comb filter 24 and provides the output signal of symbol decoder 20 to trellis decoder 16.

The multiplexer 25 provides an ideal symbol decoding result output from the memory in the television receiver during the period when the data segment synchronization and field synchronization code groups appear in the received digital television signal. Thus, the function of the symbol decoder 20 can be improved. The improved symbol decoder is disclosed by the present inventors in U.S. patent application Ser. No. 08 / 839,691 filed Apr. 15, 1997, entitled "Digital television rec."
eiver with adaptive filter circuitry for suppressi
ng NTSC co-channel interference ".

The output signal of the VHF intermediate frequency amplifier 41 is applied to a circuit 46 that synchronizes the modulation of the NTSC video carrier to the baseband. In-phase and quadrature-lock detectors are used in the circuit 46 to synchronize the modulation of the NTSC video carrier to the baseband. Synchronization is performed in the digital domain after being converted to a final intermediate frequency band above the baseband so that the final intermediate frequency can be digitized.
Also, the synchronization of NTSC video carrier modulation to the baseband can be performed in the analog domain, and the outputs of the in-phase and quadrature-phase synchronization detectors used for such purpose are analog-to-digital. Each can be digitized using a converter. The response (Q) of the quadrature phase locked detector is a component of the single sideband of the NTSC signal (that is, a component of 750 kHz or more) and the artificial noise of the digital television signal appearing in the response (I) of the phase locked detector. Hilbert transform. The Hilbert transform provided by the quadrature locked detector response (Q) is phase shifted and provides a 90 ° delay at all frequencies (except the lowest frequency with no response) by the inverse Hilbert transform circuit 47. .

Addition and subtraction are performed in a selective manner that is linearly combined. One of the linear combiners 47 and 48 becomes an adder, and the other becomes a subtractor. Circuit 47
The inverse Hilbert transform output is boosted to be linearly combined with the output of the in-phase lock detector in linear combiner 48 and corrected to a level to be provided to the rest of the analog television receiver circuit. A composite video signal having the calculated high frequency. In connection with the baseband composite video signal, the remaining portion can display a sync separation circuit, a color signal reproduction circuit, and a 4: 3 screen ratio NTSC image on a 16: 9 screen displaying a digital television image. Is included.

The inverse Hilbert transform output of the circuit 47 is linearly combined with the in-phase baseband output (I) of the synchro-dyne circuit 46 by a linear combiner 49 to be 750 kHz.
A luminance signal that cuts off the above is generated, and this luminance signal does not have digital television artificial noise. Whether the linear combiners 48 and 49 are an adder and a subtractor, respectively, or the linear combiners 48 and 49 are a subtractor and an adder, respectively, depends on whether the operation of the quadrature phase lock detector is the same. It is determined by whether the action is advanced or delayed.

FIG. 1 shows a squared representation of the energy of the NTSC co-channel interference signal after it has been filtered by a low-pass filter 50 having a cut-off frequency of 1 MHz and during the operation of receiving a digital television signal. 4 shows a band-limited luminance signal output from the linear combiner 49 squared by a squarer 31. The squarer 31 is a multiplier (multiplie
r) and a multiplier (multiplicand) to receive a signal.
It is even more practical to implement this with ly memory). The output signal of the squarer 31 indicates the energy of the NTSC co-channel interference signal at the time of receiving the digital television signal.

The digital threshold detector 32 sets a lower threshold than the NTSC co-channel interference signal that is insufficient to cause uncorrected errors to flow into the data slicer 21 so that the energy of the NTSC co-channel interference signal exceeds the threshold. Determine when it is big enough. Threshold detector 31
Is applied to the multiplexer control circuit 33. The multiplexer control circuit 33 controls an operation in which the multiplexer 25 selects the output of the first and second symbol decoders that determines the output of the final symbol decoder provided as the output signal of the symbol decoder 20. The multiplexer control circuit 33 controls the multiplexer 25 to select the output of the first symbol decoder as the output circuit of the symbol decoder 20 during the symbol decoder initialization period. During other periods, the multiplexer control circuit 33 determines that the output of the threshold detector 32 is insufficient to cause the uncorrected NTSC co-channel interference signal to flow into the operation of the data slicer 21. While showing
The multiplexer 25 controls so as to select the output of the first symbol decoder as the output signal of the symbol decoder 20. Otherwise, the multiplexer 25 controls the multiplexer 25 to select the output of the second symbol decoder as the output signal of the symbol decoder 20. Control to make it.

FIG. 2 shows the output of the linear combiner 49 as the squarer 3
1 shows a modified version of the apparatus of FIG. 1 to provide a digital threshold detector 32 without squaring by one. Since the output of the linear combiner 49 is essentially a baseband luminance signal of up to 750 kHz, it always has the same polarity. Therefore, the squarer 31 can be omitted, and the digital threshold detector 32 has a digital threshold detector 32 having a predetermined threshold corresponding to the square root of the predetermined threshold of the digital threshold detector 032. Can be substituted. That is, the predetermined threshold value of the digital threshold detector 32 is the square of the predetermined threshold value of the digital threshold detector 032.

In the television receivers of FIGS. 1 and 2, the requirement for the inverse Hilbert transform circuit 47 is reduced by including the low-pass filter 50, which reduces the cut-off frequency of the low-pass filter 50. This is because it is not necessary to provide a delay of exactly 90 ° in the frequency portion in order to suppress digital television artifacts in the frequency portion. If the inverse Hilbert transform circuit 47 provides a exactly 90 ° delay at a frequency of 4.2 MHz, the low-pass filter 50 can be replaced by a straight-through connection. In order to boost the composite video signal to a high frequency by combining the output of the inverse Hilbert transform filter 47 with the in-phase base low-pass output (I) of the synchro-dyne circuit 46 by the linear combiner 48, the inverse Hilbert transform circuit 47 operates as follows. Accurate 90 ° for frequencies up to 2MHz
It is not necessary to provide a delay in the video peaking circuit to compensate for the high frequency roll-off of the composite video signal with inaccurate delays if the delay error is not severe. because circuitry) can be used.

FIG. 3 shows a modification of the television receiver shown in FIGS. In FIG. 3, instead of the inverse Hilbert transform circuit 47 shifting the quadrature baseband response Q of the synchro dyne circuit 46 and applying it to the linear combiners 48 and 49, the inverse Hilbert transform circuit 51 uses the quadrature phase of the synchro dyne circuit 46 instead. The baseband response Q is shifted and applied to the linear combiner 48, and another inverse Hilbert transform circuit 49 shifts the quadrature baseband response Q of the synchro dyne circuit 46 and applies it to the linear combiner 49. The inverse Hilbert transform circuit 51 provides a precise 90 ° delay for frequencies between 0.5 MHz and 4.2 MHz to optimize the spectral response of the composite video signal, but a 90 ° delay for frequencies below 0.5 MHz. There is no need to provide. This allows for 0.5 MHZ with the high digital sampling ratio required to provide a 90 ° delay for frequencies up to 4.2 MHz.
The need for the many tap finite-impulse-response (FIR) filters required to provide a 90 ° delay at frequencies below z is eliminated. The high digital sampling ratio is also necessary to provide a 90 ° delay for frequencies up to 4.2 MHz. Because of the use of the low pass filter 50, the inverse Hilbert transform circuit 52 should provide an accurate 90 ° delay only for frequencies up to 1.0 MHz.

Inverse Hilbert transform circuit 52 provides a 90 ° delay at frequencies below 0.5 MHz below the NTSC scan line rate. Such a requirement can be satisfied at a digital sampling ratio of 1/10, which is four times lower than the digital sampling ratio used for the inverse Hilbert conversion circuit 51.
Reduces the need for temporary storage to provide differentially delayed sampling for FIR filtering within. 0.5 MHz for low-pass filter 50
With the following low cut-off frequency, the digital sampling ratio used in the inverse Hilbert transform circuit 52 can be designed to be eight times lower than the digital sampling ratio used in the inverse Hilbert transform circuit 51. . Further, the cutoff frequency of the low-pass filter 50 is divided into two halves once or several halves, and is used in the inverse Hilbert transform circuit 52.
Digital sampling ratio is inverse Hilbert transform circuit 51
Can be set to 1/10 of the digital sampling ratio used in the above.

FIG. 4 is a flowchart showing an operation method performed by the television receiver of FIG. In the initial stage S0, a digital television signal, sometimes accompanied by a co-channel interference analog television signal having a video signal portion, is received. Such a digital television signal receiving operation is performed by the components 1 and 2 of the television receiver shown in FIG. 2,
3, 37, 38, 41. The synchro-dyne circuit 46 performs the S1 step. In this step S1, the video signal portion of the co-channel interference analog television signal is synchronized with the base band and the in-phase demodulation result including the first artificial noise of the digital television signal is obtained. And generating a quadrature demodulation result including the second artificial noise of the digital television signal. Inverse Hilbert transform circuit 4
7 performs the S2 step, and in this step S2, 750 kHz
The in-phase and quadrature-phase demodulation results are differentially shifted by 90 ° at frequencies within the following predetermined frequency range.

The linear combiner 49 performs the S3 step. In this step S3, the in-phase and quadrature-phase demodulation results that are differentially shifted by 90 ° at frequencies within the predetermined frequency range are linearly combined, A combined result of the first and second artificial noises of the digital television signal within the predetermined frequency range is generated. (The low-pass filter 50 determines the upper limit of the predetermined frequency range.) The squarer 31 performs step S4, in which the linear combination result is squared. The digital threshold detector 32 performs a final step S5. In this step S5, it is detected whether or not the result of squaring the result of the linear combination exceeds the square of the predetermined value. With the co-channel interference analog television signal having a typical amplitude.

FIG. 5 is a flowchart showing an operation method performed by the television receiver of FIG. 4. The step S2 performed by the inverse Hilbert transform circuit 47 is different from that of FIG.
In step S2 ′ of FIG. 5, the quadrature phase demodulation result is shifted by 90 ° at a frequency within a predetermined frequency range of 750 kHz or less. FIG. 6 is a flowchart showing an operation method performed by the television receiver of FIG. The method shown in FIG. 6 also uses the same steps as S0, S1, S2, and S3 in FIG. The square performing step S4 performed by the squarer 31 of the television receiver of FIG. 1 is omitted in the operation of the television receiver of FIG. The step S5 performed by the digital threshold detector 32 of the television receiver of FIG. 1 is replaced with the step S5 ′ in the operation of the television receiver of FIG. 2, but at this step (S5 ′), a predetermined frequency is determined. Detecting whether the amplitude of the linear combination result within the range exceeds a predetermined value, generates a signal indicating when the digital television signal is accompanied by a co-channel interfering analog television signal having a substantial amplitude. This step (S5 ') is performed by the digital threshold detector 32 of the television receiver of FIG.

FIG. 7 is a flowchart showing a method of operation performed by the television receiver of FIG. 3. The step S2 of FIG. 4 performed by the inverse Hilbert transform circuit 47 is S2 'of FIG.
Different from stage. In this step (S2 '), the quadrature phase demodulation result is shifted by 90 ° at a frequency within a predetermined frequency range of 750 kHz or less.

[Brief description of the drawings]

FIG. 1 is a block diagram of a television receiver that can receive NTSC analog television signals and digital television signals and uses the method of the present invention to detect the presence of co-channel interfering NTSC analog television signals in digital television signals. It is a block diagram.

FIG. 2 illustrates a television receiver that can receive NTSC analog television signals and digital television signals and uses the method of the present invention to detect the presence of co-channel interference NTSC analog television signals in digital television signals. It is a block diagram.

FIG. 3 is a configuration diagram showing a modification of the television receiver of FIGS. 1 and 2;

FIG. 4 is a flowchart illustrating a method for detecting in a digital television receiver when a digital television signal is accompanied by a co-channel interfering NTSC signal having substantial amplitude, according to one embodiment of the present invention.

FIG. 5 is a flowchart similar to FIG. 4 according to another embodiment of the present invention;

FIG. 6 is a flowchart similar to FIG. 4 according to another embodiment of the present invention.

FIG. 7 is a flowchart similar to FIG. 4 according to another embodiment of the present invention;

Claims (8)

[Claims] Receiving a digital television signal sometimes accompanied by a co-channel interference analog television signal having a video signal portion; and synchronizing a video signal portion of the co-channel interference analog television signal to a baseband. Generating an in-phase demodulation result including a first artificial noise of the digital television signal, and generating a quadrature phase demodulation result including a second artificial noise of the digital television signal; Differentially shifting the in-phase and quadrature-phase demodulation results by 90 ° with respect to frequency; and 9 differently with respect to frequencies within the predetermined frequency range.
Linearly combining the in-phase and quadrature-phase demodulation results shifted by 0 ° to generate a linear combination result in which the first and second artificial noises of the digital television signal within the predetermined frequency range are suppressed; Detecting whether the amplitude of the linear combination result within the predetermined frequency range exceeds a predetermined value and indicating when the digital television signal is accompanied by a co-channel interference analog television signal having a substantial amplitude. Generating a signal. 2. The step of differentially shifting the in-phase and quadrature phase demodulation results by 90 ° at a frequency within a predetermined frequency range of 750 kHz or less includes the step of inversely Hilbert transforming the quadrature phase demodulation result. The method of claim 1, wherein: 3. The step of detecting whether the amplitude of the linear combination result exceeds the predetermined value includes squaring the result of the linear combination and squaring the result of the linear combination to obtain the predetermined value. The method of claim 1 including detecting whether it is exceeded. Receiving a digital television signal sometimes accompanied by a co-channel interference analog television signal having a video signal portion; and synchronizing the video signal portion of the co-channel interference analog television signal to a baseband. Generating an in-phase demodulation result including a first artificial noise of the digital television signal and generating a quadrature demodulation result including a second artificial noise of the digital television signal; and converting the quadrature demodulation result to 750 kHz or less. Shifting the phase by 90 ° at a frequency within a predetermined frequency range, and linearly combining the phase-shifted quadrature phase shift demodulation result with the in-phase and quadrature phase demodulation results, Result of linear combination with suppressed first and second artificial noise of digital television signal Generating; detecting an amplitude of the linear combination result in the predetermined frequency range exceeding a predetermined value, wherein the digital television signal is accompanied by a co-channel interference analog television signal having a substantial amplitude. Generating a signal indicative of: 5. A method for detecting an amplitude of said linear combination result in said predetermined frequency range exceeding said predetermined value, wherein said digital television signal is accompanied by a co-channel interference analog television signal having a substantial amplitude. Generating a signal indicating that, the step of squaring the result of the linear combination, and detecting a time when the result of squaring the result of the linear combination exceeds the square of the predetermined value, the digital television signal Has a substantial amplitude.
5. The method of claim 4 including generating a signal indicating that a C signal is present. 6. A digital television receiver, comprising: a circuit for detecting a time when an analog television signal having a substantial amplitude occupies a television broadcast channel, wherein the circuit occupies the television broadcast channel. A video signal portion of an analog television signal is shown, and a residual carrier side amplitude modulation signal including a video carrier and all sidebands is selected from a television broadcast channel together with the residual sideband, and the selected residual sideband is selected. An amplitude modulation signal is converted to an intermediate frequency signal, an input circuit that amplifies the intermediate frequency signal and provides an amplified intermediate frequency signal, and the video carrier signal and the quadrature carrier of the video carrier signal. Synchronously detecting the amplified intermediate frequency signal to generate an in-phase synchronization detection response and a quadrature phase synchronization detection response. A video syncdyne circuit, a first phase shift circuit that shifts all frequency components of the quadrature phase synchronization detection response equal to or higher than a predetermined frequency by 90 °, and a response of the in-phase synchronization detection response and the response of the first phase shift circuit A portion of the video signal that appears in the full sideband and the vestigial sideband in a linear combination to produce a response of the first linear combinational circuit unrelated to the response to the digital television signal occupying the television broadcast channel. First to recover
A linear combinational circuit; and a threshold for determining whether a response of the first linear combinational circuit exceeds a predetermined threshold to generate a signal indicating that the co-channel analog television signal has a substantial amplitude. A digital television receiver comprising a detector. 7. A linear combination circuit for recovering a response of a second linear combination circuit to the video signal by linearly combining a response of the in-phase synchronization detection and a response of the first phase shift circuit. The digital television receiver according to claim 6, wherein 8. A second phase shift circuit for generating a response of a second phase shift circuit by shifting all frequency components of the quadrature phase synchronization detection response of 500 kHz or more by 90 °, and a response of the in-phase synchronization detection. 7. The digital television according to claim 6, further comprising a second linear combination circuit that restores the response of the second linear combination circuit to the video signal by linearly combining the response of the second phase shift circuit with the response of the second phase shift circuit. John receiver.