JPS60102879A - Power converter - Google Patents

Abstract

PURPOSE:To reduce the switching loss by modulating a power source side PWM inverter by a carrier of the prescribed frequency, and modulating a motor side inverter by a carrier of variable frequency. CONSTITUTION:A capacitor C01 is connected to the DC side output of an inverter INV11, and connected to the DC side terminal of an inverter INV12. An induction motor M is connected to the AC side of the inverter INV12. An AC reactor LS2, an inverter INV13, a capacitor C02, an inverter INV14, and an induction motor M are connected similarly to the second secondary winding. Power source side inverters 11, 13 are modulated by a carrier of the prescribed frequency, and motor side inverters 12, 14 are modulated by a carrier wave of variable frequency.

Description

DETAILED DESCRIPTION OF THE INVENTION [Technical Field of the Invention] The present invention relates to a power converter device particularly suitable for driving a vehicle electric motor.

[Technical drawbacks and problems of invention]

FIG. 1 is a block diagram showing a conventional power conversion device. In the figure, TR is a power transformer, LS is an AC reactor, xnv
l is the first (D PWM inverter, INV2 is the second PWM inverter, co is the DC smoothing capacitor, I
M is induction motor, PG is rotary pulse generator, C0NT1
C0NT2 indicates a first PWM inverter control circuit, and C0NT2 indicates a second PwM inverter control circuit.

The first PWM inverter INVI is a thyristor Stt~
814 and diodes Dll-D14, and controls the current Isi supplied from the power supply so that the DC voltage Vo of the smoothing capacitor Co is constant. Therefore, the thyristor S 11-814 is for example gate turn-off (G
TO) Elements such as thyristors are used. In some cases, all forced commutation circuits are provided.

Second PWM inverter IN'V 21l-j: Cyrisk 821-826 and diodes D21-D26
and controls the three-phase current supplied to the motor IM. Similarly, elements such as a GTO thyristor are used for the thyristors S21 and ``''' s2+1, or a forced commutation circuit is added.

Figure 2 shows the implementation of the control circuit of the first 2wM inverter.
FIG. In the figure, ゛self, c2. c3 is a comparator, G1 (S), G2 (S) are control compensation circuits,
Km is a calculated amplifier, ML is a multiplier, TRG is a triangular wave generator, and GC is a gate circuit.

FIG. 3 is a block diagram showing an embodiment of the first control circuit of the second PWM inverter. In the figure, F/v is a frequency-voltage converter, Af is an adder, (r N + CN l +
CU 2 + cv I +CV2, CW+, CW
2 is a comparator, VRN l″i speed II f&9 unit, SF
C is slip frequency control circuit, GN(S), GIJ
(S) + Gv(S), (i\(S) is a control compensation circuit, PTG is a three-phase pattern generator, MLU, MLv
, MLw is a multiplier, 'rRG21d triangular wave generator, G
C, J, GCv, and GClv indicate gate circuits.

Hereinafter, the operation of the water device will be explained with reference to the above, FIG. 2, and FIG. 3.

A method for controlling the current supplied to the electric motor M will be described below.

Rotational speed Nf'i of electric motor IM, rotational pulse generator +℃
and an analog quantity is provided by an F/v converter. A comparator CN compares the speed command value N from the speed setting device VRN with the actual speed N. The deviation εN''=N''-N is input to the full control compensation circuit GN(s), and its output ILmk is used as a load current peak value command.

Further, a slip frequency afst of the electric motor IM is given by a slip frequency control circuit S'FC. Adder Af adds slip frequency fst and motor rotation frequency fm, and f. == Add fm+fst. The PTG generates a three-phase half-phase sine wave with a frequency fo. The three-phase pattern output PUIPV, PW is expressed by the following equation.

PU = sin (2πfo-t) Pv = sin (2πfo-t -2π/3)
Pw = sin (2πfo−t + 2π/3
) The multiplier MLU multiplies the peak value command ILrn and the unit sine wave pU to obtain a command value Ill of the current IU of the U-phase armature winding of the motor IM. Comparator CU
I is the detected value 1 of the U-phase current. and its command value IU, and the deviation εu=Iu-LU is input to the k-th control compensation circuit CU(S). Gu (S) is 17
! I is simply composed of a proportional amplifier, and the input ε1. A voltage proportional to 67. ′ is output. In another example, an integral element or a differential element may be added to optimize the current control system.

TRG2 generates an alternating current triangular wave V with a constant wave quotient, and serves as a carrier wave for a so-called PWM inverter.

The triangular wave Va and the output signal ε of the above GU(S). ' is compared, and when ε,'≧Va, an output signal "1" is outputted when ε.'<Va. Zyristor S2I of GCUi inverter INV 2
When the output signal is 11, S21 is turned on and 324k is turned off. Conversely, when the output (i) is 0", S21 is turned off and S24 is turned on, so when the period of 11" is wider than the period of 10", the U phase is increased in the positive direction, Conversely, when the period of 0" is wider than the period of 11", the U-phase current IU is increased in the negative direction.In other words, the control compensation circuit Go(S)
The voltage proportional to the output voltage of 6゜' is
is applied to the U-phase armature winding.

For example, if the command value IU of U@current becomes larger than the detected value IU, ε. =I, -IU is a positive value, 6
0', that is, the U-phase output voltage v'U is increased. Therefore ■. increases, and finally I,=1. It is controlled so that On the other hand, when I, (,I,), εU becomes a negative value, 80', that is, VTJ is made a negative value, and IUt is decreased.In the end, it settles down to IU. Here, if the command value IU is changed in the form of a total sine wave, the actual current flow is also controlled in a sine wave form following it.

Similarly, the V-phase and W-phase armature currents I, , Iw are also controlled according to the respective command values I, , I, .

As mentioned above, the peak values IL of the armature current command values IU, I, , Iw are given from the speed control circuit. When the speed command N is larger than the actual speed N, the deviation ε, =N-N becomes a positive value, increases the wave height value LLm, and increases the torque generated by the electric motor IM. vice versa,
In the case of N less than N, the deviation εN becomes a negative value, which reduces the above-mentioned wave height value Lmk and reduces the generated torque of the IM.

Therefore, in the end N = N and the drop tf<. Here, the control compensation circuit GN(S) uses a proportional and integral element, and is controlled so that the steady-state deviation εN is -rx. Differential elements may also be used to further enhance control response.

The slip frequency fSt of the electric motor IM is set to a positive value during power braking, and a phantom negative value during regenerative braking.
)'1-Slip frequency fst--method of controlling constant ■method of changing according to current peak value command ■Furthermore,
The primary current vector is controlled so that the excitation current of the motor IM and the secondary current maintain an orthogonal relationship (two-slip frequency f
There is a method that has changed. Here, an example in which a constant f8t is given from the slip frequency control circuit SFC is shown. 1. Slip frequency by f St> 07! When r L fc,
7. J% pj) 491 M 't'1 Power running mode (
1゛c, 11III machine mode) From the DC smoothing capacitor C8, which is a constant voltage source, 'ITt mg +'lQ
Power is supplied to the IM. On the other hand, when the collapsing frequency fst<0, tt' is the motive IM in regeneration mode (
II''l: Electric power is regenerated from the L motor IM to the DC smoothing capacitor C8, and a regenerative brake is applied to the 'tlLtU machine IM.

Next, a method of controlling the first PV, the current supplied from the power source by the 7M inverter NV1, will be explained.

In FIG. 2, vo is the DC voltage command value of the smoothing capacitor C6, which is the voltage detected value V by the comparator CI. compared to The temperature difference ε,=Vo−Vo is input to the next control compensation circuit C+(S), where it is amplified or integrated. The output of the control compensation circuit Gl(S) is the peak value of the electric current I8. I, , and is input to the multiplier ML.

On the other hand, detect the power supply voltage vS-■□・sinω8・L,
A unit sine wave synchronized with the power supply is generated through an operational amplifier Kmf. -j, that is, the peak value ■ of the power supply voltage is multiplied by the reciprocal number at Km. This unit sine wave and the above wave height value Ism(r
By multiplying, the current command value expressed by the following equation can be obtained.

*l5=Isrllll-8inωs-t However, for ωS, the power supply angular frequency comparator C2 compares the current command value Is with the power supply current detection value Isi to obtain the deviation ε2=Is−isf.

This signal is input manually to the next control circuit (ff circuit G2(S)) and amplified.

TRG is the conveyance skin of the PS width impark, that is, 50
07, which generates a unit triangular wave Vb, comparator C
By 3, the above sideill tjli borrowed circuit G2(S)
is compared with the output signal ε6 of .

From comparator C3, When εi≧vb, output signal %IN When εi<Vb, output signal 10 generated and input to the gate circuit GC.

The thyristor of the main circuit, when the above output signal is 91",
SI2 and S13 are turned on, and Sll and SI4 are turned off. Reverse (2. When the output signal is "0", S11 and 81
4 is turned on, and S12 and S13 are turned off.

11i 'tA current Is is determined by voltage vL applied to AC reactor Ls. In addition, the reactor voltage VL is the power 1' pressure Vs and the first PWM inverter I
The difference is determined by the voltage Ve generated by NV1.

The power converter with the configuration shown in Figure 1 always has an input power factor of 1.
．． Although it has all the features of being controlled to zero and having little harmonic current,
There are some drawbacks in terms of large capacity and high performance.

The first problem is the series or parallel connection of elements as capacity increases. Since it is difficult to make the turn-off times of GTOs and the like that have self-extinguishing ability to be the same, it is extremely difficult to connect the elements in series. Considering that capacitors require high voltage resistance, connecting elements in series is not a good idea. When increasing capacity (~
In this case, the elements must be connected in parallel, but the switching operation of the elements causes resonance between the lead wire inductance between the elements and the snubber circuit capacitor.
Make the total forward and reverse voltages of the element higher than necessary.

The second problem is an increase in switching loss associated with PWM operation. In general inverters and converters, one on/off operation is normally performed in one cycle of AC input (or output).

However, when PWM operation is performed to improve the current waveform, switching loss increases in proportion to the carrier frequency, and no improvement in system efficiency can be expected. Therefore, in order to improve the system efficiency kT%, it is important to reduce the carrier frequency range of inverter INV,l and inverter INV,2 without deteriorating the input characteristics such as power factor and harmonics, and the severity of motor control. is important.

The third problem is the redundancy of control 1.

It depends on the application, but if we assume a vehicle application, for example, it would be undesirable for even a minor failure to render the entire system inoperable and paralyze the route. Even if only 1/2 to 1/3 of the 11 deer can be used, it is required to minimize the disruption of the diamond. The 11th Yami-Yama - Power Change 1 Negative Device does not have such control redundancy.

[Purpose of the invention]

The present invention has been made to solve the above-mentioned technical problems in conventional devices, and it is possible to completely reduce switching loss without connecting multiple semiconductor devices in series and parallel, and to provide redundant power. The purpose is to provide a conversion device.

[l of invention! T,-! AI]

In order to achieve all of the above objects, the present invention uses a plurality of P'WM inverter units connected in parallel to provide control redundancy, and in order to reduce switching loss, a PWM inverter on the power supply side is used. The modulation is controlled by a constant carrier frequency, and the 2wM inverter on the load side is modulated by a variable frequency that is inversely proportional to the frequency of the motor.

[Embodiments of the invention]

FIG. 4 shows an embodiment of the present invention. Inverter INV,
11 and INV43 are the inverqueue NV, 1 in FIG.
It has exactly the same circuit configuration as the inverter NV, 12
and INV,14 have the same circuit configuration as the inverter INV,2. In the figure, Vs is a power supply, and TR is a transformer having two secondary windings. The first secondary winding is connected to the inverter IMV', 11 through the AC reactor L81''.
connected to. A capacitor Col is connected to the DC side output of the inverter INV,11.
Connected to the DC side terminal of No.2. Also inverter INV.

An induction motor M is connected to the downstream side of 12. Similarly, for the second secondary winding, AC reactor L8□, inverter INV, 13, capacitor C02, inverter I
NV, 14, guidance plan 111D machine M is in contact.

In this embodiment, since the configuration is described assuming all AC trains, the power supply is single-phase, and a case is cited in which there are a plurality of induction electric rvj machines. Since the rotational axes of the plurality of electric motors are commonly connected by a rail, it is sufficient to detect the rotational speed by one pulse generator PG.
One of the features is that both ends of the DC circuit, ie, capacitors C01 and C02, are connected in common, aiming for control redundancy.

Hereinafter, the entire operation of the embodiment will be explained using FIGS. 5 to 7. FIG. 5 is a control block diagram of inverters INV, 11 and INV, 13. The basic idea is the same as in Figure 2, but the input current command Is'(r''j, which is common to the two inverters so that the inverters INV, t■ and INV, 13 handle the same power supply) It has all independent 1u current control loops.The capacitor's 1σ current voltage command value Vd and upper side output value Vd are compared, and the deviation is given to the pressure control device Gv.The output of the control device tt, Gv and the power supply Multiply the voltage by the in-phase unit sine law sin act,
Assuming that it is 8, St/'i is the input current command necessary to make the capacitor voltage match the command value.

The input current command Is has a peak value that is proportional to the DC voltage deviation vd*-va, and is an effective current command that is in phase with the power supply voltage. Input current command value Is is inverter J IN
V, 11 current control device and inverter INV.

13 current control devices in parallel. Inverter 1
Regarding 1ff and 11, the command value Is and the detected value ISl are compared, and the deviation thereof is applied to the total current control device G1 and converted into a voltage command e1. TRGlt is a carrier wave of the PWM inverter, that is, a unit triangular wave Vbi is generated, and the inverter INV, lle is controlled via the gate control circuit GC1i according to the relationship between the voltage command e1 and the triangular wave ■. The current control method of the inverter INV, 13 is exactly the same, but the carrier wave for performing 2wM control, that is, the output signal of TRG12, can be made to have the same phase as the signal of TRGII, or it is possible to intentionally control TRG1. It is also possible to shift the phase with respect to the signal. The frequencies of the two carrier waves are constant.

As described above, the DC voltage (2) is established while drawing a current of 11 T (voltage) and voltage (Al), that is, while maintaining the input power factor of 1.0. Inverter INv, 11 and IN
V.

13 are controlled by giving the same current command 1111, so that the input currents are balanced in the same way.

Inverter INV, 12t INV, 14I:i,i
This is a two-way one inverter for converting all AC to RJ flow and driving an induction electric IIIJ machine. The control circuit is shown in Fig. 6 and is basically the same as that in Fig. 3, with one ? llr
, the rotational speed of the motor is detected by the pulse generator PG, and the speed is controlled.

Whether it is power or regeneration is determined by completely changing the slip frequency. iWM inverter INV, 12
The gate circuits of INV and j4 are common, and as shown in FIG.
On the one hand, all inverters rNV, 12-\On the other hand, all inverters INV, 14 are given. 1 The difference from Figure 3 is the configuration of the triangular wave generator TRG2. Conventionally, the frequency of the triangular wave was constant, or it rotated to vary the number of pulses in one cycle. In addition to the number, a method was used to improve the frequency of the triangular wave. In such a PwM control method, switching loss is extremely large, and no improvement in system efficiency can be expected. In electric motor control, the negative effects caused by current waveform distortion such as torque ripple are noticeable in the low speed range, and tend to be absorbed by the inertia in the high speed range. , the efficiency is improved by introducing a carrier wave that is a raw angle wave that is high in the low speed range and low in the high speed range.

A schematic block diagram is shown in FIG. 7(a). Each triangular wave is stored as a digital value in memory ROM. The rotational frequency fm of the electric motor detected by the pulse generator PG is converted by a frequency converter into a frequency approximately inversely proportional to f-2, as shown in FIG.

This frequency converter output fcZ is given to the memory ROM as the total clock frequency, and when the output is converted into D-+A, a triangular wave that is inversely proportional to the rotational frequency fm is generated. In this way, by using a variable frequency carrier wave according to the rotational speed of the electric motor, the moving parts of the system are improved. .

Another feature of embodiments of the invention is control redundancy. Although the detailed sequence is omitted, the DC circuit is common and the power supply side has multiple controllers (- and 9) in parallel.
The load side also has multiple innocutors (-1) and multiple electric motors.The power supply side innocutors (INV, 11 and INV, 1)
When one of the three inverters breaks down, if that inverter is removed, the motor can be driven even with half the power, and the power can also be regenerated. Motor side inverter INV, 12 and INV.

The same is true if any one of the 14 inverters fails, and as long as the failed inverter is removed, the minimum level of operation is guaranteed even if it is incomplete. Such redundancy is important in applications such as public transportation.

〔Effect of the invention〕

As explained using the embodiments, the present invention establishes a common DC voltage from a plurality of PWM inverters (-ta;2) from an AC power supply, and
A: Is there redundancy because the configuration is fully driven? In addition, the PWM inverter on the motor side can be modulated with a carrier wave of a constant frequency, and the switching loss can be reduced by modulating the PWM inverter on the motor side with a carrier wave of a variable frequency. Therefore, it is natural that the method can be applied even if the power supply is three-phase.

[Brief explanation of drawings]

FIG. 1 is a configuration diagram of a conventional power conversion device, FIGS. 2 and 3 are control block diagrams for explaining the operation of the device, and FIG. 4 is a power conversion device showing one embodiment of the present invention. The configuration diagrams of the apparatus, FIG. 5, FIG. 6, and FIG. 7 are control block diagrams for realizing an embodiment of the present invention. Vs...AC power supply TR...Power transformer L8.
Ls engineering, Ls□ ... AC reactor INV, l, I
NV, 2. INV, 1l-INV, 14 ・=-PWM
Inverter M...Induction motor PG...Pulse generator C0NTl, C0NT2-INV, 1, IN
V, 2] Control circuit co+Col+Co11'...
Capacitor TRG1. TRGzl TRG 11. T.R.
G 12...Triangular wave generator (7317) Representative patent attorney Kensuke Chika (and 1 other person) Figure 4 Figure 7 (0,)

Claims (1)

[Claims] an AC power supply, a first pulse width modulation inverter connected to the AC power supply via an AC reactor, a capacitor connected to the DC side of the pulse width modulation inverter, and a second pulse width modulation inverter that uses this capacitor as a voltage source. In a power conversion device comprising a pulse width modulation inverter, at least two of the first pulse width modulation inverters are provided and modulation is controlled using a carrier wave of a constant frequency, and at least two of the second pulse width modulation inverters are provided and the number of the second pulse width modulation inverters is variable. A power conversion device whose entire feature is modulation control using frequency carrier waves.