US20130009655A1 - Current sensor - Google Patents

Current sensor Download PDF

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Publication number
US20130009655A1
US20130009655A1 US13/580,377 US201113580377A US2013009655A1 US 20130009655 A1 US20130009655 A1 US 20130009655A1 US 201113580377 A US201113580377 A US 201113580377A US 2013009655 A1 US2013009655 A1 US 2013009655A1
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Prior art keywords
instrumentation amplifier
input terminal
current
instrumentation
switch
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Abandoned
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US13/580,377
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English (en)
Inventor
Victor Marten
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Sensata Technologies Inc
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Sendyne Corp
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Priority to US13/580,377 priority Critical patent/US20130009655A1/en
Publication of US20130009655A1 publication Critical patent/US20130009655A1/en
Assigned to SENDYNE CORP. reassignment SENDYNE CORP. ASSIGNMENT OF ASSIGNORS INTEREST (SEE DOCUMENT FOR DETAILS). Assignors: MARTEN, VICTOR
Abandoned legal-status Critical Current

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    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
    • G01R1/00—Details of instruments or arrangements of the types included in groups G01R5/00 - G01R13/00 and G01R31/00
    • G01R1/20—Modifications of basic electric elements for use in electric measuring instruments; Structural combinations of such elements with such instruments
    • G01R1/203—Resistors used for electric measuring, e.g. decade resistors standards, resistors for comparators, series resistors, shunts
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
    • G01R1/00—Details of instruments or arrangements of the types included in groups G01R5/00 - G01R13/00 and G01R31/00
    • G01R1/30—Structural combination of electric measuring instruments with basic electronic circuits, e.g. with amplifier
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
    • G01R15/00—Details of measuring arrangements of the types provided for in groups G01R17/00 - G01R29/00, G01R33/00 - G01R33/26 or G01R35/00
    • G01R15/14—Adaptations providing voltage or current isolation, e.g. for high-voltage or high-current networks
    • G01R15/146—Measuring arrangements for current not covered by other subgroups of G01R15/14, e.g. using current dividers, shunts, or measuring a voltage drop
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
    • G01R19/00—Arrangements for measuring currents or voltages or for indicating presence or sign thereof
    • G01R19/0092—Measuring current only
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
    • G01R29/00—Arrangements for measuring or indicating electric quantities not covered by groups G01R19/00 - G01R27/00
    • G01R29/24—Arrangements for measuring quantities of charge
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
    • G01R35/00—Testing or calibrating of apparatus covered by the other groups of this subclass
    • G01R35/005—Calibrating; Standards or reference devices, e.g. voltage or resistance standards, "golden" references
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01R—MEASURING ELECTRIC VARIABLES; MEASURING MAGNETIC VARIABLES
    • G01R19/00—Arrangements for measuring currents or voltages or for indicating presence or sign thereof
    • G01R19/0023—Measuring currents or voltages from sources with high internal resistance by means of measuring circuits with high input impedance, e.g. OP-amplifiers

Definitions

  • an electric power meter is a familiar fixture; it allows the power-providing companies to charge their customers for the energy used.
  • the instantaneous power is a product of the voltage applied to, and the current flowing through, the load.
  • An integration of this product (power) over a specific time interval yields the total consumption of the electrical energy within that time interval.
  • Current-measuring apparatus further differ in their ability to measure DC (direct current) or AC (alternating current), and also differ in the total range of frequencies that can be processed with a high degree of accuracy.
  • the apparatus that can accommodate large currents also differ in their ability to measure small currents accurately.
  • Typical current sensors in industrial, commercial, and household environments furnish the AC measurements; the parameter of interest is usually the peak-to-peak or Root-Mean-Square (RMS) value of the current, averaged across at least a single cycle of the AC frequency; in such measurements the DC component of the current is assumed to be exactly zero.
  • RMS Root-Mean-Square
  • the ultimate use of these measurements is for an overall consumption of energy, and the sensor is made to be quite accurate when a large level of current is flowing. Errors resulting from inadequate precision of measurements for small levels of current are simply ignored, as they contribute only minimally to the total energy assessment. Stated differently, when one is measuring large currents to assess overall energy consumptions, one does not mind that small currents are not measured accurately.
  • FIG. 1 A current sensor circuit based on this principle is illustrated in FIG. 1 .
  • input terminals 1 allow connection of the current shunt 2 into the circuit where current is to be measured.
  • Instrumentation amplifier (IA) 6 senses voltage across the current shunt 2 , and provides (voltage) output on terminals 10 .
  • Pick-up points 3 on the current shunt 2 are located following the principle of “Kelvin sensing” that reduces errors associated with resistance of the sense connections and wires.
  • RFI filter 4 is utilized for reduction of effects due to interfering RF signals from surrounding machinery and environment.
  • This part of the circuit may have various configurations, and may consist of resistive, capacitive, and inductive components, as well as semiconductor transient protectors.
  • the particular configuration 4 shown in FIG. 1 is exemplary but there are many possible arrangements which might be employed.
  • FIG. 1 shows an instrumentation amplifier 6 modeled as containing an ideal amplifier 7 .
  • the ideal amplifier 7 is imagined to have infinite input impedance, zero output impedance, linear gain, and zero offsets or drift, all across some defined dynamic range of inputs and from DC to at least some frequency defining some frequency response.
  • Also shown in FIG. 1 are modeled sources of error 5 , 8 , and 9 that make the performance less than ideal when compared to the requirements above.
  • the modeled voltage sensing error sources 5 appear in series with the proper signal, and introduce unpredictable offset errors at the inputs of the IA 6 . These errors can be further modeled as illustrated in FIG. 2 .
  • FIG. 2 items comprising a string of dissimilar materials 13 , 14 , 15 , 17 , 18 , and 20 are joined by solder 16 .
  • Exemplary items may be a Kelvin connection lead 13 from the current shunt 2 , a surface-mount component 15 such as a resistor, an IC (integrated circuit) lead 17 , a lead-to-die bonding wire 18 , and IC die metallizations 20 at the IC die 19 .
  • a printed circuit (PC) board 12 is composed in part of a substrate material such as FR-4.
  • a modeled temperature gradient 21 is shown, with a higher temperature at one end of this string and a lower temperature at the other end of this string.
  • thermocouples each capable of producing a voltage when a temperature difference (temperature gradient 21 ) exists between the extremes of the circuit at points 13 and 20 , or inside of the string of the serially connected elements.
  • This error voltage could potentially be on the order of several millivolts (mV).
  • mV millivolts
  • the error source 8 is a modeled offset error in the IA 6
  • the error source 9 is a modeled noise error in the IA 6 .
  • thermocouple-induced errors are to some extent common-mode in nature (affecting each line of the two differential inputs in somewhat the same way). But any non-identical temperature distributions as between the two paths can give rise to errors which are not automatically compensated by the use of differential sensing.
  • the circuits similar to FIG. 1 are generally used for measuring currents up to perhaps 10 A (Amperes).
  • the designer of the current sensing system will typically select a shunt value such that the voltage drop across the sensing resistor 2 is 100 mV or more, so that this voltage will swamp or overwhelm the error sources just described.
  • the resistance of the shunt 2 will be chosen so that the voltage developed across it is within some range of voltages. This leads to a situation where the heat dissipated in the sense resistor 2 grows at least linearly with the rated maximum current for which the particular circuit is being designed.
  • the physical size of the sense resistor 2 can thus get to be a problem, as can be the need to providing adequate cooling of the sense resistor. For this reason, for higher currents a different class of current sensing devices is used, typically utilizing an indirect method of current sensing by assessment of the magnetic field that is created around the conducting wire which is carrying the current.
  • Hall-effect apparatus is common for current measurements in excess of several Amperes.
  • a Hall-effect device is able to produce a (differential) voltage signal when a continuous (supply) current is sent through the device, and a magnetic field is present that is perpendicular to the flow of that supply current.
  • the voltage signal in the Hall-effect device is linearly proportional to both the supply current and the magnetic field, within limitations of power dissipation resulting from the supply current, and some additional anomalous effects.
  • the current that needs to be measured generates the magnetic field acting on the device.
  • the sensitivity of the Hall-effect device to the magnetic field depends, among many things, for example upon mechanical dimensions of the sensing element, material composition and uniformity, attachment of the electrodes, stability and accuracy of the supply current, and construction of the magnetic core that concentrates the desired magnetic field and rejects interfering magnetic fields (of which the Earth's magnetic field is just a single example).
  • the native offset of the unaided Hall-effect device is nonzero. For this reason various auto-zeroing schemes have been utilized in the prior art.
  • One approach is to arrange for the supply current to the sensing element to be AC rather than DC. Within such an approach, differing schemes make use of various shapes or waveforms of the AC current, for example sine waves or square waves or even more complex shapes.
  • the AC signal that is the output of the Hall-effect sensing element is further processed by a synchronous detector.
  • a further potential difficulty with such magnetically coupled measurements is that during the zero crossings of the excitation voltage (that is, near the zero values for the sine-wave excitation), the Hall-effect device is altogether insensitive to the magnetic field, and simply discards any information for the duration of the zero-crossing transitions.
  • Hall-effect devices make use of an active feedback loop that tries to zero-out the total magnetic field acting on the sensing element, thus reducing possible non-linearities in the sensing element (since in this case, the Hall-effect sensing element only needs to indicate if the magnetic field is smaller or higher than zero, and does not need to supply the actual value).
  • a winding on the magnetic path elements creates a magnetic field that is opposite to the field from the measured current.
  • the opposing winding is normally constructed with many turns (typically, with several thousands of turns), so as to minimize the current that needs to be injected into the winding in order to zero out the total magnetic field.
  • the servo loop that drives the zeroing-out winding will of course take some nonzero time to settle and to respond to perturbations.
  • Hall-effect devices utilizing the active feedback will require additional operating energy in order to supply the feedback circuitry and winding. These factors make the Hall effect sensing less than ideal, particularly for a battery-powered system.
  • MR magnetoresistance
  • GMR Giant Magnetoresistance
  • CMR Colossal Magnetoresistance
  • An apparatus and method make use of a single shunt and two or more instrumentation amplifiers, switchably measuring voltages at the shunt. Dynamic range is several orders of magnitude better than known current measurement approaches, permitting coulometry.
  • the current invention teaches a circuit and an algorithm (for the control part of the circuit, and for combined operations of the analog and digital parts of the circuit).
  • This circuit and the algorithm empower the creation of a current sensor that is accurate at much larger range of measured currents than in the prior art; it provides several-orders-of-magnitude dynamic range improvement.
  • the current invention reduces the waste of energy in the sensing element to near zero.
  • the energy consumption of the circuit itself can be reduced to near zero levels under conditions of low reporting rates for the measured current and/or accumulated charge. Due to very low measurement offset for the DC current, an accumulated charge value is quite accurate, even under conditions of a load current that is many times smaller than the maximum rated load current.
  • the high accuracy of the current sensor is ensured due to continuous assessment of most of the interfering voltage errors, and subtracting these errors from the actual measured value.
  • the output of the circuit according to the current invention is continuous, if such an operating mode is desired; this is in spite of the fact that all circuits in the sensor alternate between the actual measurement and the calibration functionality (in order to measure and zero-out the errors).
  • the ability to provide a continuous output is important if fast-acting electronic fuse functionality is desired.
  • the ability to provide a continuous output is important as well if it is desired to integrate the measured current so as to achieve coulometry.
  • the actual frequency, with which the calibration cycles are performed can be fixed or it can depend on the changes in the temperature or temperature gradients that are the dominant causes of the errors. In other words, the calibration cycles can be performed more frequently if the temperatures and/or temperature gradients change rapidly.
  • FIG. 1 depicts the prior art circuit
  • FIG. 2 illustrates generation of voltage errors
  • FIG. 3 describes one current measurement approach according to the invention
  • FIG. 4 shows the algorithm for activations of FETs 73 / 74 and Mux 48 signal in FIG. 3 ,
  • FIG. 5 presents a physical implementation that reduces uncompensated errors
  • FIG. 6 demonstrates an alternative embodiment with two RFI filters in each channel
  • FIG. 7 shows the apparatus in a second embodiment according to the invention, with a single shunt and a two-FET switching circuit, and
  • FIG. 8 shows an alternative embodiment according to the invention, with a two-FET switching circuit and two RFI filters in each channel.
  • FIG. 3 One current sensor according to the invention is shown in FIG. 3 .
  • a current shunt 72 providing a voltage signal proportional to the current flowing in the shunt 72 to resistors 75 , which are in turn connected to field effect transistor (FET) switches 73 / 74 .
  • FET field effect transistor
  • current sense signals pass via RFI filters 37 and 38 , and then get amplified by instrumentation amplifiers (IAs) 41 and 42 .
  • An analog selector switch 43 delivers the signal to output terminals 44 ; this signal can be either the output from IA 41 , or IA 42 .
  • the FETs 73 / 74 are turned either fully on or off, as required for the execution of the Algorithm detailed in FIG. 4 .
  • FETs 73 / 74 can short-circuit the signal voltage input to RFI filters 37 / 38 ; offset voltage errors associated with the RFI filters and the whole amplification chain 37 / 39 / 41 or 38 / 40 / 42 can then be calibrated out.
  • the current sense signal is removed from one of the measurement channels by activation of the shunt FET (e.g. 73 or 74 ); the current sense signal is reduced in proportion to the ratio of resistors 75 and on-resistance (so-called Rds-on) of the switches 73 / 74 .
  • Control Circuit 51 adjusts the output of Digital-to-Analog (D/A) converter 46 , until the output of the channel being calibrated is equal to zero.
  • D/A Digital-to-Analog
  • switches 73 and 74 there are only two MOSFET switches (as opposed to four in one previous circuit), and these switches 73 and 74 can be much smaller, and are only required to be able to carry a very small current, and thus are much less expensive than the units described in one previous circuit.
  • the on resistance (Rds-on) parameter of switches 73 and 74 can be tailored to the values of resistors 75 , and indeed the ratio between the resistances of resistors 75 and Rds-on for the switches 73 / 74 define the ultimate accuracy and dynamic range of the circuit.
  • Current shunt 72 ( FIG. 5 ) is constructed from copper utilizing common Printed Circuit Board (PCB) techniques. This current shunt does not have explicit sensing leads; sense lines 78 going to resistors 75 are continuous in respect to the body of the current shunt 72 .
  • PCB Printed Circuit Board
  • thermoelectric voltages there is no junction of dissimilar materials between the sensing leads on the current shunt and copper traces connecting to the rest of the circuit, thus no thermoelectric voltages can be generated.
  • the substrate of this PCB 70 may be a standard material called FR-4, or any other suitable for the required operating temperature range.
  • Components 75 , 73 , and 74 are closely grouped together in a small area of the PCB adjacent to the current shunt 72 .
  • thermoelectric voltage across each of two resistors 75 that are connected to the same MOSFET switch 73 or 74 will have the alike value, and will cancel each other due to the differential-sensing circuit that follows it.
  • a slot (air gap) 77 is cut into the PCB; this prevents the heat flux from escaping the area of the grouped components through heat conduction via the PCB substrate, effectively enforcing the same temperature across the whole area of the grouped components 75 , 73 , and 74 .
  • connections to the rest of the circuit are made through vias 76 . While these connections to the rest of the circuit will pass through the areas with uneven temperatures and will indeed produce thermoelectric error voltages, their effects will be totally negated by the calibration action of the circuit.
  • the two amplification channels are present in order to allow a single channel to provide the continuous readout of the current, while the other channel is calibrated. Calibration is performed when the current sense signal has been short-circuited by one of the MOSFET switches 73 or 74 , and RFI filter 37 or 38 with corresponding IA 41 or 42 have settled to a final value that represents zero value of measured current; then auto zeroing action is executed by Control Circuit 51 (e.g. adjustment of the D/A to null-out the output of the appropriate IA 41 or 42 ). Once calibration is finished, the inactive channel is switched back into service, by opening one of the MOSFET switches 73 or 74 .
  • the Analog Switch 43 is used to switch the Output terminals 44 to this newly calibrated channel.
  • the Control Circuit 51 can cross correlate the two signals; total accuracy of the Current Sensor can be improved by averaging the two values, and problems with the circuit can be discovered and reported if the readings from two channels start to diverge.
  • the user may elect to connect the local circuit's common potential (e.g. ground) to the middle of the current shunt 72 through the connection 79 .
  • the ground connection 79 to the shunt provides a reference voltage defining a fixed level at the sources of the MOSFETs 73 , 74 . This permits the system designer to select the output voltage at drivers 45 to fully turn the MOSFETs 73 , 74 on or off.
  • resistors 75 It may be instructive to say a little more about the resistors 75 .
  • the value of these resistors cannot be permitted to be too large because mismatch in the bias current of the AI will create systematic offset error due to I*R losses.
  • a 1 nA bias mismatch (which is a possible value for an IA) across a 100 Ohm resistor will create an offset of 100 nV.
  • the same sorts of errors from resistors in block 37 do not affect the output of an IA, since those errors are “after” the MOSFETs 73 , 74 in the signal path, and these errors are zeroed by the calibration/autozero action.
  • resistors 75 While it would be a problem if the resistors were too large in value, it is also the case that resistors 75 cannot be permitted to be too small, since we want to attenuate the signal across shunt 72 as much as possible when autozero is being performed. In other words—the ratio between the value of resistors 75 and the “on” resistance of MOSFETs 73 , 74 should be as high as possible. So consider for example the use of 100-Ohm resistors 75 , with MOSFETs 73 , 74 that have 10 milliOhms on resistance (which is quite low). In this example, the largest error coming from this non-infinite attenuation when a MOSFET is on is on the order of a full-scale signal divided by 10k, or only 80 dB of signal-to-noise.
  • An arbitrarily selected point in time 81 is chosen as the beginning of the algorithmic cycle; at this time both MOSFET switches 73 and 74 are off, and both channels are providing the current signal; however Mux 48 signal has just selected the output of IA 41 to be passed to output terminals 44 .
  • the switch 74 is turned on, and the current sense signal (voltage) at RFI filter 38 is short-circuited by MOSFET switch 74 .
  • the voltage at the input terminals of IA 42 should be exactly zero; however, due to previously described effects, the voltage at the input terminals of IA 42 , after passage through the sense connections and RFI filter 38 , contains some systematic offset error (mostly due to thermoelectric voltages generated at junctions of dissimilar materials having different temperatures).
  • the RFI filter 38 starts to settle toward the final stable value.
  • the RFI filter 38 has finished settling and provides a stable value to the IA 42 ; the output voltage of IA 42 is sampled by one of the channels of A/D 47 and/or one of the D/A 46 outputs is changed until output from IA 42 is equal to zero.
  • the channel related to open switch 73 supplied the output signal.
  • the complete cycle 80 repeats indefinitely.
  • the duration of the complete calibrate/measure/settle/switchover cycle 80 can be adjusted, depending on the rate of change of the temperature and/or temperature gradients over the circuit. If it is desired to adjust this cycle, for example to reduce the amount of time spent in calibration and zeroing activity, then the temperature should be sensed with a suitable sensor or sensors. Such sensors are omitted for clarity in FIG. 3 .
  • temperature measurements will be used to compensate for the temperature dependency of the resistance of the current shunt 72 .
  • the current shunt is created with a copper pattern on a PCB (and thus avoids the necessity to procure and assemble separate discrete components); in this case the dependency of the copper used for the conductive tracks on the PCB would be well known and predictable, and the resistance value changes could be readily compensated by digital calculations in the control circuit 51 .
  • the continuous analog output at output terminals 44 can also be adjusted to compensate for the current shunt's resistance changes; however, the main application of the continuous analog output is to provide the fast electronic solid state fuse capability which may not require very high precision, and may well be served sufficiently with a value that has not been compensated for resistance changes with temperature. Nevertheless, the circuit required for the adjustment of the continuous analog output will involve utilization of programmable-gain functionality in IA 41 or 42 ; this will be familiar to the reader.
  • the initial full-scale error created by an inaccurate value (due to manufacturing tolerances) of the shunt 72 at room temperature may be calibrated out by passing an accurately known current through the shunt 72 , and recording (preferably in the local non-volatile memory) an appropriate value for the digital adjustment of gain for the whole current sensor system.
  • the current invention provides sufficient and arbitrarily long time for settling of the RFI filters 37 and 38 , and is able to accommodate filters with extremely long time constants. This is beneficial for the accuracy of the measurements when output of the current sensor is used for accumulation of the total charge passed through the input terminals 30 (e.g. Coulomb counting). If such an integrated current value is maintained and calculated in the digital part of the circuit, an increase in the time constant of the RFI filter can be taken advantage of for lowering of the required sampling rate of the A/D converters 47 ( FIG. 3 ).
  • FIG. 6 A solution for this problem according to the current invention is disclosed in FIG. 6 .
  • This drawing shows one channel of measurements that contains two RFI filters 37 and 52 , and two corresponding amplifiers 41 and 54 ; all of these are associated with switch 73 .
  • the other channel (related to switch 74 ) is not shown for clarity of the drawing; nonetheless, it also contains two RFI filters and two instrumentation amplifiers.
  • the output of the IA 41 is connected to the output terminals 44 via Switch 43 (as shown in FIG. 3 );
  • RFI filter 37 has a time constant that is appropriate for the solid-state fuse functionality.
  • the output of the IA 54 is only connected to the additional A/D input at the control circuit 51 , the value on the output of IA 54 will only be reported in digital form after processing in the control circuit 51 , and it does not require the offset-compensation D/A (as the calibration value would be subtracted digitally).
  • the RFI filter 52 has a time constant that is suitable for the measurements of the charge passing through the current shunt 72 , while utilizing a slow sampling rate of the A/D.
  • RFI filter 52 has a time constant that is many times longer than the time constant of RFI filter 37 . The timing of the operating algorithm (as illustrated in FIG. 4 ) is thus set to properly service the timing requirements of the RFI filter 52 (since RFI filter 37 has a shorter time constant, its settling time requirements will certainly be satisfied).
  • FIG. 6 can be used simultaneously for fast electronic solid state fuse functionality via a continuous analog output, and for precise measurements of the accumulated charge that is reported digitally by control circuit 51 .
  • the approach shown in FIG. 6 has a “slow” signal processing path that maintains accuracy and is not dependent upon any requirement of high performance by the “fast” signal processing path.
  • the circuit firstly limits the bandwidth with a “slow” filter 52 (utilizing simple passive components for the filter 52 that are naturally highly linear even in the presence of transients). Then the output of the “slow” filter 52 is amplified by IA 54 . It is noted that IA 54 can thus be a relatively low-bandwidth unit, with correspondingly very low power consumption.
  • control circuit 51 will be best served by the use of a microprocessor; the whole circuit 51 can in fact be contained in a single-chip microcontroller that also includes the required functionality of A/D 47 , D/A 46 , communication interface 50 , and port pins that will serve as Mux signal 48 and FET Drivers 45 .
  • the instrumentation amplifiers can reside on the same chip for a compact and inexpensive solution.
  • FIG. 7 and FIG. 8 Alternate circuits are shown in FIG. 7 and FIG. 8 . These circuits replace the resistors 75 of FIG. 3 , FIG. 5 and FIG. 6 with FETs 96 / 97 . Depending on exact implementation details and requirements, the circuits in FIG. 7 and FIG. 8 may have advantages over the circuits in FIG. 3 and FIG. 6 , and vice versa.

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  • Physics & Mathematics (AREA)
  • General Physics & Mathematics (AREA)
  • Measurement Of Current Or Voltage (AREA)
  • Measuring Instrument Details And Bridges, And Automatic Balancing Devices (AREA)
  • Amplifiers (AREA)
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JP2015045640A (ja) * 2013-08-15 2015-03-12 リニアー テクノロジー コーポレイションＬｉｎｅａｒ Ｔｅｃｈｎｏｌｏｇｙ Ｃｏｒｐｏｒａｔｉｏｎ 電流センス回路、およびモータの巻線を駆動するための回路
US9041390B1 (en) * 2014-09-03 2015-05-26 Neilsen-Kuljian, Inc. Digitally controlled high-current DC transducer
US20150192642A1 (en) * 2014-01-07 2015-07-09 Qualcomm Incorporated Compensation technique for amplifiers in a current sensing circuit for a battery
US9217759B2 (en) 2011-12-23 2015-12-22 Sendyne Corporation Current shunt
US20160245848A1 (en) * 2014-08-27 2016-08-25 Sendyne Corporation Attachment of leads having low thermoelectric errors
DE102015204519A1 (de) * 2015-03-12 2016-09-15 Dialog Semiconductor (UK) Ltd Genaue Stromerfassung
US20160349775A1 (en) * 2014-02-13 2016-12-01 Robert Bosch Gmbh Current detection device and method for sensing an electrical current
US20160349289A1 (en) * 2014-02-13 2016-12-01 Robert Bosch Gmbh Circuit arrangement and method for current measurement
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US20150268277A1 (en) 2015-09-24
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US9588144B2 (en) 2017-03-07
WO2012117275A3 (fr) 2012-11-08

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