US9444342B2 - Clocked pulse frequency modulation buck DC-to-DC converter - Google Patents

Clocked pulse frequency modulation buck DC-to-DC converter Download PDF

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US9444342B2
US9444342B2 US14/516,859 US201414516859A US9444342B2 US 9444342 B2 US9444342 B2 US 9444342B2 US 201414516859 A US201414516859 A US 201414516859A US 9444342 B2 US9444342 B2 US 9444342B2
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current
limit
signal
converter
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US20160111956A1 (en
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Mark Childs
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Dialog Semiconductor UK Ltd
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/02—Conversion of DC power input into DC power output without intermediate conversion into AC
    • H02M3/04—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters
    • H02M3/10—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/158—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/02—Conversion of DC power input into DC power output without intermediate conversion into AC
    • H02M3/04—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters
    • H02M3/10—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/02—Conversion of DC power input into DC power output without intermediate conversion into AC
    • H02M3/04—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters
    • H02M3/10—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/02—Conversion of DC power input into DC power output without intermediate conversion into AC
    • H02M3/04—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters
    • H02M3/10—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/158—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
    • H02M3/1582—Buck-boost converters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/02—Conversion of DC power input into DC power output without intermediate conversion into AC
    • H02M3/04—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters
    • H02M3/10—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/158—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
    • H02M3/1588—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load comprising at least one synchronous rectifier element
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0032—Control circuits allowing low power mode operation, e.g. in standby mode
    • H02M2001/0032—
    • H02M2003/1566—
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/02—Conversion of DC power input into DC power output without intermediate conversion into AC
    • H02M3/04—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters
    • H02M3/10—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/1566—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators with means for compensating against rapid load changes, e.g. with auxiliary current source, with dual mode control or with inductance variation
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • This disclosure relates generally to circuits and methods for controlling operation of switching power converters. More particularly, the present disclosure relates to circuits and methods for controlling operation of a pulse frequency modulated buck DC-to-DC converter to decrease noise coupling and permit a variable current to accommodate large output currents.
  • a buck DC-to-DC converter is a voltage step down and current step up converter.
  • a buck DC-to-DC converter has a power switching section and a low pass filter section. The power switching section reduces the DC component of the power supply voltage source and the filter section removes the high frequency harmonics created by the power switching section to generate the desired DC output voltage level.
  • the power switching section has a first switch with a first terminal connected to one terminal of a power supply voltage source.
  • the power supply voltage source may be a battery or the rectified AC power mains.
  • the second terminal of the first switch is connected to a filter section of the buck DC-to-DC converter.
  • a second switch in the power switching section has a first terminal connected to a ground reference voltage terminal.
  • the second terminal of the second switch is connected to the second terminal of the first switch and the filter section of the buck DC-to-DC converter.
  • the first and second switches each have a control terminal that is connected to control circuitry that determines the switching frequency and duration of the activations of the first and second switches based on a feedback signal from an output of the buck DC-to-DC converter.
  • the input of the filter section is a first terminal of an inductor and the second terminal of the inductor is connected to a first terminal of a filter capacitor.
  • the second terminal of the filter capacitor is connected to the ground reference voltage terminal.
  • the output of the buck DC-to-DC converter is the common connection of the second terminal of the inductor and the first terminal of the filter capacitor.
  • a sense circuit is commonly applied to the output terminal of the buck DC-to-DC converter to provide the feedback signal for the control circuitry.
  • the buck DC-to-DC converter operates in a continuous, synchronous, or pulse width modulated mode for higher current or heavily loaded operation.
  • the first and second switches are activated and deactivated at a fixed frequency and the period between each activation and deactivation is determined by comparing the feedback signal with a desired reference signal to create the desired output voltage.
  • the switches do not supply the current from the power supply voltage source on each cycle and the current then supplied during the commutation mode where current is provided from the collapsing field of the inductor.
  • the discontinuous mode is used in portable electronics such as smart cellular telephones, tablet computers, digital readers, etc. as a “sleep mode”.
  • the only current required by the system in these applications is monitoring current for system maintenance (i.e. system clocking and timers, cellular network monitoring, wireless network monitoring).
  • the buck DC-to-DC converter turns on the first switch to apply the power supply voltage source to the inductor when the output voltage falls below a reference voltage.
  • the first switch is then turned off when the current in the coil reaches a threshold value (sleep current limit).
  • the second switch is turned on when the first switch is turned off.
  • the second switch is then turned off when the current in the coil is fully discharged.
  • the pulse frequency modulation mode is not typically used for large currents as the current limit is normally set low to maximize efficiency.
  • Buck DC-to-DC converter converters operate in the pulse frequency modulation mode have serious problems with noise coupling when operating a high current levels. Further, when the second switch is open, there is no path from the filter section for negative currents resulting from overvoltage situations at the output of the buck DC-to-DC converter.
  • An object of this disclosure is to provide circuits and methods for operating a buck DC-to-DC converter in a pulse frequency mode with variable current limits to provide the ability to manage large output currents.
  • Another object of this disclosure is to provide circuits and methods for operating a buck DC-to-DC converter in hysteretic mode where switching of the power supply voltage source is governed by output current and voltage thresholds.
  • a hysteretic mode control circuit within a DC-to-DC converter.
  • the hysteretic mode control circuit is configured for varying the current limit that controls the switching interval and duration of a power switching section of the DC-to-DC converter to permit the DC-to-DC converter to manage large changes in its output current load.
  • the hysteretic mode control circuit has a positive current limit circuit and a negative current limit circuit.
  • the positive current limit circuit is configured for determining a first reference voltage that is used for controlling activation a first switching device of a switching section of the DC-to-DC converter for transferring current to a load device placed at the output of the DC-to-DC converter.
  • the positive current limit circuit has a first matching switching device having geometry and impurity implantations matching the first switching device.
  • the matching switching device is connected to a first reference current source configured to develop a first reference limit signal for turning on and turning off the first switching device.
  • the first reference limit signal is compared to an output voltage of the power switching section to determine if the first switching device is to be turned on or turned off.
  • the negative current limit circuit is configured for determining a second reference voltage that is used for controlling activation a second switching device of a switching section of the DC-to-DC converter for accepting current from the DC-to-DC converter.
  • the negative current limit circuit has a second matching switching device having geometry and impurity implantations matching the first switching device.
  • the matching switching device is connected to a second reference current source configured to develop a second reference limit signal for turning on and turning off the second switching device.
  • the second reference limit signal is compared to the output voltage of the power switching section to determine if the second switching device is to be turned on or turned off.
  • the positive current section has a dynamic current limit circuit that has a first reference current source providing a maximum reference current to a reference leg of a first current mirror.
  • a mirror leg of the first current mirror is connected to provide a reference limit voltage for an output of the positive current section to determine the switching interval and duration of the first switching device of the power switching section to provide current to the filter section of the DC-to-DC converter.
  • a feedback signal from the output of the DC-to-DC converter and a first reference voltage are inputs to a comparator for determining if the feedback signal is greater than or less than the first reference voltage.
  • An output of the comparator is an input to a switching device that is activated or deactivated to divert current from the reference leg of the current mirror and thus modify the current in mirror leg of the current mirror and thus adjust the voltage level of the reference limit voltage.
  • a load device is connected to the mirror leg of the current mirror for developing the reference limit voltage.
  • the dynamic current limit circuit has a second current source connected in parallel with the mirror leg of the current mirror to provide an optional minimum reference current.
  • hysteretic mode control circuit has a variable current limit circuit.
  • a driver control circuit receives a first control signal developed by a comparison of a feedback signal from the output of the DC-to-DC converter with a reference voltage and a second control signal developed by the variable current limit circuit for controlling activation the first switching device of a switching section of the DC-to-DC converter for transferring current to a load device placed at the output of the DC-to-DC converter.
  • the variable current limit circuit is configured for determining the second control signal by sensing a voltage level present at the input to the low pass filter of the DC-to-DC converter.
  • the voltage level sensing signal is applied to a first terminal of a adjustable differential current source.
  • a control terminal of the differential current source is controlled by the comparison of the feedback signal from the output of the DC-to-DC converter with the reference voltage.
  • the second control signal developed across the adjustable differential current source is applied to the driver control circuit to permit activation of the first and second switching devices according to the level of the necessary voltage across or current through the low pass filter.
  • a compensation current source is connected in parallel with the differential current source.
  • the compensation current source provides a fixed ramp current that is summed with the differential current source for providing compensation to prevent sub-harmonic oscillation.
  • a DC-to-DC converter includes a hysteretic mode control circuit configured for varying the current limit that controls the switching interval and duration of a power switching section of the DC-to-DC converter to permit the DC-to-DC converter to manage large changes in output current load of the DC-to-DC converter.
  • a method for providing hysteretic mode control within a DC-to-DC converter provides the mode control through a hysteretic mode control circuit that varies the current limit that controls the switching interval and duration of a power switching section of the DC-to-DC converter to permit the DC-to-DC converter to manage large changes in output current load of the DC-to-DC converter.
  • the method begins by determining a limit signal proportional to a positive limit current and a negative limit current for the current flowing in the low pass filter of the DC-to-DC converter.
  • the limit signal is a voltage that is compared to a voltage that is developed at the input of the low pass filter of the DC-to-DC converter. If a positive voltage that is developed at the input of the low pass filter of the DC-to-DC converter is greater than a positive limit signal voltage, a first latching circuit is reset and a positive switching device is disabled to prevent current from flowing into the low pass filter.
  • the first latching circuit is not reset and the positive switching device is enabled to allow current to flow into the low pass filter.
  • a second latching circuit is reset and a negative switching device is disabled to prevent current from flowing from the low pass filter.
  • second first latching circuit is not reset and the negative switching device is enabled to allow current to flow from the low pass filter.
  • first setting a maximum reference current and a minimum reference current develop the positive and negative limit signals.
  • a difference between a reference voltage of the DC-to-DC converter and a feedback voltage of the DC-to-DC converter is determined as difference voltage.
  • the difference voltage is converted to a difference current.
  • the difference current is subtracted from the maximum reference current to form a variable limit current.
  • a positive variable limit current is mirrored and converted to the positive limit signal and a negative variable limit current is mirrored and converted to the negative limit signal.
  • FIG. 1 is a schematic of a DC-to-DC converter operating with a pulse width modulation mode and pulse frequency modulation mode.
  • FIG. 2 is a schematic of DC-to-DC converter operating with a pulse width modulation mode and pulse frequency modulation mode embodying the principals of the present disclosure.
  • FIG. 3 is a schematic of a positive and negative current limit circuit incorporated within a DC-to-DC converter embodying the principals of the present disclosure.
  • FIG. 4 is a schematic of an embodiment of a positive dynamic current limit circuit and a negative dynamic current limit circuit incorporated within a DC-to-DC converter of FIG. 2 .
  • FIG. 5 is a schematic of a DC-to-DC converter with a dynamic sleep mode embodying the principals of the present disclosure.
  • FIG. 6 is a plot of the waveforms of the signals the DC-to-DC converter of FIG. 5 with a dynamic sleep mode with a continuous loading.
  • FIG. 7 is a plot of the waveforms of the signals the DC-to-DC converter of FIG. 5 in dynamic sleep mode operation illustrating a change in load current.
  • FIGS. 8 and 9 are flowcharts of a method for providing hysteretic mode control within a DC-to-DC converter embodying the principals of the present disclosure.
  • FIG. 1 is a schematic of a DC-to-DC converter operating with a pulse width modulation mode and pulse frequency modulation mode.
  • the power switching section 120 has a switching control circuit 125 that generates control signals that are applied to a positive input of a driver circuit 130 a and a negative input of a driver circuit 130 b .
  • the output of the driver circuit 130 a is applied to the gate of the PMOS transistor MP 1 and the output of the driver circuit 130 b is applied to the gate of the NMOS transistor MN 1 .
  • the source of the PMOS transistor MP 1 is connected to the power supply voltage source VDD and the source of the NMOS transistor MN 1 is connected to the substrate supply voltage source VSS.
  • the substrate supply voltage source VSS is often the ground reference voltage source, but in some applications is a negative voltage level.
  • the commonly connected drains of the PMOS transistor MP 1 and the NMOS transistor MN 1 are connected to an input terminal of the filter section 135 .
  • the input terminal is a first terminal of an inductor L 1 .
  • the control circuit 125 determines that during the continuous mode or pulse width modulation mode the control signals 116 and 118 are applied to the driver circuit 130 a and the driver circuit 130 b such that the PMOS transistor MP 1 is turned on and the NMOS transistor MN 1 is turned off, a current from the power supply voltage source VDD from the first terminal of the inductor L 1 out the second terminal of the inductor L 1 into the first terminal of the output capacitor C OUT and to the substrate supply voltage source VSS.
  • the output voltage V OUT is present at the junction of the second terminal of the inductor L 1 and the output capacitor C OUT .
  • V L1 the voltage across the inductor L 1 is determined by the formula:
  • V L ⁇ ⁇ 1 L ⁇ d I L d t
  • the output voltage V OUT is equal to the difference of the power supply voltage source VDD and the voltage V L1 across the inductor L 1 in the on state and equal to the negative of the voltage ⁇ V L1 across the inductor L 1 in the off state.
  • the duty cycle of the buck DC-to-DC converter determines the on state time and the off state time. It can be shown that the output voltage V OUT is equal to the duty cycle of the buck DC-to-DC converter multiplied by the voltage level of the power supply voltage source VDD.
  • the feedback stage 140 has three inputs.
  • the first input 107 is the feedback voltage V FB that is developed from the output voltage V OUT at common connection of the second terminal of the inductor L 1 and the first terminal of the output capacitor C OUT .
  • the second and third inputs are the first and second reference voltages V REF1 and V REF2 generated by the switch control circuit 105 .
  • the switch control circuit 105 has a digital-to-analog converter 110 that receives a reference control word 112 and an offset control word 114 .
  • the digital-to-analog converter 110 converts the reference control word 112 to the first reference voltage V REF1 and the offset control word 114 to the second reference voltage V REF2 .
  • the first reference voltage V REF1 and the second reference voltage V REF2 are the second and third inputs to the feedback stage 115 .
  • the feedback control stage 115 has a first comparator 117 for providing a first control signal 116 and a second comparator 119 for providing a second control signal 118 .
  • the first control signal 116 is applied to a data input D of a first data flip-flop 127 of the switching control circuit 125 .
  • the second control signal 118 is applied to a data input D of a second data flip-flop 129 of the switching control circuit 125 .
  • An oscillator 150 generates the clock pulse signal 152 that is applied to the clock CK input of the first data flip-flop 127 and the second data flip-flop 129 .
  • the output of the first data flip-flop 127 is applied to the input of the positive driver circuit 130 a and the output of the second data flip-flop 129 is applied to the input of the negative driver circuit 130 b .
  • the output of the first driver circuit 130 a is applied to the gate of the PMOS transistor MP 1 and the output of the second driver circuit 130 b is applied to the gate of the NMOS transistor MN 1 .
  • a current/voltage sense circuit 140 is placed at the junction of the second terminal of the inductor L 1 and the output capacitor C OUT .
  • the current/voltage sense circuit 140 determines the feedback voltage V FB and the feedback current I FB .
  • the feedback voltage V FB is fed back as the first input to the first comparator 117 and the second comparator 118 .
  • the feedback current I FB is fed back to a pulse width modulation/pulse frequency modulation control circuit 145 .
  • the pulse width modulation/pulse frequency modulation control circuit 145 resets the first data flip-flop 127 and the second data flip-flop 129 .
  • the first data flip-flop 127 and the second data flip-flop 129 are not reset and the data outputs of the first data flip-flop 127 and the second data flip-flop 129 are controlled by the feedback voltage V FB to determine the pulse width of the control signals driving the first driver circuit 130 a and the second driver circuit 130 b and thus the PMOS transistor MP 1 and the NMOS transistor MN 1 .
  • the DC-to-DC converter When the load current I OUT decreases to a predetermined level, the DC-to-DC converter is set to the pulse frequency modulation mode.
  • the feedback current I FB is interpreted by the pulse width modulation/pulse frequency modulation control circuit 145 such that the second data flip-flop 129 is reset. This turns off the NMOS transistor MN 1 .
  • the pulse width modulation/pulse frequency modulation control circuit 145 resets the first data flip-flop 127 and the PMOS transistor MP 1 is turned off.
  • the pulse width modulation/pulse frequency modulation control circuit 145 releases the reset of the first data flip-flop 127 to activate the PMOS transistor MP 1 to allow current to flow into the filter section 135 .
  • the NMOS transistor MN 1 Since the NMOS transistor MN 1 is turned off, the DC-to-DC converter can not accept current from the load, therefore any overvoltage of the output voltage V OUT is not discharged. Further, the PMOS transistor MP 1 is turned on asynchronously at high load creating serious implications for noise being coupled to the system that is load being powered.
  • FIG. 2 is a schematic of a DC-to-DC converter operating with pulse width modulation mode and pulse frequency modulation mode embodying the principals of the present disclosure.
  • a switch control circuit 205 has a digital-to-analog converter 210 that receives a reference control word 212 and an offset control word 214 .
  • the digital-to-analog converter 210 converts the reference control word 112 to the first reference voltage V REF1 and converts the offset control word 114 to the second reference voltage V REF2 .
  • the first reference voltage V REF1 and the second reference voltage V REF2 are two of the three inputs to the feedback stage 215 .
  • the feedback stage 215 has a first comparator 217 that receives the first reference voltage V REF1 and a second comparator 219 that receives the second reference voltage V REF2 .
  • the third 207 of the three inputs to the feedback stage 215 receives the feedback voltage V FB that is compared to the first reference voltage V REF1 and the second reference voltage V REF2 .
  • the feedback voltage V FB that is developed from the output voltage V OUT at common connection of the second terminal of the inductor L 1 and the first terminal of the output capacitor C OUT of the filter section 235 .
  • the outputs 216 and 218 of the first comparator 217 and a second comparator 219 are the inputs to the switching control circuit 225 .
  • the first comparator 217 provides the first control signal 216 is applied to a data input D of a first data flip-flop 227 of the switching control circuit 225 .
  • the second comparator 219 provides the second control signal 218 that is applied to a data input D of a second data flip-flop 229 of the switching control circuit 225 .
  • An oscillator 250 generates the clock pulse signal 252 that is applied to the clock CK input of the first data flip-flop 227 and the second data flip-flop 229 .
  • the output of the first data flip-flop 227 is applied to the input of the positive driver circuit 230 a and the output of the second data flip-flop 229 is applied to the input of the negative driver circuit 230 b .
  • the output of the first driver circuit 230 a is applied to the gate of the PMOS transistor MP 1 and the output of the second driver circuit 230 b is applied to the gate of the NMOS transistor MN 1 .
  • the source of the PMOS transistor MP 1 is connected to the power supply voltage source VDD and the source of the NMOS transistor MN 1 is connected to the substrate supply voltage source VSS.
  • the substrate supply voltage source VSS is often the ground reference voltage source, but in some applications is a negative voltage level.
  • the commonly connected drains of the PMOS transistor MP 1 and the NMOS transistor MN 1 are connected to an input terminal of the filter section 235 .
  • the input terminal is a first terminal of an inductor L 1 .
  • the control signals 216 and 218 are applied to the driver circuit 230 a and the driver circuit 230 b such that the PMOS transistor MP 1 is turned on and the NMOS transistor MN 1 is turned off, a current from the power supply voltage source VDD from the first terminal of the inductor L 1 out the second terminal of the inductor L 1 into the first terminal of the output capacitor C OUT and to the substrate supply voltage source VSS.
  • the output voltage V OUT is present at the junction of the second terminal of the inductor L 1 and the output capacitor C OUT .
  • a first current limit circuit is designated as a positive current limit circuit 240 a that is connected in proximity with the PMOS transistor MP 1 and the negative current limit circuit 240 b is connected in proximity with the NMOS transistor MN 1 .
  • the positive current limit circuit 240 a and the negative current limit circuit 240 b are connected between the power supply voltage source VDD and the substrate supply voltage source VSS.
  • the output of the driver circuit 230 a is connected to the positive current limit circuit 240 a and the driver circuit 230 b is connected to the negative current limit circuit 240 b .
  • the output of the positive current limit circuit 240 a is a first reference limit signal designated as a positive reference limit voltage 242 and the output of the negative current limit circuit 240 b is a second reference limit signal designated as a negative reference limit voltage 244 .
  • the positive reference limit voltage 242 and the negative reference limit voltage 244 are the inputs to the pulse width modulation/pulse frequency modulation control circuit 245 .
  • the pulse width modulation/pulse frequency modulation control circuit 245 compares the positive reference limit voltage 242 with the voltage V LX developed at the first terminal of an inductor L 1 for selectively resetting of the first data flip-flop 227 to control operation of PMOS transistor MP 1 .
  • the pulse width modulation/pulse frequency modulation control circuit 245 compares the negative reference limit voltage 244 with the voltage V LX developed at the first terminal of an inductor L 1 for selectively resetting of the second data flip-flop 229 to control operation of the NMOS transistor MN 1 .
  • FIG. 3 is a schematic of the positive current limit circuit 240 a and the negative current limit circuit 240 b incorporated within a DC-to-DC converter embodying the principals of the present disclosure.
  • the positive current limit circuit 240 a has a PMOS transistor MP 2 that is a dummy transistor having characteristic that are matched to the geometry and impurity implantations of the PMOS transistor MP 1 .
  • the PMOS transistor MP 2 is used to generate the reference voltage V LIM+ for the current limit of the current passing through the PMOS transistor MP 1 across the positive reference current source I 1 .
  • the PMOS transistor MP 2 has a gate connected to the output of the positive driver circuit 230 a .
  • the source of the PMOS transistor MP 2 is connected to the power supply voltage source VDD and the drain of the PMOS transistor MP 2 is connected to a first terminal of the positive reference current source I 1 .
  • the second terminal of the positive reference current source I 1 is connected to the substrate supply voltage source VSS.
  • the negative current limit circuit 240 a has an NMOS transistor MN 2 that is a dummy transistor characteristic that are matched to the geometry and impurity implantations of the NMOS transistor MN 1 .
  • the NMOS transistor MN 2 is used to generate the reference voltage V LIM ⁇ for the current limit of the current passing through the NMOS transistor MN 1 across the negative reference current source I 2 .
  • the first terminal of the negative reference current source I 2 is connected to the power supply voltage source VDD.
  • the drain of the NMOS transistor MN 2 is connected to a second terminal of the negative reference current source I 2 and the source of the NMOS transistor MN 2 is connected to the substrate supply voltage source VSS.
  • the NMOS transistor MN 2 has a gate connected to the output of the negative driver circuit 230 b.
  • junction 242 of the PMOS transistor MP 2 and the first terminal of the first current source I 1 is connected to a first terminal of the third comparator 247 of the pulse width modulation/pulse frequency modulation control circuit 245 .
  • junction 244 of the NMOS transistor MN 2 and the second terminal of the second current source I 2 is connected to a first terminal of the fourth comparator 249 of the pulse width modulation/pulse frequency modulation control circuit 245 .
  • the second terminals of the third comparator 247 and fourth comparator 249 are connected to the connection of the drains of the PMOS transistor MP 1 and the NMOS transistor MN 1 with the first terminal of the inductor L 1 of FIG. 2 .
  • the reference voltage V LIM+ and reference voltage V LIM ⁇ are compared with the voltage V LX developed at the connection of the drains of the PMOS transistor MP 1 and the NMOS transistor MN 1 with the first terminal of the inductor L 1 during the corresponding part of the duty cycle to determine when the output current is too high.
  • the results of the comparison of the reference voltage V LIM+ and reference voltage V LIM ⁇ with the voltage V LX are applied to the control logic circuit 246 to generate the reset signals V RST+ and V RST ⁇ that are transferred respectively to the reset terminals R of the first data flip-flop 227 and the second data flip-flop 229 .
  • the output of the first driver circuit 230 a controls the gate of the PMOS transistor MP 1 and the output of the second driver circuit 230 b controls the gate of the NMOS transistor MN 1 .
  • the input of the driver circuit 230 a being controlled by the output of the first data flip-flop 227 and the input of the driver circuit 230 b being controlled by the second data flip-flop 229 .
  • the output of the first data flip-flop 227 and the output of the second data flip-flop 229 turn on the corresponding PMOS transistor MP 1 or NMOS transistor MN 1 .
  • the activated PMOS transistor MP 1 or NMOS transistor MN 1 is then turned off by the corresponding reset signals V RST+ and V RST ⁇ .
  • control logic circuit 246 has circuitry that will permit the reset signals V RST+ and V RST ⁇ to turn on the either the activated PMOS transistor MP 1 or NMOS transistor MN 1 , as required.
  • the second comparator 219 is offset by approximately 10 mV as determined by the second reference voltage V ref2 to allow a small range of output voltages for which the PMOS transistor MP 1 or NMOS transistor are not switched to provide a saving in power for very low loads.
  • FIG. 4 is a schematic of an embodiment of a positive dynamic current limit circuit 240 a and a negative dynamic current limit circuit 240 b incorporated within a DC-to-DC converter of FIG. 2 .
  • a first differential amplifier 241 receives the first reference voltage V REF1 and the feedback voltage V FB to be compared.
  • the output of the differential amplifier 241 is connected to a gate of a first NMOS switching transistor N 1 .
  • the drain of the first NMOS switching transistor N 1 is connected to a first terminal of a third reference current source I 3 that sources a maximum positive reference current I MAX+ .
  • a second terminal of the third current source I 3 is connected to the power supply voltage source VDD.
  • the NMOS transistors N 2 and N 3 form a first limit current mirror designated as a positive limit current mirror.
  • the NMOS transistor N 2 is a diode-connected transistor that forms the reference leg of the positive limit current mirror and has its gate and drain commonly connected to the first terminal of the third current source I 3 .
  • the source of the NMOS transistor N 2 is connected to the substrate supply voltage source VSS.
  • the NMOS transistor N 3 forms the mirror leg of the positive limit current source and has its gate connected to the commonly connected gate and drain of the NMOS transistor N 2 of the reference leg.
  • the drain of the NMOS transistor N 3 is connected to the drain of a dummy PMOS transistor P 1 .
  • the source of the dummy PMOS transistor P 1 is connected to the power supply voltage source VDD and the gate of the dummy PMOS transistor P 1 is connected to the substrate supply voltage source VSS.
  • the dummy PMOS transistor P 1 is matched to the geometry and impurity implantations of the PMOS transistor MP 1 of FIG. 2 .
  • the first differential amplifier 241 compares the first reference voltage V REF1 and the feedback voltage V FB voltage.
  • the voltage level of the output of the differential amplifier 241 causes the NMOS transistor N 1 to steal current from the third current source I 3 that sources a maximum positive reference current I MAX+ .
  • the maximum positive reference current I MAX+ sets the maximum current limit possible.
  • the current mirror formed by the NMOS transistors N 2 and N 3 then mirrors the remaining current through the dummy PMOS transistor P 1 .
  • a fourth reference current source I 4 provides a fixed minimum current that may optionally be necessary.
  • the output reference voltage V LIM+ is applied to the input of the pulse width modulation/pulse frequency modulation control circuit 245 .
  • a second differential amplifier 242 receives the second reference voltage V REF2 and the feedback voltage V FB to be compared.
  • the output of the differential amplifier 243 is connected to a gate of a second PMOS switching transistor P 2 .
  • the drain of the second PMOS switching transistor P 2 is connected to a first terminal of a fifth current source I 5 that sources a maximum negative reference current I MAX ⁇ .
  • a second terminal of the fifth reference current source I 5 is connected to the substrate supply voltage source VSS.
  • the PMOS transistors P 3 and P 4 form a second limit current mirror designated as a negative limit current mirror.
  • the PMOS transistor P 3 is a diode-connected transistor that forms the reference leg of the positive limit current mirror and has its gate and drain commonly connected to the first terminal of the fifth current source I 5 .
  • the source of the PMOS transistor P 3 is connected to the power supply voltage source VDD.
  • the PMOS transistor P 4 forms the mirror leg of the negative limit current source and has its gate connected to the commonly connected gate and drain of the PMOS transistor P 3 of the reference leg.
  • the drain of the PMOS transistor P 4 is connected to the drain of a dummy NMOS transistor N 4 .
  • the source of the dummy NMOS transistor N 4 is connected to the substrate supply voltage source VSS and the gate of the dummy PMOS transistor P 1 is connected to the power supply voltage source VDD.
  • the dummy NMOS transistor N 4 is matched to the geometry and impurity implantations of the NMOS transistor MN 1 of FIG. 2 .
  • the second differential amplifier 243 compares the second reference voltage V REF2 and the feedback voltage V FB voltage.
  • the voltage level of the output of the second differential amplifier 243 causes the PMOS transistor P 2 to steal current from the fifth current source I 5 that sinks a maximum negative reference current I MAX ⁇ .
  • the maximum negative reference current I MAX ⁇ sets the maximum negative current limit possible.
  • the current mirror formed by the PMOS transistors P 3 and P 4 then mirrors the remaining current through the dummy NMOS transistor N 4 .
  • a sixth reference current source I 6 provides a fixed minimum current that may optionally be necessary.
  • the output reference voltage V LIM ⁇ is applied to the input of the pulse width modulation/pulse frequency modulation control circuit 245 .
  • the reference voltage V LIM+ and reference voltage V LIM ⁇ are compared in the comparators 247 and 249 with the voltage V LX developed at the connection of the drains of the PMOS transistor MP 1 and the NMOS transistor MN 1 with the first terminal of the inductor L 1 during the corresponding part of the duty cycle to determine when the output current is too high.
  • the results of the comparison of the reference voltage V LIM+ and reference voltage V LIM ⁇ with the voltage V LX are applied to the control logic circuit 246 to generate the reset signals V RST+ and V RST ⁇ that are transferred respectively to the reset terminals R of the first data flip-flop 227 and the second data flip-flop 229 .
  • the output of the first driver circuit 230 a controls the gate of the PMOS transistor MP 1 and the output of the second driver circuit 230 b controls the gate of the NMOS transistor MN 1 .
  • the input of the driver circuit 230 a is controlled by the output of the first data flip-flop 227 and the input of the driver circuit 230 b is controlled by the second data flip-flop 229 .
  • the output of the first data flip-flop 227 and the output of the second data flip-flop 229 turn on the corresponding PMOS transistor MP 1 or NMOS transistor MN 1 .
  • the activated PMOS transistor MP 1 or NMOS transistor MN 1 is then turned off by the corresponding reset signals V RST+ and V RST ⁇ .
  • control logic circuit 246 has circuitry that will permit the reset signals V RST+ and V RST ⁇ to turn on the either the PMOS transistor MP 1 or the NMOS transistor MN 1 , as required.
  • the second comparator 219 is offset by approximately 10 mV as determined by the second reference voltage V ref2 to allow a small range of output voltages for which the PMOS transistor MP 1 or NMOS transistor MN 1 are not switched to provide a saving in power for very low loads.
  • FIG. 5 is a schematic of a DC-to-DC converter with a dynamic sleep mode embodying the principals of the present disclosure.
  • the switch control circuit 305 has a digital-to-analog converter 310 that receives a reference control word 314 .
  • the digital-to-analog converter 310 converts the reference control word 314 to the reference voltage V REF .
  • the reference voltage V REF is one of the two inputs to the feedback stage 315 .
  • the feedback stage 315 has a first comparator 317 and a differential amplifier 319 that receive the reference voltage V REF .
  • the second input 307 of the feedback stage 315 receives the feedback voltage V FB that is compared to the reference voltage V REF .
  • the feedback voltage V FB is developed from the output voltage V OUT at common connection of the second terminal of the inductor L 1 and the first terminal of the output capacitor C OUT of the filter section 235 .
  • the output 316 of the first comparator 317 is the first input to the switching control circuit 325 .
  • the first comparator 317 provides the positive control signal V UNDER+ that is applied to a data input of a first data flip-flop 326 of the switching control circuit 325 .
  • the output 318 of the differential amplifier 319 provides a current control signal that is applied to a differential current source I 7 .
  • the differential current source I 7 develops a reference limit current I LIM .
  • the sense circuit 340 provides a load for the differential current source I 7 and thus the current limit flag signal V ILIMFLG .
  • the current control signal from the output 318 of the differential amplifier 319 adjusts the differential current source I 7 such that the voltage levels are such that the data input of the second data flip-flop 328 of the switching control circuit 325 are at the correct logic levels for controlling the switching of the NMOS transistor MN 1 .
  • An oscillator 350 generates the clock pulse signal 352 that is applied to the clock CK input of the first data flip-flop 326 and the second data flip-flop 328 .
  • the output of the first data flip-flop 326 is applied to the input of the positive driver circuit 330 a and the output of the second data flip-flop 329 is applied to the input of the negative driver circuit 330 b .
  • the output of the first driver circuit 330 a is applied to the gate of the PMOS transistor MP 1 and the output of the second driver circuit 330 b is applied to the gate of the NMOS transistor MN 1 .
  • the source of the PMOS transistor MP 1 is connected to the power supply voltage source VDD and the source of the NMOS transistor MN 1 is connected to the substrate supply voltage source VSS.
  • the substrate supply voltage source VSS is often the ground reference voltage source, but in some applications is a negative voltage level.
  • the commonly connected drains of the PMOS transistor MP 1 and the NMOS transistor MN 1 are connected to an input terminal of the filter section 235 .
  • the input terminal is a first terminal of an inductor L 1 .
  • the data output of the first data flip-flop 326 is applied to the driver circuit 330 a and the data output of the second data flip-flop 327 is applied to the driver circuit 330 b .
  • the reference limit current I LIM from the differential current source I 7 is allowed to vary and thus enable the DC-to-DC converter to support very high loads.
  • the sense circuit 340 senses the output current I OUT when the PMOS transistor MP 1 is turned on.
  • the sense circuit 340 generates a sense current I SENSE produces a current proportional to the current in the PMOS transistor MP 1 .
  • the sense current I SENSE is compared to the reference limit current I LIM . When the sense current I SENSE from sense circuit 340 is greater than the reference limit current I LIM then the voltage on current limit flag signal V ILIMFLG assumes a first logic level (1). When the sense current I SENSE from sense circuit 340 is less than the reference limit current I LIM then the voltage on current limit flag signal V ILIMFLG assumes a second logic level (0).
  • the current limit flag signal V ILIMFLG indicates that current limit is achieved and the control circuit 328 generates the data applied to the input of the first flip-flop 326 that will force the first driver circuit 330 a to turnoff the PMOS transistor MP 1 .
  • the reference limit current I LIM is modulated by the output voltage of the differential amplifier 319 as a result of the comparison of the feedback voltage V FB with the reference voltage V REF . As the feedback voltage V FB falls below the reference voltage V REF , the reference current is increased, and so the current limit value is also increased.
  • FIG. 6 is a plot of the waveforms of the signals the DC-to-DC converter in dynamic sleep mode operation of FIG. 5 .
  • the feedback voltage V FB falls below the reference voltage V REF from digital-to-analog converter 310
  • the output of the first comparator 317 rises from the second level (0) to the first level (1) at the time ⁇ 0 to activate the positive control signal V UNDER+ .
  • the first level (1) is applied through the control circuit 328 to the first data flip-flop 326 .
  • the data present at the data input D of the first data flip-flop 326 is not transferred to the output of the first data flip-flop 326 until the rising edge of the clock CLK at the time ⁇ 1 at which time the gate of the PMOS transistor MP 1 is brought to the first level (0) to turn on the PMOS transistor MP 1 such that current I COIL passes to the first terminal of the inductor L 1 .
  • the current I COIL through the inductor L 1 rises from the time ⁇ 1 to the time ⁇ 2 when the current I COIL reaches the magnitude of the reference limit current I LIM .
  • the voltage of the current limit flag signal V ILIMFLG developed at the top of the differential current source I 7 is applied to the control circuit 328 and the control circuit 328 sets the data at the data inputs D of the first and second data flip-flops 326 and 327 such that the PMOS transistor MP 1 and the NMOS transistor MN 1 are turned off. With the PMOS transistor MP 1 and the NMOS transistor MN 1 are turned off, the current I COIL falls toward zero and the DC-to-DC converter operates in the discontinuous mode.
  • the voltage of the current limit flag signal I LIMFLG is deactivated to essentially the first level (0) at the time ⁇ 1 .
  • the limit current I LIM is controlled by the sum of two reference currents.
  • the first reference current is provided by the adjustable differential current source I 7 , as described above.
  • the second reference current is fixed ramp current source I 8 .
  • the reference limit current I LIM of the adjustable differential current source I 7 and the compensation current I COMP of the fixed ramp current source I 8 are additively combined to provide a degree of compensation to prevent sub-harmonic oscillation.
  • the ramp waveform of the compensation current I COMP should start at a high value and have a negative slope.
  • the compensation current I COMP has a negative value thus subtracting current from differential current source I 7 .
  • FIG. 7 is a plot of the waveforms of the signals the DC-to-DC converter of FIG. 5 in dynamic sleep mode operation illustrating a change in output current I OUT to the external load circuit.
  • the output current I OUT increases at the time ⁇ 0
  • the feedback voltage V FB starts to fall until it is below the below the reference voltage V REF from digital-to-analog converter 310
  • the output of the first comparator 317 rises at the time ⁇ 1 from the second logic level (0) to the first logic level (1) to activate the positive control signal V UNDER+ .
  • the first logic level (1) is applied through the control circuit 328 to the first data flip-flop 326 .
  • the data present at the data input D of the first data flip-flop 326 is not transferred to the output of the first data flip-flop 326 until the rising edge of the clock CLK at the time ⁇ 2 at which time the gate of the PMOS transistor MP 1 is brought to the first level (0) to turn on the PMOS transistor MP 1 such that current I COIL passes to the first terminal of the inductor L 1 .
  • the current I COIL through the inductor L 1 rises from the time ⁇ 2 to the time ⁇ 3 when the coil current I COIL reaches the magnitude of the reference limit current I LIM1 .
  • the voltage of the current limit flag signal V ILIMFLG developed at the top of the differential current source I 7 is applied at the time ⁇ 3 to the control circuit 328 and the control circuit 328 sets the data at the data inputs D of the first and second data flip-flops 326 and 327 such that the PMOS transistor MP 1 and the NMOS transistor MN 1 are turned off.
  • the current I COIL falls toward zero from the time ⁇ 3 to the time ⁇ 4 and the DC-to-DC converter operates in the discontinuous mode.
  • the voltage of the current limit flag signal V ILIMFLG is deactivated to essentially the first level (0) shortly after the time ⁇ 3 .
  • the feedback voltage V FB has also decreased from the time ⁇ 3 to the time ⁇ 4 and remains less than the reference voltage V REF such that at the time ⁇ 4 , the gate of the PMOS transistor MP 1 is activated with the rising edge of the clock pulse CLK.
  • the coil current I COIL begins to rise between the time ⁇ 4 and the time ⁇ 5 .
  • the first reference limit current I LIM1 from the differential current source I 7 is adjusted by the differential voltage ⁇ V from the differential amplifier 319 to a second reference limit current I LIM2 .
  • the current limit flag signal V ILIMFLG is activated shortly before the time ⁇ 5 .
  • the gate of the PMOS transistor MP 1 is set to the first logic level (1) and the PMOS transistor MP 1 is turned off at the time at the time ⁇ 5 .
  • the voltage of the current limit flag signal V ILIMFLG is deactivated to essentially the second level (0) shortly after the time ⁇ 5 .
  • the coil current I COIL falls between the time ⁇ 5 and the time ⁇ 6 such that the feedback voltage V FB is decreasing from the time ⁇ 5 to the time ⁇ 6 and remains less than the reference voltage V REF such that at the time ⁇ 6 gate of the PMOS transistor MP 1 is activated with the rising edge of the clock pulse CLK.
  • the cyclic operation from the time ⁇ 4 to the time ⁇ 12 is equivalent to that described between the time ⁇ 4 and the time ⁇ 6 .
  • This process continues until the output current I OUT decreases the normal dynamic sleep mode or the system commands the DC-to-DC converter to resume the normal continuous operation mode.
  • Varying between the first reference limit current I LIM1 and the second reference limit current I LIM2 controls an interval and duration of the time at which the switching section 320 is switched to permit the DC-to-DC converter to manage large changes in an output current load I OUT , while operating in the discontinuous operation mode.
  • FIG. 8 is flowchart of a method for providing hysteretic mode control within a DC-to-DC converter 200 of FIG. 2 .
  • the most positive voltage limit V ILM+ representing a maximum current I LIM+ to pass through the inductor L 1 of the low pass filter section 235 is determined (Box 400 ).
  • the most negative voltage limit V ILM ⁇ representing a minimum current I LIM ⁇ to pass through the inductor L 1 of the low pass filter section 235 is determined (Box 405 ).
  • the voltage V LX present at the junction of the drains of the PMOS transistor MP 1 and the NMOS transistor MN 1 with the first terminal of the inductor L 1 is compared (Box 405 ) with the positive voltage limit V ILM+ .
  • the first data flip-flop 227 is reset (Box 405 ) and the PMOS transistor MP 1 is turned off.
  • the voltage V LX is compared (Box 430 ) with the negative voltage limit V ILM ⁇ . If the voltage V LX is not greater than or equal to the negative voltage limit V ILM ⁇ , the NMOS transistor MN 1 is turned on (Box 445 ).
  • the operational mode of the DC-to-DC converter 200 is queried (Box 450 ) to determine if the dynamic sleep mode is to be terminated and the normal continuous operational mode resumed.
  • the process then returns to the comparing (Box 405 ) of the voltage V LX with the positive voltage limit V ILM+ . If, this instance, the voltage V LX is not greater than or equal to the positive voltage limit V ILM+ , the PMOS transistor MP 1 is turned on. The voltage V LX is compared (Box 430 ) with the negative voltage limit V ILM ⁇ . If the voltage V LX is greater than or equal to the negative voltage limit V ILM ⁇ , the data flip-flop 229 is reset (Box 435 ) and the NMOS transistor MN 1 is turned off (Box 440 ).
  • the operational mode of the DC-to-DC converter 200 is again queried (Box 450 ) to determine if the dynamic sleep mode is to be terminated and the normal continuous operational mode resumed. If the normal continuous operational mode is to be resumed the process is ended. If the dynamic sleep mode is to be continued, the process as above described, continues.
  • FIG. 9 is flowchart of a method for providing hysteretic mode control within a DC-to-DC converter 300 of FIG. 5 .
  • the DC-to-DC converter 300 is set to begin (Box 500 ) the dynamic sleep mode, the voltage reference V ref and the feedback voltage V FB are compared (Box 505 ) to determine the positive control signal V UNDER+ .
  • the differential ⁇ V between the voltage reference V ref and the feedback voltage V FB is amplified (Box 510 ) and then converted (Box 515 ) to a differential current ⁇ V.
  • the limiting current I LIM is set (Box 520 ) as the subtractive combination of the sense current I SENSE and the differential current ⁇ V.
  • the limiting current I LIM determines (Box 525 ) a current limit flag signal V ILIMFLG .
  • the state of the voltage to be applied to the gate of the PMOS transistor MP 1 and the gate of the NMOS transistor MN 1 are determined (Box 530 ).
  • the states of the voltage applied to the gate of the PMOS transistor MP 1 and the gate of the NMOS transistor MN 1 may be such that either the PMOS transistor MP 1 is turned on and the NMOS transistor MN 1 is turned off, or the PMOS transistor MP 1 is turned off and the NMOS transistor MN 1 is turned on, or the PMOS transistor MP 1 is turned off and the NMOS transistor MN 1 is turned off.
  • the state of the gate of the PMOS transistor MP 1 is examined (Box 535 ) to determine if the PMOS transistor MP 1 is turned on.
  • the NMOS transistor MN 1 is turned off (Box 540 ) and the PMOS transistor MP 1 is turned on (Box 545 ), if the state of the gate of the PMOS transistor MP 1 indicates the PMOS transistor MP 1 is to be turned on. If the state of the gate of the PMOS transistor MP 1 indicates the PMOS transistor MP 1 is to be turned off, the PMOS transistor MP 1 is turned off (Box 545 ) and the state of the gate of the NMOS transistor MN 1 is examined (Box 555 ) to determine if the NMOS transistor MN 1 is turned on.
  • the NMOS transistor MN 1 is turned on (Box 560 ), if the state of the gate of the NMOS transistor MN 1 indicates the NMOS transistor MN 1 is to be turned on. If the state of the gate of the NMOS transistor MN 1 indicates the NMOS transistor MN 1 is to be turned off, the NMOS transistor MN 1 is turned off (Box 565 ). If the PMOS transistor MP 1 is turned on (Box 545 ) or the NMOS transistor MN 1 is turned on (Box 560 ), or the PMOS transistor MP 1 is turned off (Box 550 ) and NMOS transistor MN 1 is turned off (Box 565 ), the sleep mode is examined (Box 570 ) to determine if the continuous mode is to be resumed. If the sleep mode is to be continue, the next cycle is started 500 and the process as described above is repeated. If the sleep mode is to be discontinued, the continuous mode is resumed and the dynamic sleep mode is ended.

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