WO2012120788A1 - Dispositif de commande de correcteur du facteur de puissance de convertisseur survolteur - Google Patents

Dispositif de commande de correcteur du facteur de puissance de convertisseur survolteur Download PDF

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Publication number
WO2012120788A1
WO2012120788A1 PCT/JP2012/000950 JP2012000950W WO2012120788A1 WO 2012120788 A1 WO2012120788 A1 WO 2012120788A1 JP 2012000950 W JP2012000950 W JP 2012000950W WO 2012120788 A1 WO2012120788 A1 WO 2012120788A1
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Prior art keywords
voltage
switching element
terminal
current
source
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English (en)
Japanese (ja)
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修二 玉岡
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Panasonic Corp
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Panasonic Corp
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    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
    • H02M1/4208—Arrangements for improving power factor of AC input
    • H02M1/4233—Arrangements for improving power factor of AC input using a bridge converter comprising active switches
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0083—Converters characterised by their input or output configuration
    • H02M1/0085—Partially controlled bridges
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02P—CLIMATE CHANGE MITIGATION TECHNOLOGIES IN THE PRODUCTION OR PROCESSING OF GOODS
    • Y02P80/00—Climate change mitigation technologies for sector-wide applications
    • Y02P80/10—Efficient use of energy, e.g. using compressed air or pressurized fluid as energy carrier

Definitions

  • the present invention relates to a high-input power factor and high-efficiency step-up PFC (Power Factor Correction) controller that rectifies an input AC voltage and outputs a DC voltage larger than the peak value of the AC voltage.
  • PFC Power Factor Correction
  • FIG. 13 is a circuit block diagram of a step-up PFC control device without a rectifier diode bridge described in Patent Document 1.
  • the step-up PFC control device 500 without a rectifier diode bridge described in FIG. 13 is intended to reduce the size and cost of the high-frequency block filter 502.
  • two sawtooth wave oscillators 514 and 515 having phases different from each other by 180 degrees are used, and the phase of the switching operation for boosting is shifted twice by 180 degrees. That is, the first switching operation by the switching elements T1 and T2 and the second switching operation by the switching elements T3 and T4. As a result, the ripple of the current flowing through the high-frequency blocking filter 502 has a second harmonic compared with the switching frequency. As a result, the high-frequency blocking filter 502 can be reduced in size and cost.
  • the above-described conventional step-up PFC control device 500 generates a loss due to a voltage drop due to the forward voltage VF of the diode D1 or D2 and the forward voltage VFP of the parasitic diode of the switching elements T1 to T4 made of silicon semiconductor. Have the problem of doing.
  • the switching loss during the turn-on operation of the switching elements T1 to T4 is large due to the recovery current of the parasitic diodes of the switching elements T1 to T4 made of silicon-based semiconductor. Further, the switching loss increases in proportion to the switching frequency of the PFC control device. Therefore, increasing the switching frequency in order to reduce the size of the magnetic induction means 504 causes an increase in switching loss.
  • the present invention provides a step-up PFC control device that realizes low power consumption and miniaturization by eliminating the parasitic diodes of the switching elements and the diodes D1 and D2 that occupy most of the power consumption in the PFC control device.
  • a step-up PFC control device is a step-up PFC control device including an AC / DC converter unit that rectifies and boosts an input single-phase AC voltage.
  • the AC / DC converter unit includes a plurality of AC switch units that are bridge-connected, a boosting coil, a smoothing capacitor, and a drive logic circuit.
  • the boosting coil stores magnetic energy corresponding to the bidirectional current that flows in response to the application of the single-phase AC voltage.
  • the smoothing capacitor stores a boosting coil and a charge corresponding to the magnetic energy stored in the boosting coil.
  • the drive logic circuit switches the current path of bidirectional current by controlling the on / off state of a plurality of AC switch units, and alternately stores magnetic energy in the boosting coil and charges in the smoothing capacitor. Make it.
  • Each of the plurality of AC switch units has a gate terminal, a drain terminal, and a source terminal.
  • the AC switch unit is composed of a switching element having FET characteristics, reverse FET characteristics, and reverse conduction characteristics.
  • the FET characteristic is a characteristic in which current flows from the drain terminal to the source terminal according to the polarity of the drain-source voltage when the gate-source voltage, which is the gate voltage with respect to the source voltage, is higher than the threshold voltage.
  • the reverse FET characteristic is a characteristic in which current flows from the source terminal to the drain terminal according to the polarity of the drain-source voltage when the gate-source voltage is higher than the threshold voltage.
  • Reverse conduction characteristics means that when the gate-source voltage is equal to or lower than the threshold voltage (hereinafter referred to as “below”), the current flowing from the drain terminal to the source terminal is cut off, and the gate voltage relative to the drain voltage is This is a characteristic in which a current flows from the source terminal to the drain terminal when it is equal to or higher than the threshold voltage (hereinafter referred to as “more”).
  • the plurality of AC switch units have first to fourth switching elements.
  • the first electrode of the boosting coil is connected to a first AC terminal to which a single-phase AC voltage is applied.
  • the drain terminal of the first switching element and the drain terminal of the third switching element are connected to the first electrode of the smoothing capacitor.
  • the source terminal of the second switching element and the source terminal of the fourth switching element are connected to the second electrode of the smoothing capacitor.
  • the source terminal of the first switching element and the drain terminal of the second switching element are connected to the second electrode of the boosting coil.
  • the source terminal of the third switching element and the drain terminal of the fourth switching element are connected to a second AC terminal to which a single-phase AC voltage is applied.
  • the gate terminals of the first to fourth switching elements are connected to the respective predrive circuits that shift the level of the control signal output from the drive logic circuit.
  • the drive logic circuit controls so that the first switching element and the second switching element are not turned on at the same time, or the third switching element and the fourth switching element are not turned on at the same time. May be.
  • the AC / DC converter unit is constituted by a basic bridge circuit in which four bidirectional switching elements are bridge-connected. Therefore, the AC / DC converter unit can control and boost the bidirectional current path simply and with high efficiency.
  • the drive logic circuit further includes a case where the current flowing between the AC power source that outputs a single-phase AC voltage and the AC / DC converter unit is zero (hereinafter, including substantially zero).
  • At least two of the plurality of AC switch units may be configured by two switching elements in which source terminals or drain terminals are connected in series.
  • step-up PFC control device like the step-up PFC control device described above, it is possible to reduce power consumption while maintaining the rectification step-up operation of the conventional step-up type PFC control device. Furthermore, a step-up PFC control device having high input voltage resistance can be realized. In addition, the step-up PFC control device can completely disconnect the load connected between the output terminal and the GND terminal of the step-up PFC control device from the input AC voltage source.
  • the switching element includes a stacked body formed of a plurality of nitride semiconductor layers formed on a semiconductor substrate, a gate terminal formed on the stacked body, and a gate terminal sandwiched between the gate terminals. It is desirable to provide a drain terminal and a source terminal formed on both sides.
  • the switching element is generally known as a heterojunction field effect transistor using a gallium nitride semiconductor and is called a GaN transistor.
  • a GaN transistor has FET characteristics and reverse FET characteristics when the gate / source voltage is higher than a certain threshold voltage, and has reverse conduction characteristics when the gate / source voltage is equal to or lower than the threshold voltage.
  • the GaN transistor also serves as a bidirectional switching element having high breakdown voltage characteristics, and is also an FET transistor having a very low on-resistance value in FET characteristics and reverse FET characteristics. The on-resistance is the resistance of the channel portion when a current flows through the switching element.
  • the switching frequency can be increased by reducing the switching loss, whereby the size of the boosting coil can be reduced, and as a result, the device can be reduced in size.
  • the present invention it is possible to provide a step-up PFC control device that does not include a diode and a parasitic diode of a switching element.
  • the power consumption of the conventional boost type PFC control device is largely due to the diode and the parasitic diode of the switching element, but the power consumption can be greatly reduced in the present invention.
  • the PFC control unit included in the boost type PFC control device can use the control unit of the conventional boost type PFC control device as it is, the conventional boost type PFC control device changes to the boost type PFC control device of the present invention. Is easy to replace.
  • FIG. 1 is a circuit block diagram of a step-up PFC control apparatus according to an embodiment of the present invention.
  • FIG. 2A is an IV characteristic diagram showing the FET characteristics of the bidirectional switching element used in the present invention.
  • FIG. 2B is an IV characteristic diagram showing the reverse FET characteristic of the bidirectional switching element used in the present invention.
  • FIG. 2C is an IV characteristic diagram showing reverse conduction characteristics of the bidirectional switching element used in the present invention.
  • FIG. 2D is a diagram for explaining equivalent conversion in an on-operation state of the switching element.
  • FIG. 2E is a diagram for explaining equivalent conversion in an OFF operation state of the switching element.
  • FIG. 3A is a circuit block diagram of the drive logic circuit according to the embodiment of the present invention.
  • FIG. 3A is a circuit block diagram of the drive logic circuit according to the embodiment of the present invention.
  • FIG. 3B is an operation timing chart of internal signals of the drive logic circuit according to the exemplary embodiment of the present invention.
  • FIG. 4A is a diagram illustrating a current path of the boost operation of the AC / DC converter unit according to the embodiment when the voltage polarity of the input AC power supply is positive.
  • FIG. 4B is a diagram illustrating a current path of the boost operation of the AC / DC converter unit according to the embodiment when the voltage polarity of the input AC power supply is negative.
  • FIG. 5 is a cross-sectional view of a bidirectional switching element included in the step-up PFC control device of the present invention.
  • FIG. 6 is a circuit block diagram of a step-up PFC control device according to a first modification of the embodiment of the present invention.
  • FIG. 7A is an equivalent circuit diagram illustrating a state of the AC switch according to the first modification example of the embodiment when the voltage polarity of the input AC power supply is positive.
  • FIG. 7B is an equivalent circuit diagram illustrating a state of the AC switch according to the first modification example of the embodiment when the voltage polarity of the input AC power supply is negative.
  • FIG. 8A is a circuit block diagram of a drive logic circuit according to a second modification example of the embodiment of the present invention.
  • FIG. 8B is an operation timing chart of internal signals of the drive logic circuit according to the second modification example of the exemplary embodiment of the present invention.
  • FIG. 9A is a diagram illustrating a current path of the boosting operation of the AC / DC converter unit according to the second modification example of the embodiment when the voltage polarity of the input AC power supply is positive.
  • FIG. 9B is a diagram illustrating a current path of the boosting operation of the AC / DC converter unit according to the second modification example of the embodiment when the voltage polarity of the input AC power supply is negative.
  • FIG. 10 is a circuit block diagram of a boost type PFC control device according to a comparative example.
  • FIG. 11A is an equivalent circuit diagram illustrating a state of the switching element when the voltage polarity of the input AC power supply according to the comparative example is positive.
  • FIG. 11B is an equivalent circuit diagram illustrating a state of the switching element when the voltage polarity of the input AC power supply according to the comparative example is negative.
  • FIG. 12A is a diagram illustrating transient characteristics of the switching element when the voltage polarity of the input AC power supply according to the comparative example is positive.
  • FIG. 12B is a diagram illustrating transient characteristics of the switching element when the voltage polarity of the input AC power supply according to the comparative example is negative.
  • FIG. 13 is a circuit block diagram of a step-up PFC control device described in Patent Document 1 without a rectifier diode bridge.
  • FIG. 1 is a circuit block diagram of a step-up PFC control apparatus according to an embodiment of the present invention.
  • a step-up PFC control device 100 shown in FIG. 1 includes an AC / DC converter unit 30, a current detection element 3, an output load 8, a second error amplifier 9, a multiplier circuit 10, and a second absolute value circuit 11. And a PWM comparator 12 and a sawtooth wave oscillator 14. Further, the step-up PFC control device 100 illustrated in FIG. 1 includes a differential amplifier 17, a reference voltage source 18, a first error amplifier 19, a comparator 20, and a first absolute value circuit 21.
  • the AC / DC converter unit 30 has a function of rectifying and boosting the AC voltage input from the input AC power supply 1.
  • the AC / DC converter unit 30 includes a boosting coil 4, a smoothing capacitor 7, AC switches S 1 to S 4, four pre-drive circuits 51, three power supplies 52, and a drive logic circuit 53.
  • the block configuration excluding the AC / DC converter unit 30 is the same as the block configuration of the conventional step-up PFC control device 500 shown in FIG. Therefore, in the description of the embodiment of the present invention, the AC / DC converter unit 30 will be mainly described.
  • the silicon-based semiconductor switching elements T1 and T2 shown in FIG. 13 are replaced with AC switches S1 and S2, which are bidirectional switching elements, respectively. Further, the diodes D1 and D2 shown in FIG. 13 are replaced with AC switches S3 and S4 using bidirectional switching elements.
  • AC switches S1 to S4 are bridge-connected as follows.
  • the input side electrode which is the first electrode of the boosting coil 4
  • the drain terminal of the AC switch S1 and the drain terminal of the AC switch S3 are connected to the first electrode of the smoothing capacitor 7.
  • the source terminal of the AC switch S2 and the source terminal of the AC switch S4 are connected to the ground electrode that is the second electrode of the smoothing capacitor 7. Further, the source terminal of the AC switch S1 and the drain terminal of the AC switch S2 are connected to the output side electrode that is the second electrode of the boosting coil 4.
  • the source terminal of the AC switch S3 and the drain terminal of the AC switch S4 are connected to the second AC terminal of the input AC power supply 1 to which the input AC voltage is applied.
  • the gate terminals of the AC switches S1 to S4 are connected to the respective predrive circuits 51 that shift the level of the control signal output from the drive logic circuit 53.
  • the boosting coil 4 stores magnetic energy corresponding to a bidirectional current that flows when a single-phase input AC voltage is applied by the input AC power source 1.
  • the smoothing capacitor 7 stores a charge corresponding to the magnetic energy stored in the boosting coil 4.
  • the bidirectional switching element is a bidirectional switching element having the IV characteristics shown in FIGS. 2A, 2B, and 2C. This characteristic will be described below.
  • FIG. 2A is an IV characteristic diagram showing an FET (Field Effect Transistor) characteristic of the bidirectional switching element used in the present invention.
  • FIG. 2B is an IV characteristic diagram showing the reverse FET characteristic of the bidirectional switching element used in the present invention.
  • FIG. 2C is an IV characteristic diagram showing reverse conduction characteristics of the bidirectional switching element used in the present invention.
  • the bidirectional switching element has a gate terminal, a drain terminal, and a source terminal.
  • the gate / source voltage Vgs is higher than the threshold voltage Vth, the current Ids from the drain terminal to the source terminal or from the source terminal to the drain terminal depending on the polarity of the differential voltage Vds between the drain terminal and the source terminal. Can flow.
  • the gate / source voltage Vgs is a differential voltage of the gate terminal voltage with respect to the source terminal voltage.
  • the current Ids is a positive value when flowing from the drain terminal to the source terminal.
  • the IV characteristic has a triode region and a saturation region like the IV characteristic of a MOSFET (Metal Oxide Semiconductor Field Effect Transistor).
  • the triode region is a region in the vicinity of zero voltage until Vds reaches a certain voltage value from zero voltage, and the saturation region is a characteristic similar to a constant current characteristic in which Ids does not change much even if Vds changes. It is an area
  • the IV characteristic has linearity, and the slope of Vds with respect to Ids can be defined as the on-resistance Ron of the switching element.
  • the on-resistance Ron in the triode region is sufficiently smaller than the on-resistance in the saturation region.
  • the characteristics of the bidirectional switching element in the triode region are important as the operation characteristics of the AC switches S1 to S4. In the following description, the characteristics of the triode region are limited.
  • FIG. 2A is an IV characteristic diagram when the differential voltage Vds is positive, that is, when the drain voltage is higher than the source voltage. As can be seen from the IV characteristic diagram, the current Ids has a positive value, and the current flows from the drain terminal to the source terminal.
  • the IV characteristics shown in FIG. 2A will be referred to as FET characteristics.
  • FIG. 2B is an IV characteristic diagram when the differential voltage Vds is negative, that is, when the drain voltage is lower than the source voltage. As can be seen from the IV characteristic diagram, the current Ids has a negative value, and the current flows from the source terminal to the drain terminal.
  • the IV characteristics shown in FIG. 2B will be referred to as inverse FET characteristics.
  • the bidirectional switching element cannot flow the current Ids from the drain terminal to the source terminal when the gate / source voltage Vgs is lower than the threshold voltage Vth.
  • the bidirectional switching element even when the gate / source voltage Vgs is lower than the threshold voltage Vth, the drain terminal voltage is lower than the gate terminal voltage, and the difference voltage (Vgs ⁇ Vds) is higher than the threshold voltage Vth.
  • a current Ids can flow from the source terminal to the drain terminal.
  • This characteristic is called reverse conduction characteristic.
  • FIG. 2C is an IV characteristic diagram showing the reverse conduction characteristic. As can be seen from FIG.
  • the I ⁇ of the diode whose forward voltage is the threshold voltage Vth with the source terminal as the anode and the drain terminal as the cathode. It is the same as the V characteristic.
  • the bidirectional switching element performs FET operation and reverse FET operation when the gate / source voltage Vgs is equal to or higher than the threshold voltage Vth, and the on-resistance It can be regarded as a resistor having the value Ron.
  • the gate-source voltage Vgs is equal to or lower than the threshold voltage Vth
  • the operation is performed according to the reverse conduction characteristic, and it can be regarded as a diode having the source terminal as an anode and the drain terminal as a cathode.
  • the forward voltage of this diode is (Vth ⁇ Vgs).
  • the bidirectional switching element is equivalently converted as follows.
  • the switching element In the ON operation state of the bidirectional switching element in which the gate / source voltage Vgs is higher than the threshold voltage Vth, the switching element is regarded as a resistance having an ON resistance value Ron.
  • the switching element In the off operation state of the bidirectional switching element in which the gate terminal and the source terminal are short-circuited, the switching element is regarded as a diode having the source terminal as an anode and the drain terminal as a cathode.
  • FIG. 2D is a diagram for explaining equivalent conversion between the FET mode and the reverse FET mode in the on-operation state of the switching element
  • FIG. 2E is a diagram for explaining equivalent conversion in the reverse conduction mode in the off-operation state of the switching element. .
  • FIG. 3A is a circuit block diagram of the drive logic circuit 53 according to the embodiment of the present invention
  • FIG. 3B is an operation timing chart of internal signals of the drive logic circuit 53 according to the embodiment of the present invention.
  • the drive logic circuit 53 when a PWM (Pulse Width Modulation) signal is input, the drive logic circuit 53 outputs an LPWM signal for PWM driving one of the AC switches S1 and S2 and the other of the AC switches S1 and S2.
  • a UPWM signal for PWM driving is generated.
  • the PWM signal is a signal output from the PWM comparator 12 shown in FIG. 1, and is a signal for PWM control of the boosting operation.
  • One of the AC switches S1 and S2 drives the boosting coil 4 for the boosting operation by the LPWM signal, and the other of the AC switches S1 and S2 operates in synchronization with one of the AC switches S1 and S2 by the UPWM signal.
  • the LPWM signal has the same waveform as the PWM signal for PWM control of the boost operation, but is delayed from the PWM signal by a certain delay time DT.
  • the UPWM signal is a signal having a polarity opposite to that of the LPWM signal, and a waveform is generated so that a section in which the UPWM signal and the LPWM signal are simultaneously a low level signal (hereinafter referred to as “Low”) is provided for the delay time DT. ing.
  • a section in which these two signals are Low at the same time is called a dead time. Due to this dead time, as long as the two AC switches S1 and S2 are operating normally, they cannot be in the ON operation state at the same time.
  • the AC switch S1 or S2 operates as a boosting diode in the dead time interval (of course, not only in the dead time interval but also in the interval where the UPWM signal is Low) according to the rules of equivalent conversion described above.
  • the AC switch S1 or S2 operates as a resistor having an on-resistance value Ron in a section where the UPWM signal is a high level signal (hereinafter referred to as “High”).
  • the section in which current flows while the AC switches S1 and S2 are operating as boosting diodes is a very short period of time, which is only the dead time section. Therefore, the power consumption of the AC switches S1 and S2 is very small compared to the switching elements T1 and T2 of the conventional step-up PFC control device.
  • the four output signals G_S1 to G_S4 of the drive logic circuit 53 control the four AC switches S1 to S4.
  • the output signals G_S1 and G_S2 output the LPWM signal or the UPWM signal
  • the output signals G_S3 and G_S4 output the high or low level signal.
  • the DIR signal shown in FIG. 3B becomes a high level signal when the AC output of the input AC power supply 1 applies a positive voltage to the boosting coil 4, and the AC output is for boosting.
  • a negative voltage is applied to the coil 4, it becomes a Low level signal.
  • FIG. 4A is a diagram illustrating a current path of the step-up operation of the AC / DC converter unit according to the embodiment when the voltage polarity of the input AC power supply is positive.
  • FIG. 4B is a diagram illustrating a current path of the boost operation of the AC / DC converter unit according to the embodiment when the voltage polarity of the input AC power supply 1 is negative.
  • the boosting operation described in FIGS. 4A and 4B is realized by the PWM operation of the drive logic circuit 53 shown in FIG. 3A.
  • a current always flows through the smoothing capacitor 7 by the magnetic energy of the boosting coil 4 and the output voltage Vo is boosted regardless of whether the voltage polarity of the input AC power supply 1 is positive or negative. .
  • the DIR signal When the input AC power supply 1 is positive, the DIR signal is at a high level.
  • the AC switch S2 receives the LPWM signal for driving the boosting coil 4, and the AC switch S1 is the UPWM signal. Receive.
  • the AC switch S3 receives a Low level signal, and the AC switch S4 receives a High level signal.
  • the AC switches S1 to S4 receive these signals and enter the operating state shown in FIG. 4A.
  • the LPWM signal is described as a PWM signal for controlling the step-up PWM operation
  • the UPWM signal that is a synchronization signal is described as a PWM signal with a bar that represents an inverted signal of the PWM signal. ing.
  • the AC switch S2 performs a PWM operation to drive the boosting coil 4, and the AC switch S1 is in an operation state in which the PWM operation is inverted.
  • the AC switch S4 functions as a resistor having an on-resistance Ron and allows a current to flow.
  • the AC switch S3 can be regarded as a diode in terms of an equivalent circuit because of an off operation state, but no current flows.
  • the AC switch S1 functions as a boosting diode.
  • the current in the booster coil 4 is smoothed by the magnetic energy stored in the booster coil 4 via the AC switch S1 in the path (b) of FIG. 4A. It flows into the terminal of the capacitor 7 which is not grounded. This is because the AC switch S1 functions as a boosting diode in the dead time period, and the AC switch S1 functions as a resistor having an ON resistance Ron when the AC switch S1 is in the ON operation state.
  • the DIR signal is at a low level. From the timing waveform diagram of FIG. 3B, the movement of the AC switches S2 and S1 compared to the case where the input AC power supply 1 has a positive polarity. It can be seen that the movements of the AC switches S4 and S3 are switched.
  • the AC switch S1 performs a PWM operation to drive the boosting coil 4, and the AC switch S2 is in an operating state in which the PWM operation is inverted.
  • the AC switch S3 functions as a resistor having an on-resistance Ron, and a current flows.
  • the AC switch S4 is in an off operation state and can be regarded as a diode in terms of an equivalent circuit, but no current flows.
  • the aforementioned dead time section is provided in the PWM operation in which the AC switches S1 and S2 are operated in synchronization. Then, the AC switch S2 performs the same operation as the AC switch S1 when the input AC power supply 1 is positive, and the current of the boosting coil 4 is passed through the AC switch S2 through the path (b) of the smoothing capacitor 7. It flows into a terminal that is not grounded.
  • the drive logic circuit 53 switches the current path of the bidirectional current by controlling the on / off states of the AC switches S1 to S4 to accumulate magnetic energy in the boosting coil 4 and charge to the smoothing capacitor 7. Accumulate and alternate.
  • the step-up PFC control device 100 shown in FIG. 1 has a function of rectifying and stepping up the AC voltage of the input AC power supply 1 even though there is no rectifier diode bridge.
  • the hysteresis comparator shown in FIG. 3A detects that almost no current flows.
  • the hysteresis comparator compares the signal Sc obtained by converting the output signal of the current detection element 3 into a positive voltage signal by the second absolute value circuit 11 and the comparison reference voltage VRB, thereby obtaining the current flowing through the boosting coil 4.
  • the AC switch S1 or S2 operates as a resistor having an on-resistance value Ron in response to a high-level gate signal. However, the AC switch S1 or S2 changes to the diode operation by changing the gate signal from the High level to the Low level by the zero current detection operation of the hysteresis comparator.
  • the step-up PFC control apparatus 100 realizes a step-up PFC control apparatus without a diode and without a parasitic diode of a switching element.
  • the conduction loss and the switching loss influenced by the diode and the parasitic diode of the switching element occupy a large proportion of the power loss of the entire device.
  • the power consumption can be greatly reduced while maintaining the conventional rectification and step-up operation by eliminating the diode and the parasitic diode of the switching element.
  • the bidirectional switching element used as an AC switch does not have a parasitic diode, and therefore there is no recovery current due to the parasitic diode.
  • the switching loss of the step-up PFC controller does not increase significantly, and the step-up coil 4 can be made smaller by increasing the switching frequency.
  • the boost type PFC control device included in the boost type PFC control device can use the control unit of the conventional boost type PFC control device as it is, the boost type PFC control device of the present invention is changed from the conventional boost type PFC control device. Replacement is also easy.
  • the switching element described above includes a stacked body formed of a nitride semiconductor layer formed on a semiconductor substrate, a drain terminal and a source terminal formed on the stacked body at intervals, and a drain terminal and a source. And a gate terminal formed between the terminals. This switching element will be described with reference to FIG.
  • FIG. 5 is an example of a cross-sectional view of a bidirectional switching element included in the step-up PFC control device of the present invention.
  • the bidirectional switching element shown in the figure is a normally-off type heterojunction FET made of a nitride semiconductor formed on a semiconductor substrate. Specifically, the switching element is realized by forming a stacked body 203 of semiconductor layers on a silicon substrate 201 via a buffer layer 202.
  • the buffer layer 202 is formed by alternately stacking aluminum nitride and gallium nitride.
  • an n-type aluminum gallium nitride layer 205 is formed on an undoped gallium nitride layer 204, and an FET having a high carrier concentration called a two-dimensional electron gas is located in the vicinity of the heterointerface between the two layers. Channel regions are generated.
  • a source terminal ohmic electrode 206a, a drain terminal ohmic electrode 206b, and a wiring 210, which are in ohmic contact with the channel region, are arranged.
  • a control layer 209 which is a p-type semiconductor layer for controlling FET characteristics, is formed on the n-type aluminum gallium nitride layer 205.
  • a protective film 207 for protecting each part is formed so as to cover each part.
  • a gate electrode 208 is formed on the control layer 209 and is in ohmic contact with the control layer 209.
  • the electric signal supplied to the gate electrode 208 controls the current flowing between the drain terminal and the source terminal of the normally-off type heterojunction FET, that is, the bidirectional switching element.
  • the distance from the drain terminal ohmic electrode 206b to the gate electrode 208 is longer than the distance from the source terminal ohmic electrode 206a to the gate electrode 208 because the withstand voltage between the drain terminal and the gate terminal is higher than the source terminal. This is because it is required to be larger than the breakdown voltage between the gate terminals.
  • the bidirectional switching element formed as shown in FIG. 5 is called a GaN transistor, and is a device that can be driven with a high voltage and a large current, such as an IGBT (Insulated Gate Bipolar Transistor).
  • the switching element does not have an offset voltage due to the PN junction in the current-voltage characteristics of the IGBT, and has a characteristic of flowing a current to both as shown in FIGS. 2A and 2B.
  • the switching element has a very small on-resistance component Ron with respect to the chip area of the device.
  • the GaN transistor also has the reverse conduction characteristics shown in FIG. 2C.
  • the GaN transistor has almost no accumulation effect due to minority carriers, and almost no tail current effect during turn-off as in IGBTs and other silicon-based semiconductor elements.
  • the AC / DC converter unit 30 is composed of an AC switch using a GaN transistor as a bidirectional switching element. Therefore, the AC / DC converter unit 30 can greatly reduce power consumption by reducing conduction loss due to its very small on-resistance value Ron and reducing switching loss due to almost no minority carrier accumulation effect.
  • the apparatus can be miniaturized.
  • FIG. 6 is a circuit block diagram of the step-up PFC control apparatus according to the first modification of the embodiment of the present invention.
  • the step-up PFC control device 150 shown in the figure is different from the step-up PFC control device 100 shown in FIG. 1 in the configuration of a part of the AC switch of the AC / DC converter unit 31.
  • the description of the same points as the step-up PFC control apparatus 100 described in FIG. 1 will be omitted, and only different points will be described.
  • AC switches S2 and S4 are each composed of two bidirectional switching elements whose drain terminals are connected in series. With this configuration, it is possible to increase the withstand voltage of the AC switch, and it is possible to realize a step-up PFC control device with a higher input withstand voltage. Further, in the step-up type PFC control device 150, the output side can be disconnected from the input AC power supply 1 of the step-up type PFC control device 150 by giving a low level signal to the illustrated off signal (OFF).
  • FIG. 7A is an equivalent circuit diagram showing the state of the AC switch according to the first modification of the embodiment when the voltage polarity of the input AC power supply is positive.
  • FIG. 7B is an equivalent circuit diagram showing the state of the AC switch according to the first modification of the embodiment when the voltage polarity of the input AC power supply is negative. 7A and 7B, the PWM signal that is the output of the PWM comparator 12 is at the low level, and the off signal is also at the low level.
  • each of the AC switches S2 and S4 is constituted by a series connection of two bidirectional switching elements.
  • two bidirectional switching elements connected in series may be applied to the AC switches S1 and S3, and the AC switches S2 and S4 may be normal AC switches.
  • the AC switches S2 and S4 have a configuration in which the drain terminals of the two switching elements are connected to each other, but may be connected in series with the source terminals connected to each other.
  • FIG. 8A is a circuit block diagram of a drive logic circuit 63 according to a second modification of the embodiment of the present invention
  • FIG. 8B illustrates a drive logic circuit 63 according to the second modification of the embodiment of the present invention. It is an operation
  • the drive logic circuit 63 illustrated in FIG. 8A When the PWM signal is input, the drive logic circuit 63 illustrated in FIG. 8A generates an LPWM signal for PWM driving one of the AC switches S2 and S4, and PWM driving one of the AC switches S1 and S3.
  • the PWM signal is a signal output from the PWM comparator 12 shown in FIG. 1, and is a signal for PWM control of the boosting operation.
  • One of the AC switches S2 and S4 drives the boosting coil 4 for the boosting operation by the LPWM signal, and one of the AC switches S1 and S3 operates in synchronization with one of the AC switches S2 and S4 by the UPWM signal.
  • the relationship between the PWM signal, the UPWM signal, and the LPWM signal is the same as the relationship described in FIG. 3B.
  • a section in which these two signals are Low at the same time is called a dead time. Due to the dead time when the UPWM signal and the LPWM signal are simultaneously Low, as long as the two AC switches S1 and S2 or the AC switches S3 and S4 are operating normally, they cannot be in the ON operation state at the same time. .
  • the AC switch S3 or S4 that performs switching operation with the UPWM signal is boosted in the dead time interval (of course, not only in the dead time interval but also in the interval in which the UPWM signal is Low) according to the rules of equivalent conversion described above. Operates as a diode. Further, in a section where the UPWM signal is High, the AC switch S3 or S4 operates as a resistor having an on-resistance value Ron.
  • the period during which the current flows while the AC switches S1 and S3 are operating as boosting diodes is a very short period of only the dead time period. Therefore, the power consumption of the AC switches S1 and S3 is very small compared to the switching elements T3 and T4 of the conventional step-up PFC control device.
  • the output signals G_S1 to G_S4 control the AC switches S1 to S4, respectively.
  • the output signals G_S2 and G_S4 output an LPWM signal or a high level signal
  • the output signals G_S1 and G_S3 output a UPWM signal or a low level signal.
  • the DIR signal shown in FIG. 8B becomes a high level signal when the AC output of the input AC power supply 1 applies a positive voltage to the boosting coil 4.
  • the DIR signal is a Low level signal when the AC output of the input AC power supply 1 applies a negative voltage to the boosting coil 4.
  • FIG. 9A is a diagram illustrating a current path of the boosting operation of the AC / DC converter unit when the voltage polarity of the input AC power supply is positive.
  • FIG. 9B is a diagram illustrating a current path of the boosting operation of the AC / DC converter unit when the voltage polarity of the input AC power supply is negative.
  • the boosting operation described in FIGS. 9A and 9B is realized by the PWM operation of the drive logic circuit 63 shown in FIG. 8A.
  • a current always flows through the smoothing capacitor 7 by the magnetic energy of the boosting coil 4 and the output voltage Vo is boosted regardless of whether the voltage polarity of the input AC power supply 1 is positive or negative. .
  • the DIR signal When the input AC power supply 1 is positive, the DIR signal is at a high level.
  • the AC switch S2 receives the LPWM signal for driving the boosting coil 4, and the AC switch S1 is the UPWM signal. Receive.
  • the AC switch S3 receives a Low level signal, and the AC switch S4 receives a High level signal.
  • the AC switches S1 to S4 receive these signals and enter the operating state shown in FIG. 9A.
  • the LPWM signal is described as a PWM signal for controlling the step-up PWM operation
  • the UPWM signal that is a synchronization signal is described as a PWM signal with a bar that represents an inverted signal of the PWM signal. ing.
  • the AC switch S2 performs a PWM operation to drive the boosting coil 4, and the AC switch S1 is in an operation state in which the PWM operation is inverted.
  • the AC switch S4 functions as a resistor having an on-resistance Ron and allows a current to flow.
  • the AC switch S3 can be regarded as a diode in terms of an equivalent circuit because of an off operation state, but no current flows.
  • the DIR signal is at a low level. From the timing waveform diagram of FIG. 8B, the movement of the AC switches S2 and S4 compared to the case where the input AC power supply 1 has a positive polarity. It can be seen that the AC switches S1 and S3 are switched.
  • the AC switch S4 performs a PWM operation to drive the boosting coil 4, and the AC switch S3 is in an operating state in which the PWM operation is inverted.
  • the AC switch S2 passes a current as a resistor having an ON resistance Ron, and the AC switch S1 is in an OFF operation state, and can be regarded as a diode in terms of an equivalent circuit, but no current flows.
  • a current for exciting the boosting coil 4 flows through the path (a) in FIG. 9B.
  • the AC switch S4 is in the OFF operation state by the PWM operation, the current flows through the AC switch S3 in the path (b) of FIG. 9B due to the magnetic energy stored in the boosting coil 4. Then, the current is charged from the terminal of the smoothing capacitor 7 that is not grounded, and the output voltage Vo of the step-up PFC control device 200 is stepped up.
  • the aforementioned dead time interval is provided in the PWM operation in which the AC switches S4 and S3 are operated synchronously. Then, the AC switch S3 performs the same operation as the AC switch S1 when the input AC power supply 1 is positive, and the current of the boosting coil 4 is passed through the AC switch S3 through the path (b) of the smoothing capacitor 7. It flows into a terminal that is not grounded.
  • the hysteresis comparator shown in FIG. 8A detects that almost no current flows.
  • the hysteresis comparator compares the signal Sc obtained by converting the output signal of the current detection element 3 into a positive voltage signal by the second absolute value circuit 11 and the comparison reference voltage VRB, thereby obtaining the current flowing through the boosting coil 4.
  • the AC switch S1 or S3 is operated synchronously with the UPWM signal with respect to the AC switch S2 or S4 that switches the boosting coil 4 with the LPWM signal.
  • the AC switch S1 or S3 has its gate signal changed to Low level by the zero current detection operation of the hysteresis comparator, and changes to diode operation.
  • the drive logic circuit 63 realizes a step-up PFC control device without a diode and without a parasitic diode of a switching element.
  • FIG. 10 is a circuit block diagram of a boost type PFC control device according to a comparative example.
  • the step-up type PFC control device 300 shown in the figure is a step-up type PFC circuit without a rectifier diode bridge.
  • a step-up PFC control apparatus 300 includes a pair of switching elements T3 and T4 and a high-frequency block filter 502 included in the step-up PFC control apparatus 500 of Patent Document 1 described in FIG. Excluded.
  • the pair of switching elements T3 and T4 is a mechanism for performing the switching operation twice while shifting the phase by 180 degrees for the boosting operation.
  • the pair of switching elements T1 and T2 that perform the switching operation included in the step-up PFC control device 500 are replaced with silicon-based semiconductor switching elements T5 and T6 such as an n-type MOSFET.
  • a pre-drive circuit 51 having a level shift function for driving the switching elements T5 and T6 and a power source 52 are added.
  • the first error amplifier 19 generates an error voltage Ve that is proportional to the difference between the output voltage Vo of the step-up PFC controller 300 and the voltage value Vref of the reference voltage source 18 that sets the output voltage.
  • a voltage signal Vina having a waveform similar to a voltage obtained by full-wave rectifying the AC voltage of the input AC power supply 1 is generated by the differential amplifier 17 and the first absolute value circuit 21.
  • the multiplier circuit 10 multiplies the error voltage Ve and the voltage signal Vina to generate a voltage signal Iref that becomes a current control command value.
  • the voltage signal Iref is proportional to the error voltage Ve and has the same pulsation waveform as Vina that is similar to the voltage waveform obtained by full-wave rectifying the AC voltage of the input AC power supply 1.
  • the current flowing from the input AC power source 1 to the boosting coil 4 by the switching operation of the switching elements T5 and T6 is detected by the current detection element 3, and converted to a positive voltage value signal by the second absolute value circuit 11. It is converted to a current-shaped waveform waveform Sc.
  • Second error amplifier 9 PWM comparator 12, sawtooth oscillator 14, switching element T5 or T6, boosting coil 4, diode D1 or D2, smoothing capacitor 7, output load 8, current detection element 3, and second absolute value
  • the circuit 11 forms a current control negative feedback loop.
  • the current waveform signal Sc is controlled to follow the voltage signal Iref by this current control negative feedback loop, and has substantially the same value as the voltage signal Iref.
  • the voltage signal Iref is a pulsation waveform similar to a voltage waveform obtained by full-wave rectification of the AC voltage of the input AC power supply 1. Therefore, the fact that the current waveform signal Sc follows the voltage signal Iref means that the AC voltage of the input AC power supply 1 and the AC current of the input AC power supply 1 have substantially the same phase and substantially the same waveform. Therefore, the power factor of the input AC power supply is approximately 1.
  • the error voltage Ve is a voltage obtained by amplifying the difference between the output voltage Vo and the voltage value Vref by the first error amplifier 19.
  • the error voltage Ve that is the output of the first error amplifier 19 is controlled to be a finite value by the PFC control loop including the first error amplifier 19, the multiplier circuit 10, and the above-described current control negative feedback loop. .
  • the error voltage Ve is described by the following equation, where A is the gain of the first error amplifier 19.
  • the output voltage Vo maintains the desired voltage set by the voltage value Vref of the reference voltage, and the power factor of the input AC power supply.
  • a step-up AC / DC converter with a high input power factor capable of reducing the power to approximately 1 is realized.
  • the differential amplifier 17, the comparator 20, the NOT logic circuit 13, the two AND logic circuits 16, the switching elements T5 and T6, and the diodes D1 and D2 realize a rectified boost switching operation without using a rectifier diode bridge. .
  • this point will be described.
  • the comparator 20 compares the output signal of the differential amplifier 17 to determine the polarity of the AC voltage of the input AC power supply 1 (the polarity of the AC voltage of the input AC power supply 1 is input to the boosting coil 4). When the AC power supply 1 applies a positive voltage, the polarity is positive). If the polarity of the alternating voltage is positive, the output signal of the comparator 20 is at a high level, and if the polarity of the alternating voltage is negative, the output is low.
  • the NOT logic circuit 13 and the two AND logic circuits 16 receive the output signal of the comparator 20 and control the switching elements T5 and T6. Specifically, if the polarity of the input AC voltage is positive, the NOT logic circuit 13 and the two AND logic circuits 16 are connected to the switching element T6 by the PWM signal output from the PWM comparator 12 for the boosting switching operation. PWM drive. Then, the NOT logic circuit 13 and the two AND logic circuits 16 set the gate voltage of the switching element T5 to the Low level to turn off the switching element T5. On the other hand, if the polarity of the input AC voltage is negative, the NOT logic circuit 13 and the two AND logic circuits 16 PWM drive the switching element T5 with a PWM signal. Then, the NOT logic circuit 13 and the two AND logic circuits 16 set the gate voltage of the switching element T6 to the Low level to turn off the switching element T6.
  • FIG. 11A is an equivalent circuit diagram showing the state of the switching element when the voltage polarity of the input AC power supply according to the comparative example is positive.
  • FIG. 11B is an equivalent circuit diagram illustrating a state of the switching element when the voltage polarity of the input AC power supply according to the comparative example is negative. That is, FIG. 11A and FIG. 11B show which is different depending on the PWM operation by the operation of the comparator 20, the NOT logic circuit 13, and the two AND logic circuits 16, depending on whether the voltage polarity of the input AC power supply 1 is positive or negative. It is a figure explaining how voltage
  • a current for exciting the boosting coil 4 flows through the path (a) in FIG. 11A. Further, when the switching element T6 is turned off by the PWM operation, a current flows by the magnetic energy stored in the boosting coil 4 through the path (b) of FIG. 11A. Then, a current is charged from a terminal of the smoothing capacitor 7 that is not grounded, and the output voltage Vo of the boost PFC control device 300 is boosted. Since the switching elements T5 and T6 are silicon semiconductor switching elements such as n-type MOSFETs, there are parasitic diodes between the source terminal and the drain terminal as shown in FIG. Therefore, as shown in FIG. 11A, the current generated by the magnetic energy of the boosting coil 4 in the path (b) flows via the parasitic diode of the switching element T5.
  • the boost type PFC control device 300 according to the comparative example has the following two problems.
  • the problem (1) regarding the forward voltage VF of the diode and the forward voltage VFP of the parasitic diode will be described.
  • the forward voltage VF of the diode D2 is generated when a current flows through the path (a) in FIG. 11A.
  • a current flows through the path (b) in FIG. 11A a voltage drop due to the forward voltage VFP of the switching element T5 in the off operation state occurs in addition to the forward voltage VF of the diode D2.
  • Power consumption that is, loss occurs in the diode D2 and the switching element T5 due to these voltage drops and the current flowing through these diodes.
  • the forward voltage VF of the diode D1 is generated when a current flows through the path (a) in FIG. 11B. Further, when a current flows through the path (b) in FIG. 11B, a voltage drop due to the forward voltage VF of the diode D1 and the forward voltage VFP of the switching element T6 occurs, and loss occurs in the diode D1 and the switching element T6. To do.
  • the power consumption of the element due to the constant flow of current is called conduction loss.
  • the conduction loss when the current flows while the switching element T5 or T6 is in the on-operation state is very small compared to the conduction loss due to the voltage drop of the forward voltage VF or VFP of the diode because the on-resistance is small. Therefore, as can be seen from the current paths shown in FIGS. 11A and 11B, it can be seen that most of the conduction loss of the conventional boost type PFC control device is due to the forward voltages VF and VFP of the diode.
  • the conduction loss of the boost type PFC control device 300 can be greatly reduced.
  • the anode of the parasitic diode existing between the source terminal and the drain terminal of the switching elements T5 and T6 is at the source of the switching element, and the cathode of the parasitic diode is at the drain of the switching element.
  • FIG. 11A when the switching element T5 is in the OFF operation state, when a current flows through the path (b), a current flows from the anode to the cathode of the parasitic diode.
  • FIG. 11B when the switching element T6 is in the OFF operation state, when a current flows through the path (b), a current flows from the anode to the cathode of the parasitic diode.
  • the recovery current component due to the minority carrier accumulation effect in the diode is generated in the parasitic diode. That is, when the current starts to flow from the path (b) state to the path (a) state, the recovery current of the parasitic diode is also driven.
  • the path (b) is a state in which one of the switching elements T5 and T6 is in an off operation state and a current flows through the parasitic diode.
  • the other of the switching elements T5 and T6 is turned on to become the path (a), not only the current for driving the boosting coil 4 but also the unnecessary parasitic diode recovery current is generated by the turn-on operation. To drive.
  • FIG. 12A is a diagram illustrating the transient characteristics of the switching element when the voltage polarity of the input AC power supply according to the comparative example is positive.
  • FIG. 12B is a diagram illustrating transient characteristics of the switching element when the voltage polarity of the input AC power supply according to the comparative example is negative.
  • the current Ids (T6) flowing through the switching element T6 includes a drive current of the boosting coil 4 as indicated by a region R1 by a turn-on operation that is a change from the state (b) to the state (a).
  • the recovery current of the parasitic diode of the switching element T5 is superimposed.
  • the current Ids (T5) flowing through the switching element T5 is changed to the current Ids (T5) flowing from the state (b) to the state (a) as shown in the region R2, as shown in FIG.
  • the recovery current of the parasitic diode of the switching element T6 is superimposed.
  • the recovery current is a large one that cannot be ignored and affects the increase in switching loss, which means power consumption during switching operation of the switching element.
  • This switching loss increases in proportion to the switching frequency of the step-up PFC control device. Therefore, increasing the switching frequency for the purpose of reducing the size of the boosting coil 4 causes an increase in switching loss of the switching element. That is, it is difficult for the boost type PFC control apparatus 300 according to this comparative example to downsize the boost coil in the future.
  • a considerably large switching loss is generated, so that it is not suitable as a step-up PFC control device in the current continuous mode.
  • Appropriate conditions for using this boosting method must be based on the premise of so-called soft switching in which the switching element performs switching operation when no current flows. That is, in this boosting method, there is a limitation on the PFC control method.
  • the parasitic diode of the switching element is not only involved in the increase in power consumption due to the forward voltage VF of the parasitic diode itself, but also the different switching element. It also has a problem of affecting the switching loss.
  • the step-up PFC control device includes an AC / DC converter unit that rectifies and boosts an input single-phase AC voltage. Furthermore, the AC / DC converter unit includes a plurality of AC switch units that are bridge-connected, a boosting coil, a smoothing capacitor, and a drive logic circuit.
  • the boosting coil stores magnetic energy corresponding to the bidirectional current that flows in response to the application of the single-phase AC voltage.
  • the smoothing capacitor stores a charge corresponding to the magnetic energy stored in the boosting coil.
  • the drive logic circuit switches the current path of bidirectional current by controlling the on / off state of a plurality of AC switch units, and alternately stores magnetic energy in the boosting coil and charges in the smoothing capacitor. Make it.
  • Each of the plurality of AC switch units has a gate terminal, a drain terminal, and a source terminal.
  • the AC switch includes a bidirectional switching element having FET characteristics, reverse FET characteristics, and reverse conduction characteristics. FET characteristics means that when a gate / source voltage, which is a differential voltage of a gate terminal voltage with respect to a source terminal voltage, is higher than a threshold voltage, a current flows from the drain terminal to the source terminal according to the polarity of the drain / source voltage.
  • the reverse FET characteristic is a characteristic that allows a current to flow from the source terminal to the drain terminal according to the polarity of the drain / source voltage when the gate / source voltage is higher than the threshold voltage.
  • Reverse conduction characteristics means that when the gate-source voltage is less than or equal to the threshold voltage, the current from the drain terminal to the source terminal is cut off, but when the gate terminal voltage exceeds the threshold voltage with respect to the drain terminal voltage, the source terminal The current can flow from the drain terminal to the drain terminal.
  • the boosting PFC control unit for controlling the bidirectional current path can use the boosting PFC control unit of the conventional boosting PFC control device as it is, and the control operation as the boosting PFC control device is conventional. Is almost the same.
  • the drive logic circuit receives a control signal from a hysteresis comparator that detects that there is no current when the current flowing between the AC / DC converter unit and the input AC power supply is substantially zero. Then, control is performed so that the gate / source voltage of at least one of the plurality of AC switch units is equal to or lower than the threshold voltage.
  • At least two of the AC switch units are two bidirectional switching elements in which source terminals or drain terminals are connected in series. Composed.
  • step-up PFC control device like the step-up PFC control device described above, it is possible to reduce power consumption while maintaining the rectification step-up operation of the conventional step-up type PFC control device. Furthermore, a step-up PFC control device having high input voltage resistance can be realized. In addition, the step-up PFC control device can completely disconnect the load connected between the output terminal and the GND terminal of the PFC control device from the input AC voltage source.
  • the bidirectional switching element is formed on a stacked body including a nitride semiconductor layer formed on a semiconductor substrate, and on the stacked body. And a drain terminal and a source terminal formed on both sides of the gate terminal.
  • the bidirectional switching element is generally known as a heterojunction field effect transistor using a gallium nitride semiconductor and is called a GaN transistor.
  • a GaN transistor has FET characteristics and reverse FET characteristics when the gate / source voltage is higher than a certain threshold voltage, and has reverse conduction characteristics when the gate / source voltage is equal to or lower than the threshold voltage.
  • the GaN transistor also serves as a bidirectional switching element having high breakdown voltage characteristics, and is also an FET transistor having a very low on-resistance value in FET characteristics and reverse FET characteristics. Further, there is no minority carrier effect as in a silicon-based semiconductor element, and there is almost no influence of an increase in switching loss due to a recovery current. Therefore, by using a GaN transistor as a bidirectional switching element included in the step-up PFC control device of the present invention, a step-down PFC control device with lower power consumption can be realized.
  • the boost type PFC control device of the present invention has been described based on the embodiment, the boost type PFC control device according to the present invention is not limited to the above embodiment and its modifications. Another embodiment realized by combining arbitrary constituent elements in the embodiment and its modifications is also included in the present invention. Furthermore, the present invention also includes modifications obtained by making various modifications conceivable by those skilled in the art without departing from the spirit of the present invention to the embodiments and modifications thereof. Furthermore, various devices incorporating the boost type PFC control device according to the present invention are also included in the present invention.
  • the present invention can be applied to an AC / DC converter that outputs a DC voltage from an AC input, and is particularly useful as an AC / DC converter having a high-input power factor and high-efficiency step-up PFC controller that does not require a rectifier diode bridge. It is.

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Abstract

La présente invention concerne un dispositif de commande de correcteur du facteur de puissance de convertisseur survolteur (100), qui redresse une tension en courant alternatif monophasée et qui lui procure un effet survolteur, comprenant un convertisseur alternatif/continu (30). Ce convertisseur alternatif/continu (30) comprend : une pluralité d'interrupteurs en courant alternatif (S1 à S4) ; et un circuit logique d'excitation (53), qui réalise en alternance un stockage de charge vers un condensateur de lissage (7) et un stockage d'énergie magnétique vers une inductance à effet survolteur (4) en commandant l'état marche/arrêt des interrupteurs en courant alternatif (S1 à S4). Lorsqu'une tension grille-source est supérieure à une tension de seuil, les interrupteurs en courant alternatif (S1 à S4) laissent circuler le courant entre un drain et une source. Lorsque la tension grille-source est inférieure ou égale à la tension de seuil, et lorsque la tension de grille par rapport à une tension de drain est au moins égale à la tension de seuil, les interrupteurs en courant alternatif (S1 à S4) laissent circuler le courant entre la source et le drain.
PCT/JP2012/000950 2011-03-07 2012-02-14 Dispositif de commande de correcteur du facteur de puissance de convertisseur survolteur Ceased WO2012120788A1 (fr)

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JP6959167B2 (ja) * 2018-03-07 2021-11-02 シャープ株式会社 力率改善回路
CN112425057A (zh) 2018-07-19 2021-02-26 三菱电机株式会社 电力变换装置、马达驱动装置以及空气调节器
JP2020145842A (ja) * 2019-03-06 2020-09-10 ローム株式会社 電力変換装置
JP7281589B2 (ja) * 2019-10-16 2023-05-25 日立ジョンソンコントロールズ空調株式会社 直流電源装置及びこれを搭載した空気調和機
WO2021171572A1 (fr) * 2020-02-28 2021-09-02 三菱電機株式会社 Dispositif convertisseur

Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2005168136A (ja) * 2003-12-01 2005-06-23 Makita Corp モータの制御装置
JP2008125310A (ja) * 2006-11-15 2008-05-29 Sakae Shibazaki スイッチング電源装置
JP2008125312A (ja) * 2006-11-15 2008-05-29 Sakae Shibazaki スイッチング電源装置
WO2009147774A1 (fr) * 2008-06-05 2009-12-10 パナソニック株式会社 Dispositif semi-conducteur
JP2010004697A (ja) * 2008-06-23 2010-01-07 Panasonic Corp 双方向スイッチのゲート駆動方法およびそれを用いた電力変換装置

Patent Citations (5)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2005168136A (ja) * 2003-12-01 2005-06-23 Makita Corp モータの制御装置
JP2008125310A (ja) * 2006-11-15 2008-05-29 Sakae Shibazaki スイッチング電源装置
JP2008125312A (ja) * 2006-11-15 2008-05-29 Sakae Shibazaki スイッチング電源装置
WO2009147774A1 (fr) * 2008-06-05 2009-12-10 パナソニック株式会社 Dispositif semi-conducteur
JP2010004697A (ja) * 2008-06-23 2010-01-07 Panasonic Corp 双方向スイッチのゲート駆動方法およびそれを用いた電力変換装置

Cited By (9)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
JP2015023606A (ja) * 2013-07-16 2015-02-02 新電元工業株式会社 力率改善回路
CN110809854A (zh) * 2017-07-07 2020-02-18 三菱电机株式会社 交流直流变换装置、电动机驱动控制装置、送风机、压缩机以及空调机
EP3651337A4 (fr) * 2017-07-07 2020-06-24 Mitsubishi Electric Corporation Dispositif de conversion ca/cc, dispositif de commande d'entraînement de moteur, ventilateur, compresseur, et climatiseur
US10938318B2 (en) 2017-07-07 2021-03-02 Mitsubishi Electric Corporation AC-DC converting apparatus, motor drive control apparatus, blower, compressor, and air conditioner
CN110809854B (zh) * 2017-07-07 2021-04-09 三菱电机株式会社 交流直流变换装置、电动机驱动控制装置、送风机、压缩机以及空调机
JP2022052838A (ja) * 2020-09-24 2022-04-05 株式会社東芝 電流検出回路、電流検出システム、および電源回路
JP7434129B2 (ja) 2020-09-24 2024-02-20 株式会社東芝 電流検出回路、電流検出システム、および電源回路
JP2024041935A (ja) * 2020-09-24 2024-03-27 株式会社東芝 電流検出回路及び電流検出システム
JP7673263B2 (ja) 2020-09-24 2025-05-08 株式会社東芝 電流検出回路及び電流検出システム

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