Classifications

• • H—ELECTRICITY
  • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
  • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
  • H02M3/00—Conversion of DC power input into DC power output
  • H02M3/02—Conversion of DC power input into DC power output without intermediate conversion into AC
  • H02M3/04—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters
  • H02M3/10—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
  • H02M3/145—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
  • H02M3/155—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
  • H02M3/156—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
  • H02M3/158—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
• • H—ELECTRICITY
  • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
  • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
  • H02M3/00—Conversion of DC power input into DC power output
  • H02M3/22—Conversion of DC power input into DC power output with intermediate conversion into AC
  • H02M3/24—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters
  • H02M3/28—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC
  • H02M3/325—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal
  • H02M3/335—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only
  • H02M3/33507—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of the output voltage or current, e.g. flyback converters
  • H02M3/33523—Conversion of DC power input into DC power output with intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode to produce the intermediate AC using devices of a triode or a transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of the output voltage or current, e.g. flyback converters with galvanic isolation between input and output of both the power stage and the feedback loop
• • H—ELECTRICITY
  • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
  • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
  • H02M1/00—Details of apparatus for conversion
  • H02M1/0048—Circuits or arrangements for reducing losses
  • H02M1/0054—Transistor switching losses
  • H02M1/0058—Transistor switching losses by employing soft switching techniques, i.e. commutation of transistors when applied voltage is zero or when current flow is zero
• • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
  • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
  • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
  • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
  • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

• This present invention relates generally to a power converting appartuus, and more specifically to a DC-DC converter using an LLC circuit in the region of voltage gain above unity.
• Direct current (DC) architectures are well known, for example for the transmission and distribution of power.
• DC architectures generally provide efficient (low loss) diiiribution of electrical power relative to alternating c urrent (AC) architectures.
• DC architectures The importance of DC architectures has increased because of factors including; (1) the reliance of computing and telecommunications equipment on DC input power; (2) the reliance of variable speed AC and DC drives on DC input ,>ower; (3) the production of DC power by solar photovoltaic systems, fuel cells, and various wind turbine technologies; (4) propulsion systems in electric and hybrid vehicles, marine applications; (5) aerospace applications; (6) micro-grids and smart grids, including the above, energy storage and electric charging stations; and (7) other systems that require converters with varying input voltage and load.
• cost reduction is achieved in part by (1) reducing the components of DC-DC power converters, and (2) increasing the switching frequency of DC-DC power converters.
• These cost reduction methods can be achieved by implementing transformerless DC-DC converters that switch at high frequency. High frequency operation allows the circuit designer to reduce the size, and therefore the cost, of expensive components such as transformers, inductors and capacitors.
• Two of the most common transformerless DC-DC converters are the buck converter 10, as shown in FIG. 1 , for stepping down the voltage, and the boost converter 12, as shown in PIG. 2, for stepping up the voltage.
• For medium frequency applications (approx. 20kH3 ⁇ 4 - 100kHz) such devices are not leadily available thus they need to be created out of a series combination of an insulated- gate bipolar transistor (“IGBT”) and a diode, or a metal oxide semiconductor field effect transistor (“MOSFET”) MOSFET and a diode. This not only further increases system cost but it also nearly doubles the device conduction losses of the converter.
• IGBT insulated- gate bipolar transistor
• MOSFET metal oxide semiconductor field effect transistor
• Galvanic isolation and larger voltage boost and buck ratios are possibk with resonant and quasi-resonant DC-DC converters. These converters use inductive and capacitive components to shape the currents and/or voltages so that the switching losses are reduced allowing higher switching frequencies without a large efficiency penalty as explained in N. Mohan, T. Undeland, . Robbins, "Power electronics: converters, applications, and design," Wiley, 1995. Rt sonant and quasi-resonant DC-DC conveners can be implemented with or ithout galvanic isolation.
• a resonant converter with galvanic isolation is found in Bor-Ren Liu ami Shin- Feng Wu, "ZVS Resonant Converter With Series-Connected Transformers," ndustrial Electronics, IEEE Transactions on, vol, 58, No. 8, pp. 3547-3554, Aug- 2011, I this work, a series resonant converter is implemented with mu ltiple transformers connected in series.
• the proposed convener is designed to be ui ed as a power factor pre-regulator in consumer electronic applications.
• the converter operates near the characteristic frequency defined by the resonant capacitor and resonant inductor. ZVS is achieved for all of ihe input switching components.
• This conveiter analyzed by Bor-Ren Lin and Shin-Feng Wu uses a convencional resonant converter design approach.
• the resonant tank is only able to piovide minimal voltage boosting, if necessary, and any voltage boosting or bucking must come entirely from the iransformET turns ratio.
• the small amount of v illage boosting that can be provided is used when the input voltage is low. Furthei more, due to the resonant tank design, this converter would not be suitable to conti ol the power flow between an input and an output voltage source.
• Series resonant conveners and parallel resonant converters are known to bi very efficient for a small range of operating points. They can be implemented v ithout galvanic isolation or with galvanic isolation . For applications that require a large range of input voltages and loads, they are not ideal. As shown in B. Yang, "Topology Investigation for Front End DC/DC Power Conversion for Disb imped Power System", Ph.D. Dissertation, Virginia Tech, 2003, both series resonant conveners and parallel resonant converters suffer from large circulating currents, and large switching currents when the input voltage is high.
• a method of operating a resonant DC-DC converter comprising a high voltage boost L LC circuit, wherein the method comprises providing variable power flow control to the LLC circuit with externally determined input and output voltages using frequency control.
• the externally determined output voltage is created by cither a single externally determined output voltage, or a series connection of two externally determined output voltages to create a bi- olar output.
• a method wherein frequency control is applied such that it emulates different loading conditions thus operating along horizontal curves on the voltage gain compared Lo the switching frequency operating plane.
• the LLC circuit includes an LLC resoaant tank, and wherein the LLC resonant tank operates with a minimum boo; ting having an eiPfeclive value that is above unity over Lhe entire operating range.
• the minimum boosting results in controllable transfer of power via change of switching frequency.
• the method further comprises maintaining an externally determined voltage gain and using frequency control to enable movement between the load curves, and lo control this movement within a frequency control region where there is horizontal separation amongst the load curves.
• the method further comprises: (A) operating the high voltage boost LLC circuit in a region close to a resonant frequency determined by a resonant inductor, magnetizing inductor and a resonant capat itor, to achieve a high vohage boosL; and (B) utilizing unipolar or bipolar resonant tank excitation to improve converter efficiency in the high voltage boost circuit.
• a balanced bipolar DC output is pro ided wherein the output capacitor voltages are automatically balanced.
• the DC-DC convener further includes a resonant inductor, a magnetizing inductor and a resonant capacitor, and the method comprises the further step of selecting these components such that the yield over the entire range of operation is an effective voltage gain that is greater than unity.
• the LLC converter is implemented wiih a transformer to allow decoupling of the resonant circuit gain from the externally determined voltage gain.
• the effective voltage gain value and the components are selected so as to minimize the effective voltage gain of the resonant circuit, while being greater than unity, and provide controllability oi the DC-DC converter via frequency.
• the method further comprises operating .it a range of input stage switching frequencies in an LLC circuit whereby a c nge in input voltage results in a change in load or transferred power, such th t a decoupling between the input voltage and load is not required.
• a resonant DC-DC converter for I dgh voltage step-up ratio
• the resonant DC-DC converter for high voltage step- up ratio comprises: (A) a low voltage full-bridge or half-bridge DC-AC conve ter; (B) an LLC resonant tank; (C) a high voltage AC-DC converter or rectifier; and (D) a high voltage controllable switch; wherein the high voltage controllable switch is controllable to regulate power flow from an input to an output of the DC-DC converter based on an externally determined voltage gain ratio, wherein the LLC resonant tank operates with a minimum boosting having an effec tive value above unity over the entire operating range.
• the DC-DC converter is designed to provide variable power flow control using frequency control.
• a DC-DC converter wherein application of frequency control emulates different loading conditions hus enabling operation along horizontal curves on a voltage gain compared 10 a switching frequency operating plane.
• a DC-DC converter wherein the minimum boosLing results in controllable transfer of power based on ch nge of switching frequency.
• a DC-DC converter that maintains an externally detennined voltage gain, and uses frequency control to enable movement between the load curves, and controls this movement with in a frequency control region where there is horizontal separation amongst the load curves.
• a DC-DC converter is provided that is designed for: (A) operation of a high voltage boost LLC circuit in a region close to a resonant frequency determined by a resonant inductor, magnetizing inductor and a resonant capacitor, to achieve a high voltage boost; and (B) use of unipolar or bipolar resonant tank excitation to improve converter efficiency in the high voltage boost circuit.
• a DC-DC converter is provided that further comprises a balanced bipolar DC output wherein output capacitor volt iges are automatically balanced.
• a DC-DC converter that further includes a resonant inductor, a magnetizing inductor and a resonant capacitor, these components being selected such that the yield over tie entire range of operaiii >n is an effective voltage gain that is greater than unity.
• a DC-DC converter comprises a transformer to allow decoupling of the resonant circuit gain iron the externally determined voltage gain.
• a DC-DC converter if provided wherein the components are selected so as to minimize the effective voltage gain ol the resonant circuit, while being greater than unity, and provide controllability of the DC-DC converter via frequency.
• a method of designing a resonant DC-DC converter for high voltage boost ratio comprising: ( ⁇ ) a low voltage full-bridge or half-bridge DC-AC converter; (B) an LLC resonant tank; (C) a high voltage AC-DC convener or rectifier; and (D) optionally, a high voltage controllable switch; wherein the high voltage controllable swilcli is controllable to regulate power flow from an inpui to an output of the DC DC converter based on a externally determined input to output voltage gain : atio maintained by the high voltage controllable switch using frequency control, wherein the DC-DC converter includes (i) a resonant capacitor, (ii) a resonant inductor, and (iii) a magnetizing inductor; wherein the design method comprises: (i) determining a minimum gain sufficient to enable high-resolution coiitrc l of frequency using available control hardware; (ii) selecting an
• a method of designing a resonant DC -DC converter lor high voltage boost ratio comprising: (A) a low voltage full-bridge or half-bridge DC- AC converter; (B) an LLC res ⁇ mailt tank; (C) a high voltage AC-DC converter or rectifier; and (D) optionally, a high voltage controllable switch; wherein the high voltage controllable switi h is controllable to regulate power How from an input to an oulput of the DC-DC converter based on a externally determined input to output voltage gain ;-atio. Power flow control is maintained using frequency control.
• the DC-DC com ertcr includes (i) a resonant capacitor, (ii) a resonant inductor, and (iii) a magneiizing inductor; wherein the design method comprises; (I) determining a minimum gain sufficient to enable high-resolution control of frequency using available control hardware; (2) selecting an IVL r ratio that is suitable for an application fo ⁇ the DC-DC converter; (3) generating voltage gain curves for various values of Q, and plotting these values so as to graph a boundary curve that defines LHS and RHS regions, and selecting the Q values whose voltage gain curve intersects ith boundary curve at the maximum voltage boost ratio, thereby defining a se t of normalized frequency values; and (4) using the Q values and the normalized frequency values found to calculate values for the resonant capacitor, the resonant inductor, and the magnetizing inductor so as to enable selection of suitable components for the application.
• the invention is capable of operating with other resonant converter configuration known in previous art and/or used in diffej ent applications. It is also understood that the invention is usable in applications v ith different grounding requirements including floating systems, high impedance grounded systems, and solidly grounded systems and that the use or not of a transformer may be influenced by the grounding requirements, fn this respect, before explaining at least one embodiment of the invention in detail, it is to be understood that the invention is not limited in its application to the details of construction and to the arrangements of the components set forth in the following description or illustrated in the drawings. The invention is capable of other embodiments and of being practiced and carried out in various ways. Also, it is to be understood that ihe phraseology and terminology employed her « in are for the purpose of description and should not be regarded as limiting. HJRIEF DESCRIPTION OF THE DRAWINGS
• FIG. 1 is a circuit diagram illustrating a prior art buck converter.
• FIG. 2 is a circuit diagram illustrating a prior art boost converter.
• FJGS. 3(a), 3(b) and 3(c) illustrate three representative implementationii of the half-bridge resonant DC -DC converter, having a single high voltage switch.
• FIGS. 4(a) and 4(b) illustrate an implementation of a full-bridge resonant DC-! C converter.
• FIGS. 5(a), 5(b), 5(c) and 5(d) illustrate four representative implementations of the full-bridge resonant DC-DC converter of the present invention, having a sin gle high voltage switch and a common ground on the input and the output.
• FIGS. 6(a), 6(b) and 6(c) illustrate the three representative circuits of an alternate implementation of the circuit design of the present invention that include transformer.
• FIG. 7 is the implementation of FIG. 6(c), using MOSFET switches, with he addition of a anubber diode.
• HG. 8 illustrates the voltage and current waveforms associated in operation w ith the circuit of FIG. 7.
• FIG. 9 is a specific implementation of the half-bride resonant DC-DC converter of FIG.3(a) using a combination of MOSFET and IGBT switches.
• FIG. 10 illustrates the voltage and current waveforms associated in operation with the circuit of FIG. 9.
• FIG. 11 is a specific implementation of the full-bridge resonant DC-DC convener of FIG.5(a)using a combination of MOSFET and IGBT switches, with the addition of a snubber diode.
• FIG. 12 illustrates the voltage and current waveforms associated in operation with the circuit of FIG. 11.
• FIGS. 13(a) and 13(b) are circuit diagrams illustrating alternate implementations of the full-bridge resonant DC-DC converter of the present invention, wiLli a common ground for the input and the output, but without a high voltage switch.
• FIG. 14(a), 14(b) and 14(c) arc a circuit diagrams illustrating a possible LLC converter circuit designs, that are (A) operated in a novel and innovative vay based on the methods of the present invention, and (B) redesigned also as described in this disclosure.
• FIG. 15 illustrates a classic LLC circuit equivalent model used for First Harmonic Approximation (FHA) analysis.
• FIG. 16 illustrates the votlage gain, computed from FHA, achieved with an LLC circuit topology with various loads over a large range of switching frcquencies-
• FIG. 17 illustrates the voltage gain, computed from FHA, achieved with an LLC circuii topology with various loads over a large range of switching frequencies with the conventional region of LLC operation and the Interrupt Switch Coni rol Region denoted.
• FIG. 18 illustrates the voltage gain achieved with an LLC circuit topology w ith various loads over a large range of switching frequencies with a number of operating regions denoted: (1) RHS Operation region, (2) LHS Operation region, (3) Conventional Opearation region and (4) the LHS/ HS Boundary curve.
• FIG- 19 jllustrates a voltage gain graph of the LLC converter operated at constant gain with power flow regulated by adjusting the switching frequency.
• FIG. 20 illustrates how conventional LLC resonant converters control their voltage boost, and therefore, their power flow.
• FIG. 21 illustrates an LLC tank current waveform using an interrupt switch in accordance with another aspect of the present invention.
• FIG. 22 illustrates an LLC tank current operating near lull power in accordarice with another aspect of the present invention.
• FIG. 23 illustrates an LLC tank current operating at low power in accordance with another aspect of the invention.
• FIGS. 24(a), 24(b), 24(c) and 24(d) illustrate four representative implementati ons of the resonant DC-DC converter of the present invention with extern illy determined input and output voltages and no interrupt switch, where FIGS. 2 (b) and 24(c) further illustrate representative bi-polar output configurations and 1A (d) further illustrates a possible implementation with auto-balancing bi-polar out ut voltages.
• FIG- 25 illustrates the voltage gain achieved with an LLC circuit topology vith various loads over a large range of switching frequencies with LI-IS/RHS Boundary curve denoted.
• FIGS. 26(a) and 26(b) show the general form of the current invention with ;md without an interrupt switch.
• FIG. 27 illustrates voltage gain curves for the LLC converter focused around a voltage gain of 4 in accordance with an embodiment.
• FIG. 28 illustrates a final converter design in accordance with an embodime nt, with the region of operation identified.
• the present invention describes a number of innovations related to the subject matter of the Base Application.
• the present invention includes (A) a novel and innovative resonant DC-DC converter that employs a high boost resonant tank to enable power flow control between externally determined input and output voltages using frequency control, with or without use of an interrupt switch (I he "Improved DC-DC Converter"), (B) a method of operating a resonant DC-DC converter to achieve high boost resonant tank operation, which is suitable for improving the performance of resonant converters based on different topologies ("method of operation"), including but not limited to the Improved DC-PC Converter; and (C) a metliod for designing DC-DC converters (having different topologies) for improved performance using the method of operation ("design method").
• A a novel and innovative resonant DC-DC converter that employs a high boost resonant tank to enable power flow control between externally determined input and output voltages using frequency control, with or without use of
• the design method includes identification of circuit design parameters that enable use of the method of operation. Performance improvements include improved resolution of power flow control between externally determined input and output voltages a d maximization of range of allowable voltage conversion ratios while meeting a specified power flow. Operation of the converter ove> a reduced range of frequencies may also allow circuit components to be beuer optimized for efficiency. In full-bridge embodiments, as exemplified in FIG "4, the use of uni-polar/bi-polar operation may allow enhancement of circuit efficiency over portions of the operating range. In embodiments such as FIG 24(d) the provision of a bi-polar auto-balancing output voltage eliminates need for advanced sensing and controls to achieve voltage balancing in the bi-polar output.
• use of the embodiment of FIG 24(d) for solar photovoltaic applications enables the design of a high voltage boost DC-DC Converter that offers inherent safety through low- voltage operation of the photovoltaic modulus, distributed control for increased energy yield, high-conversion efficiency ovei a wide range of input voltages and power flows, integrated galvanic isolation to isolate faults, use of long-life film capacitors, and full utilization of the AC gi id interconnection inverter.
• the converter circuit topologies may include a resonant tank and (in one aspect) a means for interrupting die t mk current to produce a near zero-loss "hold” state wherein zero current and/or icro voltage switching is provided, while providing control over the amount of power transfer.
• the converter circuit topologies may control energy tran fer by controlling the duration of the near zero-loss "hold”. This may be referred t> > as the "interrupt control mode" (again, shown for example in Figs. 19 and 21) This energy power transfer control may be achieved using a single high vol! ge controllable switch.
• the present invention may avoid unnecessary circulating current during low power operation, thereby reducing losses within the tank components and the low voltage DC/ AC converter, and also reducing switching losses based on the iero voltage switching of the low voltage DC/AC converter and zero current switching of the low voltage DC/AC converter. Also, 2£ro current switching of the Mgh voltage controllable switch within the tank may be achieved and thereby keep its own switching losses low.
• the present invention may have several embodiments ihat present con verier circuit topologies that provide high input-to-output vollage conversion and achieve high efficiency operation. Examples of tliese embodiments are disclosed herein; however a skilled reader will recognize ⁇ these examples do not limit the scope of the present invention and that oi her embodiments of the present invention may also be possible.
• the term "low voltage” is used in this disclosure to refer to components with voltage ratings comparable to that of the input, and the t ;rm “high voltage” is used in this disclosure lo refer to components with voltage raiing comparable to, or above, the peak voltage level seen across the resonant unk capacitor.
• appropriate implementation of the nea zero-loss hold state may cause zero voltage switching or zero current switching to be achieved lor all controllable switches within the circuit.
• Embodiments of the present invention may provide a lower loss convener circ uit for high input-to-output voltage conversion ratio converters.
• circuit design of the present invention may include a variety of elements. Jn one embodiment these elements may include: (1) an input DC/AC converter; (2) a resonant tank; (3) a Lank interruption means (such as a switch as described herein); and (4) an output rectifier.
• T e output rectifier may, for example, include a filter inductor that limits the rate of rise of current in the output diode.
• the input DC/AC a skilled reader will recognize that a number of different types of inverters may be suitable, for example, such as a half-bridge or full-bridge type inverter.
• the output rectifier may include any output rectifier staj'.e, for example, such as a half-bridge or full-bridge rectifier.
• a transformer may be included in the circuit, prior to the output rectification stage.
• the circuit design may be a circuit that includes: (1) a full-bridge DC/AC converter; (2) a resonant tank consisting of two L components and one C component; (3) a tank interruption switch; and ( 1) an output rectifier stage (full-bridge or half-bridge), wherein a common ground may be provided for both the input voltage and the output voltage.
• lhaL include such a circuit design are shovn in FIGS. 5a to 5d.
• the circuit may, or may not, include a transformer. In an embodiment of the present invention wherein a full-bridge output rectifier is utilized a transformer may also be required.
• the resonant L components may be integrated into the transform er design.
• the choice to include a transformer in an embodiment of the present invention may be based on specifications of the circuit of the embodiment of the present invention, or other preferences or considerations. This document discloses and describes some examples of both: embodiments of the present invention (hat include a transformer clement; and embodiments of the present invention that do not include a transformer element, and therefore are transformerless.
• FIGS. 6(a), 6(b) and 6(c) show embodiments of the present invention that ire circuits 42, 44 and 46 respectively, that include an alternate implementation, wherein additional windings were added to the main inductor's magnetic core thus decreasing the voltage stress on switch S*.
• windings may convert the inductor L into a transformer with isolation, which provides additional circuit implementation options.
• the embodiment of the present invention shown in FIG. 6(c) may provide bipolar output to allow a differential output voltage of 2 x V 2 to be achieved while maintaining a voltage to ground at level V2.
• a circuit 48 may be one practical implemeniittion of ihe circuit shown in FIG. 6(c).
• the transformer magnetizing branch may provide I he main resonant tank inductance "L”.
• the filter inductance "Lr” may also be integrated into the transformer. This may be done by designing the transformer to have leakage inductance of value "Lr”.
• Vs shown in FIG- 7, all switches may be implemented using MOSFETs.
• a snubber circuit may be employed to limit the transient voltage across the high voltage MOSFET at the end of the conduction period. Provided the voltage V 2 is low er than the voltage rating of the high voltage MOSFET, the snubber may consist o!
• embodiments of the present invention may produce particular results 50 that include gating signals for the converter of FIG- 7, together with tlie important voltage and current waveforms.
• Switch S x may turn off at the same time as Sj and S2p, though ihc MOS1FET body diode may allow conduction of the negative current. If losses in the MOSFET conduction channel are calculated to be lower than body diode conduction Josses, then the MOSFET should be kept on for ihe duration of the negative current pulse to reduce conduction losses.
• the duration of the hold state may be varied to control the amount of average power transfer from input to output. Following the hold sl ile another similar cycle of operation may follow.
• Transfer of power from the resonant tank to the output may occur twice per period, once to the positive DC output, once to the negative DC output.
• Pov er transfer to the positive output may take place immediately after the turn on of Switches Si and S3 ⁇ 4,.
• Power transfer to the negative output may lake e immediately after the tum on of switches S 2 and S lp .
• a circuit may be provided consisting of a DC-AC converter followed by a (parallel) resonant tank with single controllable high voltage switch, followed by an AC-DC converter.
• FIGS. 3' ), 3(b) and 3(c) are shown in FIGS. 3' ), 3(b) and 3(c) in three specific representative implementations.
• the embodiment of the present invention shown in FIG. 3(a) may be a circuit 14 that does not include an output filter inductor.
• FIG. 3(a) illustrates the basic circuit design concept of the present invention, and presents a half-bridge floating lank convei ter in accordance with the present invention.
• the embodiment of the present invention shown in FIG. 3(b) may be a circuit 16 that includes an output filter inductor. For most implementations of the invention, it is a practical requirement to include a filter inductor.
• FIG. 3(b) shows an embodiment of the present invention that may be a circuit 18 that includes a filter inductor integrated in the tank.
• the circuit 20 may be a "full-bridge floating tank" configuration of the circuit design illustrated in FIGS. 3(a), 3(b), and 3(c).
• FIG. 4(a) may be extension of the conve er illustrated in FIGS. 3(a), 3(b) and 3(c).
• FIG. 4(a) may be extension of the conve er illustrated in FIGS. 3(a), 3(b) and 3(c).
• Embodiments of the present invention may represent variants of the full-bridge resonant DC-DC converter of the present invention, and may include a single high voltage switch, and a common ground for the input and the output More specifically: the embodiment of the present invention shown in FIG.
• the embodiment of the present invention shown in FIG. 5(a) may be a circuit 22 wherein the indue tor current may be switched by the single high voltage switch (S ⁇ ); the embodiment of the present invention shown in FIG. 5(b), may be a circuit 24 wherein the capacitor current may be switched by the single high voltage switch (S x ); the embodiment of the present invention shown in FIG. 5(c), may be a circuit 26 t at is similar to the circuit 22 shown in PIG. 5(a), and the circuit 26 shown in FfG. 5(c) may include an inductor current that may be switched by S x and the fi lter inductor may be integrated into the tank; and the embodiment of the present invention shown in FIG.
• the DC-DC converter of the present invention may display a significant degree of asymmetry.
• the asymmetry may be displayed in that the grounding is asymmci tic, the input switch configuration is asymmetric, and the output stage is asymmetric.
• an embodiment of the present invention may use emerging reverse block IGBT devices, in which case S x may be eliminated, but Si and S2 may each need to consist of a high voltage reverse blocking IGST.
• Such an embodiment of the present invention may yield precisely the sii e voltage and current waveforms within the tank and output circuitry. Numei ous other variations are possible.
• the circuit design may be such thai the high voltage switch needs not be reverse blocking, and thus MOSFETs or IGliTs may be used instead of, for example, thrysitors (which limit Switching frequencies to excessively low values), or MOSFET-series-diode / IGBT-series-diode combiuations-
• the circuit designs may usi an electrically floating tank, as further explained below. Certain aspects of the invention are explained in greater detail below, however these details should not be read as limiting the scope of the invention in anyw iy, but as examples of embodiments of the present invention.
• the half-bridge floating tank converter may be included in embodiments of ihe present invention.
• the switching process may vary slightly based on the type of switches used and ihe location/orientation of the high voltage switch (S x ) within the tank circuit.
• a description of a possible switching process to be used in an embodiment of ihe present invention is provided herein with reference to a topology 30 wherein $
• waveform results 32 of use of the embodiment may show particular voltage and current waveforms associated with a half-bridge floating tank converter.
• ihe converter may operate in a mode where the inductor current is not continuously oscillating but is interrupted, once each period, by the single high voltage switch,
• Si and S x may fire to begin one cycle of LC resonant oscillation.
• the initial condition on the capaciior voltage may be approximately -V2.
• Current II may be positive and input voltage V jn may be positive for haif a cycle, transferring energy into the circuit.
• the output diode conductors and II may be transferred to the output, accomplishing output power transfer (the ra
• Si At zero crossing of the input current Si may be turned off and S2 may be turned on.
• the output diode may turn off at this time and the IGBT reverse conducting diode may turn on at this time. This allows the tank oscillation to continue, thereby recharging the capacitor to -V2, in preparation for he next cycle.
• the IGBT may be in an "off' iBtate, thus interrupting the tank oscillation at a current zero crossin;;. 6.
• the circuit may then in a 'hold state' until a new pulse of energy is required.
• Embodiments of the present invention may include a full-bridge floating tank converter with common ground, as shown in FIG. 1 1.
• the switching process may vary slightly based on the type of switches used and the location/orientation of the high voltage switch (S x ) within the tank circuit.
• One embodiment of the present invention include a fulJ-brid3 ⁇ 4e floating tank converter with common ground may include a topology 34 where ine four switches Sj, S] P , 3 ⁇ 4 and S 2p are implemented using MOSFETS and is implemented using a high voltage IGBT, as shown in FIG. 11.
• a snubber circuit may be employed to limit the transient voltage across the high voltage MOSFET at the end of the conduction period.
• the snubber may consist of a single diode from the collector of the IGBT to the output. This may allow energy normally lost in snubber circuitry to be transferred to tlie output, thereby yielding a near lossless snubber. Such embodiments of the present invention may improve overall converter efficiency.
• wavefoi m results 36 of use of the embodiment may show particular voltage and current waveforms associated with this a full-bridge floating tank converter with common ground.
• the converter may operate in a mode where the inductor current js not continuously oscillating but is interrupted, once each period, by the single high voltage switch, S x .
• An example of the operation of the circuit may be as follows:
• Si, Si p and S x may fire to be nn one cycle of LC resonant oscillation.
• V- K When V- K reaches power may begin being transferred to the output.
• Capacitor voltage may then be in a 'hold state' until a new pulse of cnei3 ⁇ 4y is required.
• Embodiments of the present invention may include a full-bridge floating tank converter with common ground that is operable to transfer energy during both positive and negative half cycles of the tank current, without use of a transformer, while maintaining a common ground on input and output, as required for many applications.
• the purpose of S in this circuit may be to achieve zero current/zero voltage switching while still offering control over the amount of power transfer. Thus near z ro switching loss may be achieved while simultaneously maintaining control over the amount of power transfer.
• FIGS. 13(a) and 13(b) accomplish tins. These topologies may be related to the circuit designs shown in FIGS. 5a and 5c.
• silicon carbide devices may offer grcady reduced switching losses (esp. the elimination of diode reverse recovery current), maintaining zero current zsro voltage switching may be sacrificed without negatively impacting efficiency. Power transfer may then be achieved via frequency control, as is common in ot er resonant converters, see: R. Erickson, D. Maksimovic, "Fundamentals of Po ⁇ ver Electronics,” Kluwer Academic Publishers, 2001.
• the full-bridge converter with common ground may offers important benefits compared to the conventional resonant converters as outlined in R. Erickson, D. Maksimovic, "Fundamentals of Power Electronics,” Kluwer Academic Publishers, 2001.
• the topology of an embodiment of the present invention ihat includes a full-bridge converter with common ground may offer common ground on input and output along with a high step-up ratio and may offer power Iran -fer into the tank during both positive and negative half cycles of the tank current.
• Embodiments of che present invention that include a half-bridge floating l ink converter may offer particular benefits over the prior art. Some of these bene fits include the following:
• the half-bridge circuits of the present invention may only use one high voltage device, labelled: S x . Furthermore S may not need to be a reverse blocking device.
• a single high voltage switch may be operable in embodiments of the present invention to interrupt the resonant operation of the conveicer, thereby controlling energy transfer.
• Si and Si may be implemented in embodiments of the present invent ion using, only low voltage components, reducing losses.
• embodiments of the present invention may only require a single source and sing/e tank inductor.
• Embodiments of the present invention may provide zero cmrentAtero voltage switching of the input AC-DC converter.
• Embodiments of the present invention that include a full-bridge floating tink converter with common ground may offer particular benefits over the prior irt. Some of these benefits include the following:
• the circuit of embodimf nts of the present invenLipn may operate using only one high voltage device, labeled S*. as shown in FIGS. 3(a), 3(b). 3(c), and 3(d). Furthermore S K may not need to be a reverse blocking device. .
• S* high voltage device
• the full-bridge DC-DC converter of embodiments of the present invention may provide rou ldy double power transfer since energy may be transferred from the souice into the tank during both positive and negative half cycles of the tiink current.
• Embodiments of the present invention may provide zero currcnt ⁇ ⁇ voltage switching of the input AC-DC converter.
• common ground may be provided between the input voltage source and output voltage source.
• a single high voltage switch may be operable to interrupt the resonant operation of the converter, then by controlling energy transfer.
• circuit designs of embodiments of ihe present invention may present a modular structure and therefore components m y be added or removed, while providing the Functionality of the design, as described above
• particular embodiments of the DC-DC converter of >he present invention may be transformerless.
• a transformer could be included betwe en either the resonant tank inductor or resonant tank capacitor and the diode rectif ier in the circuit shown in FIG. 4(b).
• the switching elements may employ silicon carbide devices. Switching may be carried out to provide a square wave voltage switching between +V1 and -VI lo the tank circuit. The switching carried out to provide a square wave voltage may be switching between +V1 and 0 (or between 0 and -VI) to the lank circuit. Tank input voltage switching may occur between +V1 and -VI when operating near rated power and between +V1 and 0 (or between 0 and -VI) under low power.
• the elements recited in this paragraph may be used in a topology where the inductor Lf is moved to the output path (such as is shown in FIG. 13a).
• the inventors have realized DC-DC converters may be provided that include improved performance characteristics of the DC-DC converters disclosed above, however, without the interrupt switch disclosed in the Base Patent.
• this is achieved by employing a number of concepts including: (i) achieving a high boost through the systematic design oi a resonant tank; (ii) enhancing converter efficiency using a unipolar/bipolar resonant tank excitation; and (iii) employing an output configuration with automatic voltage balancing on output capacitors in conjunction with the high boosting resonant tank circuit to yield a high step-up ratio and a balanced bipolar DC output voltage.
• FIG. 24(a) An illustrative example is shown in FIG. 24(a) where an LLC converter circuit design in accordance with an embodiment does not require a high voltage switch.
• C represents a resonant capacitor
• Lr represents a resonant inductor
• Lm represents a magnetizing inductor.
• the "Classic" LLC Circuit DC-DC converter topologies shown in FIG. 14 have been studied in literature (R.L. Lin et. al. and H. Hu et. al., above), however most of the prior ait is related to step down (buck) realizations of the technology.
• Tlus type of converter design is commonly used in conventional applications where t tie output voltage is independent of the load, such as a power supply.
• the classic LLC circuit topology offers advantages compared to otl icr circuit topologies.
• the voltage g;iin characteristics of the LLC converter can be approximated using first harmonic approximation (FHA) techniques. Assuming the circuit is stimulated by a perfect sinusoid, one can use conventional circuit analysis to determine the voltage gain of the circuit.
• FHA first harmonic approximation
• An LLC converter as illustrated in FIG. 14(a) for example may be simplified to provide the circuit shown in FIG. 15 here
• the voltage gain can be then calculated for different loadings and frequencief to produce the plots shown in FIG. 16-
• This figure shows the voltage gain achieved by an "LLC Resonant Tank", as a function of normalized switching frequency of the input stage DC-AC converler.
• f r i and fa The resonant frequencies of the circuit are defined by f r i and fa, defined below:
• a Classic LLC Circuit In conventional applications, such as power supplies, a Classic LLC Circuit is generally operated near f rI as indicated in FIG. 17 by the box titled, "Conventional Region of Operation", because a constant output voltage is desired throughout tlie entire load range.
• the desired ratio between the input voltage and output voltage is predominandy achieved using a transformer in the output stage, and not the LLC Resonant Tank itself.
• the output voltage When the input voltage changes the output voltage is maintained at a constant level by adjusting the switching frequency of the in nut stage above or below f,i-
• the value of Q may not critical to the operation of ihe circuit and it may only be verified that the circuit can provide the required out ut voltage for the maximum load- Values of Q close or even higher than 1 -ire common in conventional circuits.
• the Classic LLC Circuit topology can be operated over a frequency range well below f r] (f r j is not within the operating range) by selecting the components such that the value of Q is well below 1 for the full load range specified. Furthermore, the circuit has not been used in applications that require control of the power transfer between two regulated or unregulated DC sources.
• the LLC topology is designed to operate with switching frequencies well below f r j, close to the second resonant frequency of the circuit, fr f ).
• Operation in the area near f,o can be divided into two distinct operating regions as shown in FIG. IS- As shown in the figure, the two regions are named the “LHS Operation” and “RHS Operation” regions. The line which intersects both of these regions is called the “LHS RHS Boundary", which is also shown in Fl.G. 18.
• Operation in the "LHS Operation” region yields zero current switching (ZCS), suitable for switching devices such as IGBTs.
• ZCS zero current switching
• Operati m in the "RHS Operation” region yields zero voltage switching (ZVS), suitable ior switching devices such as MOSFETs. Operating in any one of these regions yields a voltage gain above I for loads with Q lower than 1.
• the values of tne resonant tank components can be selected such that the Q value lower than 1 cm be achieved for all load values (power transfers) required to be handled by the converter. This Q value would be lower for higher voltage boosting requiremenis.
• the system would then operate at a switching frequency below t ' -i for all steai ly state operating conditions.
• a resonant tank circuit designed in accordance wirh this embodiment will be called a "High Voltage Boost Circgir (HVBC). All >f the embodiments of the invention shown above use a HVBC resonant tank circuit design. The introduction of a transformer to the circuit does not alter the high boost nature of the tank design.
• che range of input voltage and the range of load is known.
• the output voltage is also known based on components to be powered by the converter or the externally regulated voltage bus that is to receive power.
• what follows is a possible method for designing circuits based on said LLC topology, but providing relatively high boost ratios-.
• Ln L t ratio that is suitable for the application. Typical values range from, but are not limited to, 3-10. Large values will result in higher pi ak currents in the tank, while small values will result in larger switching losses at low loads.
• step 4 Using the Q and normalised frequency values found in step 3, calculate ihe I- T and C r values.
• the first method discovered to achieve controllability of the above design was the introduction of an interrupt switch in the LLC Resonant Tank (tUe "Interrupt Switch LLC Circuit").
• the interrupt switch allows the Q value to i>e solely dependent on the input voltage and not the load. As the input volta ;e increases, the Q value decreases.
• the Input Stage switching frequency of t ie circuit is used to compensate for changes in the input voltage and the off time of the interrupt switch is used to adjust to the changes in load.
• the decoupling of the load (using the interrupt switch in the LLC Resonant Tank) from the input volfcige (using the Input Stage switching frequency) allows for a simple implementatioti of a controller and stable control.
• the introduction of an interrupt switch into the LLC Resonant Tank also enables the use of the Interrupt Swkch LLC Circuit in new applications where the LLC Resonant Tank is operated in ihe conventional region of operation close to f
• the use of the Classic LLC Resonant Circuit in this operating region is not easily realizable with the classic frequency control method.
• the Interrupt Switch LLC Circuii is suited to new applications where the objective of the LLC circuit is not to regul ite the output voltage but instead to regulate the power delivered to an output voltage regulated externally.
• Ihe LLC resonant tank is operated so as to l ie given sufficient boosting gain, a change in either the input voltage or the switching frequency results in a corresponding change in load (power transfer).
• FIG. 1 for one possible circuit design that is adapted to deliver minimum boosting as described.
• FIG. 19 illustrates maintenance of a fixed voltage gain of 2.0 wl iiie using frequency control to enable movement between the load curves.
• FIG. 18 for example in the "frequency control regions" there is horizontal separation amongst the load curves.
• Operation of the LLC Resonant Tank on a maintained basis in the frequency control regions suitably above unity g iin enables better control of power transfer based on switching frequency, while maintaining the boosting ratio shown in FIG. 19.
• This provides the reduced frequency range of operation required to control :he load, and chopping of much smaller currents than conventional non-boosting D circuits.
• components of a DC-DC converter desigr ed to embody the mode of operation described may be selected so as to improve performance within the frequency range described.
• the objective of the design method of the present invention is to provide a DC-DC converter that is designed so that the boosting gain is above unity.
• the boosting gain can be designed as close to unity as desiied provided a frequency controller with an infinite frequency resolution.
• Practic al implementations of the converter which use frequency controllers with a Finite resolution will require a minimum boosting gain above unity which achieves tlie desired controllability, i.e., the desired power flow resolution.
• tlie desired controllability i.e., the desired power flow resolution.
• a boosting gain of 1.25 may be practical to mainta in power flow controllability with practical power flow resolution over tlie entire operating range.
• FIG. 18 shows the typical voltage ga that can be achieved with an LLC circuit.
• the different lines in FtG. 18 represent the same tank circuit with different loads. The lines then trace nit the voltage gain from the converter when operated from about 0.4 times ihc resonant frequency f r] to 1.2 times the resonant frequency f,i .
• LLC power supplies are designed to operate near the reson tnt frequency defined by the resonant inductor and resonant capacitor, f rl .
• This region of operation can be seen in FIG. 17 with the resonant frequency f r i denot d.
• the circuit When operated in this region near f,j, the circuit will exhibit constant voltage gain throughout the entire load range.
• FIG. 17 also shows the Interrupt Switch Control region, which covers parts of the conventional region of operation.
• the LLC is designed such that it is operating very close to the resonant frequency determined by the resonant inductor, magnetizing inductor and the resonant capacitor, which will be referred to as In FIG. 18, this operating region is outlined and labelled "LHS Operation” a id " HS Operation".
• the circuit is able to achieve hi . ;h boost ratios yet also achieve a reduced switching loss throughout a wide lo id range.
• Output power is controlled by varying the switching frequency, whi h need only be varied by about 20% of the resonant frequency. It will also be appreciated that the regions of operation as defined by FIGS.
• FIG. 18 The "LHS Operation” and “RHS Operation” regions are focused around
• PIG- 20 shows the current waveform flowing out of the switching network in a
• the interrupt switch waits until negligible current is flowing in
• circuit is operating approximately on the ZCS ZVS boundary shown in FIG. 18. at
• FIG. 22 shown is a proposed mode of control over the prcsenily
• the waveform will resemble the full power waveform of the circuit with the interrupt switch, with switching happening
• the switching frequency are necessary to tegulale power from full load to zero load.
• the switching frequency is increased from 55-5 kffc to 59.7 kHz and the power transfer is reduced by about 25%.
• FIG. 24(b) without transformer, 24(c) wiih transformer and 24(d) w ith transformer and one possible implementation with auto-balancing output voltage.
• the waveforms are shown in FIG. 22 for full load and FIG. 23 for partial load:
• the resonance will reduce the voltage in C r and will increase the current in the inductor Lr.
• Lm has a constant voltage equal to V o] across it. T he voltage across C r will turn negative and the current across Lr will st irt decreasing.
• Switches Si and ⁇ ⁇ are then turned off; almost immediately thereafier switches S2 and Si p are turned on. This commences the second half eye le which is symmetrical to the first.
• the length of the switching period may be varied to control the power flow through the converter.
• the above control descriptions are based on the circuit using a rull bridge DC-AC converter and the split output circuit, a person skilled in the art could be able to identify that the general operation- is similar in other embodiments. Differences in the number of pulses transferred per period, ihe type of load receiving the power pulses, or the location of the components used to produce the resonance amongst others do not change the operation principles for the circuit.
• the benefits of the circuit over the classical LLC converter control are: (i i a significantly longer switching period (approximately 2 times) for a given set of components; (ii) a reduction in switching losses; (iii) a reduction in losses within the resonant tank (comprised of C r , L- and L ra ); and (iv) the ability to regul ite power transfer between two externally determined DC sources.
• switching of the DC/AC converter may be carried out such that the DC/AC converter output is either an AC waveform of +V1 and -VI, or an AC waveform of either VI and 0 or -VI and 0.
• the ability to switch betwe en these modes of operation will be called "Unipolar Bipolar Resonant Tank Excitation Control”.
• Unipolar/Bipolar Resonani Tank Excitation Control changes how the resonant rank is excited in order to operate the converter in its must efficient control mode for a given input power.
• an embodiment of the invention includes a bi-polar output voltage. This configuration is advantageous since the maximi m voltage to neutral is reduced by a factor of two. As a consequence, cabling with a lower insolation class can be used, reducing the cost of wiring the converter. Tlie use of two voltage sources to create the bi-polar output ensures that tlie output of the converter i always balanced to the neutral point. Auto Balancing Output
• an embodiment of the invention includes a voltage doubling rectifier, which creates a bi-polar output.
• This bi-polar output must be balanced in order to properly maintain the output DC link.
• the output capacitors, Co in FIG. 24(d) ire automatically balanced.
• the operating point of the converter moves vertically down the curves shown in FIG. 18. Moving down these curves corresponds to a higher Q value, or larger load. A larger load means more power will be transferred, which will in turn charge the capacitor back to its nominal operating voltage. No other control circuitry is needed.
• the focus of the present embodiment is on a unique mode of operation that yields a large voltage boost in the resonant tank.
• This voltage bo )st allows the present HVBC embodiment to achieve very high efficiencies at high conversion ratios.
• the resonant tank of an LI ,C converter can be designed lo yield high voltage gain, useful for step up convert? rs.
• the converter can be operated with a low Q over the entire load ran;;e. This is achieved by knowing the load, and designing the resonant components around it.
• the resonant tank can be stimulated near the reson.mt frequency and operation of the converter in this region yields to ZVS, and lo current switching (LCS), to yield a highly efficient, step up converter.
• This mode of operation makes is viable for the converter to transfer power between two externally determined voltage sources.
• the current, I r seen by the input ac source, the capacitor C r and the inductor L- therefore has two components:
• I M itself transfers no power to the load, it is merely required to enable the process of energy transfer.
• I c comprises a large percentage of I r , leading to highly efficii ut operation.
• the interrupt switch enables a high Ic lo I r ratio to be employed under all loading conditions. At full load the I e to l r ratio is high by its very nature, posing no challenge. To operate at reduced load the interrupt switch introduces a near zero loss hold stale. This yields an efficiency that is roughly independent of loadmg conditions- It should also be noted that each time the convert leaves the hold stile one pulse of energy is transferred to the output. For a given input and output voltage the size of this energy pulse is constant. Power transfer is controlled by merely regulating the number of energy pulses that are released by the intemipt switch.
• FIG. 22 shows a comparison of where the interrupt circuit operates versus wtn:re the frequency contiol circuit operates for a Fixed V 6 to V 0 ratio of 1:2. Note Llial only one point is shown for the interrupt circuit operation.
• the interrupt switch pulses the power to the output always at one point on this plane. By conuolling the pulse density the amount of power transfer is linearly controlled.
• Such applications include, but are not limited to, solar photovoltaic systems, fuel cells, permanent magnet wind turbines, electric and hybrid vehicles, electric charging stations, aerospace applications, marine applications, crohn o- grids, energy storage and other systems that require converters with varying input voltage and load.
• the interrupt switch topology is used in two main applications: 1. In applications where a high efficiency is desired and the converter operates at low power for long periods of times, such as standby pcwer applications.
• the theoretical gain of the LLC convener can be approximated using l irst harmonic approximation (FHA) techniques. Assuming the circuit is stimulated by a perfect sinusoid, one can use conventional circuit analysis to determine the voltage gain of the circuit.
• the phase of the resonant current determines the region of operation of the converter. For example, if the resonant current is leading the input voltage, the LLC converter is in the "LHS Operation” region. Conversely, wlien the resonant current is lagging the input voltage, the converter is in the "KHS Operation” region.
• the border between the two regions is where the resonant t- ⁇ nk behaves like a perfect resistor. The dashed line in FIG. 18 shows this border.
• the circuit For voltage boosting applications, the circuit must be designed such diat it (an operate with voltage gains greater than 1, In FIG- 18, this is achieved by designing the converter around a low Q value. As shown, lower Q values provide a larger voltage boost at the output. In addition to a low Q value, the convener will be operated at switching frequencies closer to the dashed line. Turse observations are in contrast to traditional LLC designs, where the converter is designed with larger Q values and operated near the resonant frequency, f 0 . Designs that follow these traditional constraints exhibit unity voltage gain for .ill loads.
• this particular example of a converter requires a maximum gain of 4 based on the voltage that converter will be exposed to. Therefore, the method enables the determination of the resonant components that will yield che required maximum voltage gain, while operating in the LHS region. A skilled reader will appreciate that maximum gain drives the circuit design.
• L ni /L- ratio that is suitable for the application. Typical values range from, but are not limited to, 3-10. Large values will result in higler peak currents in the tank, while small values will result in larger switching losses at low loads. 3) Generate voltage gain curves for various values of Q. On that plot, vlso graph the boundary curve separating LHS and RHS regions, similar to FIG. 25.
• step 5 Using the Q and normalized frequency values found in step 4, calculate the I T and C r values using equations 9, 10 and 11.
• the design process can be easily automated through software and can be app] led to any general form of the LLC circuit as shown in FIG. 26(a) with the interrupt switch and FIG. 26(b) without the interrupt switch.
• FIG. 28 shows the voltage gain curves of the designed converter, as well as the region of operation. Note how the region of operation remains in the "BUS Operation" region.
• the converter design described in the previous section is unique for the gi ven constraints and the selected Ln/Lr ratio. However, each time the designer selects new constraints, a new set of components must be calculated. As a consequence, there are in infinite number of different LLC converters that operate with high boosting and low Q.
• Table A shows a small sample of possible resonant Link component values for converters designed to operate at 300kHz and various Q and voltage boosting values. All of these converters may be successfully operated using frequency control to regulate load power.
• DC-DC converters of the present invem ion may provide an efficient, low cost alternative to numerous components providing high input-to-output voltage conversion.
• DC-DC converters with high amplification ratios that are embodiments of the present invention may be use* I to create a fixed voltage DC bus in renewable/alternative energy applications.
• the (A) method of operating a resonant DC - DC converter of the present invention may be used in connection with a range of different applications, including in connection with photovoltaic systems; a fuel cells; permanent magnet wind turbines; electric and hybrid vehicles; electric charge stations; aerospace systems; marine systems; power grids or smart grids including micro grids; and energy storage systems.

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• 2012
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