WO2018114528A1 - Circuit de commande muni d'un régulateur à deux positions pour la régulation d'un convertisseur cadencé - Google Patents

Circuit de commande muni d'un régulateur à deux positions pour la régulation d'un convertisseur cadencé Download PDF

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Publication number
WO2018114528A1
WO2018114528A1 PCT/EP2017/082639 EP2017082639W WO2018114528A1 WO 2018114528 A1 WO2018114528 A1 WO 2018114528A1 EP 2017082639 W EP2017082639 W EP 2017082639W WO 2018114528 A1 WO2018114528 A1 WO 2018114528A1
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WO
WIPO (PCT)
Prior art keywords
converter
clocked
clocked converter
current
lower threshold
Prior art date
Legal status (The legal status is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the status listed.)
Ceased
Application number
PCT/EP2017/082639
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German (de)
English (en)
Inventor
Thomas Pollischansky
Filippo Branchetti
Markus Heckmann
Yanshun Xue
Current Assignee (The listed assignees may be inaccurate. Google has not performed a legal analysis and makes no representation or warranty as to the accuracy of the list.)
Osram GmbH
Original Assignee
Osram GmbH
Priority date (The priority date is an assumption and is not a legal conclusion. Google has not performed a legal analysis and makes no representation as to the accuracy of the date listed.)
Filing date
Publication date
Application filed by Osram GmbH filed Critical Osram GmbH
Priority to CN201780079928.5A priority Critical patent/CN110168890B/zh
Publication of WO2018114528A1 publication Critical patent/WO2018114528A1/fr
Anticipated expiration legal-status Critical
Ceased legal-status Critical Current

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Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/02—Conversion of DC power input into DC power output without intermediate conversion into AC
    • H02M3/04—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters
    • H02M3/10—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/1563—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators without using an external clock
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/02—Conversion of DC power input into DC power output without intermediate conversion into AC
    • H02M3/04—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters
    • H02M3/10—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M3/00—Conversion of DC power input into DC power output
    • H02M3/02—Conversion of DC power input into DC power output without intermediate conversion into AC
    • H02M3/04—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters
    • H02M3/10—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M3/145—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M3/155—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M3/156—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators
    • H02M3/158—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load
    • H02M3/1588—Conversion of DC power input into DC power output without intermediate conversion into AC by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only with automatic control of output voltage or current, e.g. switching regulators including plural semiconductor devices as final control devices for a single load comprising at least one synchronous rectifier element
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0016—Control circuits providing compensation of output voltage deviations using feedforward of disturbance parameters
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0003—Details of control, feedback or regulation circuits
    • H02M1/0025—Arrangements for modifying reference values, feedback values or error values in the control loop of a converter
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes

Definitions

  • the invention relates to a control circuit for controlling a clocked synchronous rectifying rectifier such. a synchronously rectifying buck converter by means of two-point controller and superimposed threshold value control
  • the invention is based on a control circuit with a two-level controller for controlling a clocked converter according to the preamble of the main claim.
  • Fig. 1 shows a known buck converter with the main components also known.
  • a switch SO is connected in series with a freewheeling diode DF.
  • the connection point of the cathode of the freewheeling diode DF and the switch TO is connected to a throttle L.
  • the other terminal of the reactor L is connected to a filter capacitor C_filter.
  • the other end of the filter capacitor C filter and the anode of the diode DF are connected to ground.
  • the other terminal of the switch SO is together with the ground of the input of the buck converter.
  • the output of the buck converter is parallel to the filter capacitor C filter.
  • Such buck converters are widely used and work satisfactorily. However, at low output voltages, Zero Voltage Switching operation is no longer possible. As a result, the switch SO is very hot and must be sized accordingly larger.
  • Fig. 2 shows some relevant signals of the known buck converter.
  • the current IL is the current through the inductance L. It is good to see that the converter operates in operation at the gap boundary, also referred to as "transition mode.”
  • the switch When the switch is switched on, the current rises sharply due to the magnetization of the choke After that, the transformer inductor is demagnetized again, which takes much longer than the magnetization at low voltage, the current flows through the freewheeling diode DF, it is good to see that the transistor is switched on again.
  • the converter will operate at the gap limit during operation, and at input voltages above 200V, this mode of operation is a favorable compromise between good efficiency, good power density and cost, but with lower output voltages loss-less switching more possible, as based on the time course of UM in Fig .2 is good to see.
  • the natural Umschwingvor- the voltage UM at the half-bridge center reaches only a fraction of the input voltage.
  • the achievable value is twice the output voltage, or slightly more when taking into account the real reverse delay charge of the diode.
  • the remaining voltage swing must be achieved by lossy hard switching on the MOS-FETs. This can be seen from the first flat increase in the voltage UM at the half-bridge center.
  • the voltage UG shows in the equal to the gate-source voltage of the transistor SO. At the moment when the UM reaches the maximum of its natural Umschwingvorganges, the transistor SO is turned on.
  • a control circuit having a two-level controller for controlling a clocked converter having an upper threshold which characterizes the switch-off time of a first converter transistor of the clocked converter, a lower threshold which the output switching point of a second converter transistor of the clocked converter, wherein the lower threshold is adjusted depending on an output voltage or depending on an output current of the clocked converter that certain operating parameters of the clocked converter are met, and wherein the upper threshold is set so that the output current of the clocked converter corresponds to a predetermined output current of the clocked converter, the thresholds resulting from operating parameters of the clocked converter and from unavoidable delay times of real components, wherein the lower threshold is determined by means of a current through a converter choke of the clocked converter and the current is negative by the converter choke at the switching time.
  • Certain operating parameters of the clocked converter may be, for example, a favorable switching behavior, in particular a voltage-free switching of the converter transistor (so-called Zero Voltage Switching, ZVS). This is achieved at low currents only after a certain time after a zero crossing of the current through the converter choke.
  • ZVS Zero Voltage Switching
  • Unavoidable delay times of real components are e.g. the delay times of operational amplifiers, comparators or logic gates such as flip-flops.
  • negative current is meant the current through the converter choke, which again negatively alsmagne- tarra after a swing after demagnetization.
  • the lower threshold is dependent on the output voltage.
  • the lower threshold is determined based on a ratio of an input voltage of the clocked converter to an output voltage of the clocked converter. This can advantageously reduce losses at high ratios of input voltage to output voltage.
  • the lower threshold is set lower at low output voltage than at higher output voltage. This measure ensures an advantageous voltage-free switching of the converter transistors even at very low output voltages.
  • the lower threshold is set lower for a lower output current than for a higher output current. This measure ensures a voltage-free switching of the converter transistors even at very low output currents.
  • the described measures can also be mixed in a particularly preferred embodiment, so that a voltage-free switching of the converter transistors is ensured at all output voltages and output currents.
  • the lower threshold is determined based on the output power and / or the input voltage of the clocked converter. This measure allows advantageous voltage-free switching of the converter transistors in an even larger operating range of the clocked converter.
  • the upper threshold is determined based on the setpoint value of the output current of the clocked converter and on the basis of the lower threshold. This advantageously ensures operation with very accurate control of the predetermined output current of the clocked converter. Further advantageous developments and refinements of the inventive control circuit result from further dependent claims and from the following description.
  • Fig. 1 is a schematic diagram of a known buck converter according to the prior art
  • Fig. 2 is a timing diagram of the known buck converter
  • Fig. 3 is a schematic diagram of a known synchronously rectifying buck converter
  • FIG. 5 is a block diagram of an embodiment of the two-point controller
  • Fig. 6 is a timing diagram of the two-point controller
  • FIG. 7 shows a first analogue embodiment of a synchronously rectifying step-down converter with an embodiment of the two-point controller.
  • FIG. 8 shows a second digital embodiment of the synchronously rectifying step-down converter with an embodiment of the two-point controller Preferred embodiment of the invention
  • FIG. 3 shows a schematic circuit diagram of a known sync rectifying buck converter.
  • the essential difference from the topology explained in FIG. 1 is the replacement of the converter diode DF by a lower transistor SU.
  • the positive input is at a DC potential of about 400V, the negative input is a reference potential.
  • the converter inductor L is connected to the half-bridge center HSS, the other connection of the converter inductor L together with the reference potential forms the output LED + / LED- of the converter.
  • a filter capacitor C_filter is connected.
  • Fig. 4 shows a timing diagram of the known synchronously rectifying buck converter
  • the voltage UGO is the voltage at the gate of the upper transistor SO
  • the voltage UGU the voltage at the gate of the lower transistor SU.
  • the converter does not operate in operation at the gap limit, but in non-leaking operation in such a way that the transistor is turned off only at a negative inductor current, iniller Embodiment at about -0.5A.
  • the choke L is magnetized up when the converter transistor SO is switched on (signal UGO is high) and is again magnetized after switching off the converter transistor SO.
  • a positive inductor current IL always flows. After a long demagnetization time, the current becomes zero and then negative. This is because the lower transistor remains switched on and thus a current path is still present.
  • the current through the converter inductor IL thus becomes negative in this time range until the lower transistor SU is switched off. This has As a result, the transistor can be switched even with very low loads with low switching losses, as can be seen in FIG.
  • a dead time is provided between the switching off of the upper transistor SO and the switching on of the lower transistor SU, during which the transient of the half-bridge takes place.
  • the voltage across the respective switch is practically zero at the moment of switching off or on (Zero Voltage Switching - ZVS).
  • this dead time is also provided between turning off the lower transistor and turning on the upper transistor.
  • Fig. 5 now shows a block diagram of an embodiment of the two-position controller which can operate the above synchronously rectifying buck converter with low loss and optimum power in the manner described above.
  • the current ILED of a clocked converter is measured by a current measuring unit 514 and fed via a first filter 515 to a comparison unit 517.
  • a voltage signal URef corresponding to the desired output current is input via a second filter 516.
  • the result is fed to a control amplifier 51 1, which determines therefrom the upper threshold, ie the switch-off instant of a first converter switch of a clocked converter 512, and feeds this thereto.
  • the lower threshold, ie the switch-off time of a second converter switch is determined by the module 513, which uses the power P and / or the desired output current corresponding voltage signal URef and / or the output voltage UA of the clocked converter.
  • the output current ILED of the clocked converter 512 is in turn measured by the current measuring unit 514, whereby a control loop is established.
  • this control ensures an exact setting of the desired output current ILED, but also takes into account the characteristic of the clocked converter via the module 513.
  • the time of switching on the first switch of the clocked converter after maintaining a dead time to avoid short circuits in the transistor bridge determined.
  • the aim of the optimization is to enable a more favorable switching behavior of the converter transistor of the clocked converter over a wide output voltage range and, in addition, to be able to set the output current over a wide range.
  • the lower threshold may be lower than at higher output currents. This allows the frequency to be reduced for smaller currents.
  • a higher lower threshold is chosen to prevent losses in the components due to additional reactive currents. Due to the negative current in the converter choke of the clocked converter namely reactive currents occur in the clocked converter, which must be considered. These are thermally uncritical for small currents, in contrast to the switching losses and the drive losses, which would result from too high switching frequencies.
  • the lower switching threshold may also be lower than at higher output voltages.
  • the lower threshold is set at a lower value, thereby providing more energy in the inductor for the freewheeling phase to enable the converter transistor to de-energize. Other parameters for setting the lower threshold may be, for example, the power and the input voltage.
  • the upper threshold is respectively adjusted by the variable gain amplifier 51 1 to compensate for both the lower threshold change and to maintain the output current of the clocked converter at the setpoint according to the voltage URef.
  • FIG. 6 shows a timing diagram of the two-position controller which drives a clocked converter such as the synchronously rectifying step-down converter discussed in FIG.
  • the diagram is therefore similar to that in FIG. 4.
  • the signal 530 shows the current through the converter inductor L with the lower threshold 522 and the upper threshold 522.
  • ren threshold 521 which indicate the (after the dead time active) switch-on and the turn-off of the upper converter transistor UGO.
  • the drive signal of the upper converter transistor UGO and the drive signal of the lower rectifying transistor UGU is applied. It is important here in comparison to a known converter control that the current through the converter choke can also assume negative values in order to be able to always control the transistors in the clocked converter with Zero Voltage Switching (ZVS). This is referred to as Forced Continuous Conduction Mode (FCCM) as already mentioned. What is new here is the combination of a two-position controller with an additional controller, which sets the upper threshold of the two-position controller so that the desired output current and FCCM operation over very large ranges of the output voltage and the output current can be achieved.
  • ZVS Zero Voltage Switching
  • control principle is not limited to a synchronously rectifying buck converter, embodiments with a flyback converter are likewise conceivable.
  • FIG. 7 now shows a schematic circuit diagram of a first embodiment of the synchronously rectifying buck converter.
  • the converter is operated with the above-described two-position controller, wherein the switch-off of the lower transistor SU is set at about -0.5A inductor current, and the turn-off of the upper transistor for the purpose of current control of connected LEDs is variable.
  • the switch-off time of the upper switch determines the maximum current through the switch and the converter choke. This must be determined so that the average current through the choke corresponds to the specified current through the LEDs.
  • the filter capacitor at the output theoretically falsifies the correlation between the current IL through the converter inductor and the output current ILED, but this steady-state error is zero because the capacitor does not provide a DC path.
  • the current ILED through the LEDs 5 is detected with two measuring resistors RS1 and RS2, where RS1 is optional.
  • the voltage across both measuring resistors RS1 and RS2 is fed to a differential amplifier 13 with the transfer function H (s), the difference between the setpoint US and the through RS1 & RS2 supplied actual value amplified
  • the output of the differential amplifier 13 specifies the threshold value for the maximum current through the converter inductor L.
  • the transfer function H (s) must be such that the control loop is stable in terms of control technology.
  • the output signal of the differential amplifier 13 with transfer function is fed to the negative input of a first comparator 14. The positive input is the falling across the resistor RS2
  • the output of the first comparator 14 is supplied to a reset input R of a flip-flop 16.
  • the voltage drop across the resistor RS2 voltage is also fed to a negative input of a second comparator 15.
  • the positive input of the second comparator 15 is connected to a reference voltage, which is a measure of the turn-off threshold of the lower transistor SU. With this voltage, the turn-off of the lower transistor SU at a certain negative inductor current can be adjusted as described above.
  • the half-bridge driver circuit 17 ensures that a certain dead time is maintained between the switching operations of the upper and lower transistors, so that no short-circuit current through the half-bridge can occur and also the complete swinging of the half-bridge is done before the respective transistor is turned on again.
  • the logic in the half-bridge driver is as follows:
  • the lower transistor SU If the output Q of the flip-flop 16 jumps high, the lower transistor SU is turned off as quickly as possible. Then follows the dead time during which both transistors are off. After the dead time, the upper transistor SO is turned on. If the output signal Q of the flip-flop jumps back to low, the upper transistor SO is switched off as quickly as possible. Then follow again the dead time during which both transistors are off. After the dead time, the lower transistor SU is turned on.
  • the function of the overall circuit is as follows: By increasing the control deviation by means of differential amplifier 13 with the transfer function H (s), the threshold value for the comparator 14 is generated.
  • the comparator 14 compares the current value with the threshold. This results in a turn-off threshold of the upper transistor, which corresponds to the desired current value through the LEDs. If the current current value exceeds the predetermined setpoint, the output of the first comparator 14 goes high and resets the flip-flop 16.
  • the upper transistor is now switched off.
  • the current now flows from the converter inductor L through the LEDs 5 via the parasitic output capacitance of the half-bridge back to the converter inductor L and the half-bridge voltage UM oscillates to zero. Then the current commutates to the freewheeling diode of the lower transistor SU. Shortly thereafter, the dead time has elapsed and the lower transistor SU is turned on.
  • the current value is input to the negative input of the second comparator 15.
  • the minimum current value Imin is entered as the voltage at which the lower transistor is to switch off again.
  • the output of the second comparator 15 switches to high and sets the flip-flop again. This turns off the lower transistor.
  • the current now flows from the inductor into the parasitic output capacitance of the half-bridge and the voltage UM oscillates up to the value of the input voltage UE. Then the current commutates to the freewheeling diode of the upper transistor SO. Shortly thereafter, the dead time has expired and the upper transistor SO is turned on. As soon as the current through the converter choke L the
  • capacitors in the form of capacitors can also be arranged. These are typically connected to one or both MOS FETs, each between drain and Source, connected. Frequently, these capacitors are also connected in series with a resistor. These snubber circuits can further reduce the switching losses in the MOS FETs.
  • Fig. 8 shows a second embodiment of the synchronously rectifying buck converter.
  • the second embodiment of the converter is a digital embodiment with a microcontroller.
  • the second embodiment is structurally similar to the first embodiment, so that only the differences from the first embodiment will be described below.
  • the flip-flop 16 is replaced by a microcontroller 3 that has implemented more advanced control mechanisms.
  • the turn-on and turn-off thresholds are reported to the microcontroller through the first and second comparators 14 and 15, as in the analog version, but the microcontroller does not respond as a flip-flop but implements a digital controlled system and allows e.g. by additional targeted delay times a flexible adjustment of the operating parameters of the clocked converter.
  • the switch-off time of the lower switch is in one embodiment dependent on the voltage of the LED chain 5 and is chosen later by the microcontroller, the smaller the voltage of the LED chain 5 is to allow for low-loss switching as possible.
  • the threshold of Comparator 15 can be changed depending on the output voltage.
  • threshold and delay times can be changed depending on any parameters.
  • the microcontroller then controls the half-bridge driver 17 accordingly, in order to achieve a low-loss operation of the converter with maximum accuracy of the output current.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Dc-Dc Converters (AREA)

Abstract

L'invention concerne un circuit de commande muni d'un régulateur à deux positions pour la régulation d'un convertisseur cadencé, présentant un seuil supérieur qui caractérise l'instant de mise hors circuit d'un premier transistor du convertisseur cadencé, un seuil inférieur qui caractérise l'instant de mise hors circuit d'un second transistor du convertisseur cadencé, le seuil inférieur étant ajusté en fonction d'une tension de sortie ou en fonction d'un courant de sortie du convertisseur cadencé de telle manière que des paramètres de fonctionnement déterminés du convertisseur cadencé soient remplis, et le seuil supérieur étant ajusté de telle manière que le courant de sortie du convertisseur cadencé corresponde à un courant de sortie prédéfini du convertisseur cadencé. Les seuils sont obtenus à partir des paramètres de fonctionnement du convertisseur cadencé et à partir des temps de retard inévitables de composants réels, le seuil inférieur étant déterminé au moyen d'un courant traversant un étranglement du convertisseur cadencé et le courant traversant l'étranglement du convertisseur à l'instant de mise en circuit étant négatif. L'invention concerne également un procédé de régulation d'un convertisseur cadencé, comprenant les étapes consistant à mettre hors circuit un premier transistor du convertisseur cadencé à un seuil supérieur, à mettre hors circuit un second transistor du convertisseur cadencé à un seuil inférieur, à ajuster le seuil inférieur en fonction d'une tension de sortie ou en fonction d'un courant de sortie du convertisseur cadencé de telle manière que des paramètres de fonctionnement déterminés du convertisseur cadencé soient remplis, et à ajuster le seuil supérieur de telle manière que le courant de sortie du convertisseur cadencé corresponde à un courant de sortie prédéfini du convertisseur cadencé, le seuil inférieur et le seuil supérieur étant obtenus à partir des paramètres de fonctionnement du convertisseur cadencé et à partir des temps de retard inévitables de composants réels, le seuil inférieur étant déterminé au moyen d'un courant traversant un étranglement du convertisseur cadencé et le courant traversant l'étranglement du convertisseur à l'instant de mise en circuit étant négatif.
PCT/EP2017/082639 2016-12-22 2017-12-13 Circuit de commande muni d'un régulateur à deux positions pour la régulation d'un convertisseur cadencé Ceased WO2018114528A1 (fr)

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Application Number Priority Date Filing Date Title
CN201780079928.5A CN110168890B (zh) 2016-12-22 2017-12-13 调节时钟驱动的变换器的具有双点调节器的控制电路

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DE102016226001.4A DE102016226001A1 (de) 2016-12-22 2016-12-22 Steuerschaltung mit einem zweipunktregler zur regelung eines getakteten wandlers
DE102016226001.4 2016-12-22

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WO2018114528A1 true WO2018114528A1 (fr) 2018-06-28

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CN (1) CN110168890B (fr)
DE (1) DE102016226001A1 (fr)
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