WO2020157053A1 - Procédé de réglage d'un système redresseur d'impulsions triphasé avec circuit intermédiaire - Google Patents

Procédé de réglage d'un système redresseur d'impulsions triphasé avec circuit intermédiaire Download PDF

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Publication number
WO2020157053A1
WO2020157053A1 PCT/EP2020/052022 EP2020052022W WO2020157053A1 WO 2020157053 A1 WO2020157053 A1 WO 2020157053A1 EP 2020052022 W EP2020052022 W EP 2020052022W WO 2020157053 A1 WO2020157053 A1 WO 2020157053A1
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Prior art keywords
phase
current
bridge
voltage
switch
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Ceased
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PCT/EP2020/052022
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German (de)
English (en)
Inventor
Johann Walter Kolar
Mattia Guacci
Dominik Bortis
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Eidgenoessische Technische Hochschule Zurich ETHZ
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Eidgenoessische Technische Hochschule Zurich ETHZ
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Classifications

    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M7/00—Conversion of AC power input into DC power output; Conversion of DC power input into AC power output
    • H02M7/02—Conversion of AC power input into DC power output without possibility of reversal
    • H02M7/04—Conversion of AC power input into DC power output without possibility of reversal by static converters
    • H02M7/12—Conversion of AC power input into DC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode
    • H02M7/21—Conversion of AC power input into DC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal
    • H02M7/217—Conversion of AC power input into DC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only
    • H02M7/219—Conversion of AC power input into DC power output without possibility of reversal by static converters using discharge tubes with control electrode or semiconductor devices with control electrode using devices of a triode or transistor type requiring continuous application of a control signal using semiconductor devices only in a bridge configuration
    • G—PHYSICS
    • G05—CONTROLLING; REGULATING
    • G05F—SYSTEMS FOR REGULATING ELECTRIC OR MAGNETIC VARIABLES
    • G05F1/00—Automatic systems in which deviations of an electric quantity from one or more predetermined values are detected at the output of the system and fed back to a device within the system to restore the detected quantity to its predetermined value or values, i.e. retroactive systems
    • G05F1/10—Regulating voltage or current 
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
    • H02M1/4208—Arrangements for improving power factor of AC input
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
    • H02M1/4208—Arrangements for improving power factor of AC input
    • H02M1/4225—Arrangements for improving power factor of AC input using a non-isolated boost converter
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/42—Circuits or arrangements for compensating for or adjusting power factor in converters or inverters
    • H02M1/4208—Arrangements for improving power factor of AC input
    • H02M1/4233—Arrangements for improving power factor of AC input using a bridge converter comprising active switches
    • H—ELECTRICITY
    • H02—GENERATION; CONVERSION OR DISTRIBUTION OF ELECTRIC POWER
    • H02M—APPARATUS FOR CONVERSION BETWEEN AC AND AC, BETWEEN AC AND DC, OR BETWEEN DC AND DC, AND FOR USE WITH MAINS OR SIMILAR POWER SUPPLY SYSTEMS; CONVERSION OF DC OR AC INPUT POWER INTO SURGE OUTPUT POWER; CONTROL OR REGULATION THEREOF
    • H02M1/00—Details of apparatus for conversion
    • H02M1/0048—Circuits or arrangements for reducing losses
    • H02M1/0054—Transistor switching losses
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02B—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO BUILDINGS, e.g. HOUSING, HOUSE APPLIANCES OR RELATED END-USER APPLICATIONS
    • Y02B70/00—Technologies for an efficient end-user side electric power management and consumption
    • Y02B70/10—Technologies improving the efficiency by using switched-mode power supplies [SMPS], i.e. efficient power electronics conversion e.g. power factor correction or reduction of losses in power supplies or efficient standby modes
    • Y—GENERAL TAGGING OF NEW TECHNOLOGICAL DEVELOPMENTS; GENERAL TAGGING OF CROSS-SECTIONAL TECHNOLOGIES SPANNING OVER SEVERAL SECTIONS OF THE IPC; TECHNICAL SUBJECTS COVERED BY FORMER USPC CROSS-REFERENCE ART COLLECTIONS [XRACs] AND DIGESTS
    • Y02—TECHNOLOGIES OR APPLICATIONS FOR MITIGATION OR ADAPTATION AGAINST CLIMATE CHANGE
    • Y02T—CLIMATE CHANGE MITIGATION TECHNOLOGIES RELATED TO TRANSPORTATION
    • Y02T10/00—Road transport of goods or passengers
    • Y02T10/60—Other road transportation technologies with climate change mitigation effect
    • Y02T10/7072—Electromobility specific charging systems or methods for batteries, ultracapacitors, supercapacitors or double-layer capacitors

Definitions

  • the invention relates to the field of three-phase pulse rectifier systems, in particular to a method for regulating the input phase currents to a sinusoidal shape and the DC output voltage.
  • the input terminals, i.e. the center taps, of a bidirectionally lockable three-phase bridge circuit are connected directly to the mains terminals, the individual switching elements of the three bridge branches being connected, for example, by the antiseries circuit of a unidirectionally blocking component and a diode, the antiseries circuit of two unidirectionally blocking components, or a bidirectional component can be realized.
  • a switched DC voltage is now generated at the output of the three-phase bridge circuit, that is to say between the positive and negative output voltage rail, the mean value of which is exactly the same as that DC voltage to be formed corresponds, but whose instantaneous value corresponds in sections to one of the possible chained input voltages depending on the selected pulse pattern.
  • a low-pass filter consisting of an output inductance and a DC support capacitance is therefore arranged on the output terminals of the three-phase bridge circuit.
  • Clocking is understood to mean high-frequency switching of the converters or their circuit breakers, in contrast to the low-frequency profile of the line voltages.
  • a corresponding clock frequency is also called switching frequency, its reciprocal is one clock period.
  • an input current curve proportional to the mains voltage is advantageously set by appropriate selection of the pulse pattern.
  • the three-phase pulse rectifier system behaves like a three-phase resistance load and thus draws a constant power from the three-phase network, i.e.
  • an average constant DC inductance current or intermediate circuit current flows in the output inductance.
  • the timing of the bridge branches can also be understood as switching the constant DC inductance current to the three input phases, the pulse pattern being selected such that the distribution of the DC inductance current at the center taps of the three-phase bridge circuit leads to three input voltage proportional input currents.
  • the DC inductance current By switching the DC inductance current, however, pulse-shaped phase currents are obtained from the center taps of the three-phase bridge circuit, which is why input capacitors are arranged in a star or delta circuit to smooth the phase currents at the input of the three-phase pulse rectifier system.
  • an input current curve with a defined phase shift with respect to the mains voltage can also be set in a certain range.
  • a DC / DC step-up converter stage must be arranged after the system, which results in a two-stage three-phase step-up converter pulse rectifier system, the output inductance of the three-phase rectifier step-down converter stage also advantageously being used as the inductance of the step-up converter will, ie the bridge branch of the step-up converter stage is arranged between the output inductance of the three-phase rectifier step-down converter stage and the DC supporting capacitance, and the inductance thus forms a DC intermediate circuit inductor as a DC intermediate circuit inductor (cf. FIG. 1).
  • step-up converter stage is designed bidirectionally, then a power supplied by an active load can also be fed into the DC intermediate circuit between the two converter stages and from there power can be fed back into the network. This operation is e.g. when feeding photovoltaically generated power into the three-phase network or when feeding a three-phase machine from a DC voltage.
  • a high clock frequency of the converter stages is advantageously to be provided in both cases, but this results in relatively high switching losses, which reduce the efficiency of the energy conversion. Furthermore, a relatively high drive power is required for the high-frequency clocking of the electronic switches, which also affects the efficiency.
  • the object of the invention is therefore to provide a method for the control and modulation of a two-stage three-phase step-up converter pulse rectifier system which has reduced switching losses, with a sinusoidal line current curve advantageously lying in phase or in phase opposition with the line voltage and a constant one Output voltage should be ensured.
  • the method is used to regulate a three-phase pulse rectifier system, which has a three-phase bridge circuit, for power exchange between a three-phase network and an intermediate circuit inductance, and a load converter for exchanging power between the intermediate circuit inductance and a load.
  • the bridge branches of the three-phase bridge circuit for switching phase currents flowing through the intermediate circuit inductance are arranged, each in one section in chronologically consecutive sections, each lasting one sixth of the period of the three-phase network
  • the three-phase bridge circuit to which the line phase with the largest phase current in terms of magnitude hereinafter referred to as selected line phase, is assigned, this line phase optionally depending on the polarity of the phase current with a switch of the selected bridge branch, hereinafter referred to as a selected switch, is connected to a positive connection point or a negative connection point of the three-phase bridge circuit, this line phase being connected to the positive connection point in the case of positive phase current and to the negative connection point in the case of negative phase current, and remaining connected for the duration of this section,
  • the unselected switch remains open for the entire duration of the section, i.e. it is not conductive. There is therefore no free running of the intermediate circuit current through a bridge branch. Neither the selected switch nor the opposite switch in the same phase are clocked. Switching losses and a drive power requirement that would occur with a clocking are avoided.
  • Opposite switches are understood to be the upper switches with respect to a lower switch, and vice versa. For example, if the lower switch of the first bridge branch is the selected switch, the opposite switches in the non-selected bridge branches are equal to the upper switches in the second and third bridge branches.
  • a sinusoidal phase current is set in each case in a time segment by the clocking in each of the other two, not selected, bridge branches.
  • the load converter sets a voltage at a load converter-side connection of the intermediate circuit inductance by clocking, which, in cooperation with a switched direct voltage formed by the three-phase bridge circuit, results in a predetermined intermediate circuit current at the other, bridge circuit-side connection of the intermediate circuit inductance.
  • the AC side or network side is considered as the input and the DC voltage side as the output.
  • a power flow can not only take place from the entrance to the exit but also vice versa.
  • the circuit breakers used are generally also bidirectionally conductive and switching, ie they can conduct currents in both directions through the switch and switch them on or off. If a power flow is to be realized in only one direction, unidirectionally conductive and switching circuit breakers can be used.
  • the subject matter of the invention is based on preferred
  • Figure 1 Power section of the system, the input section as a bidirectionally lockable
  • Three-phase bridge circuit and a load converter is designed as a DC / DC step-up converter.
  • Figure 2 Current profiles of the network input currents (4, and 4) and the
  • Intermediate circuit current 4 The division into the six current sectors and the allocation of the network currents to the intermediate circuit current (4) takes place depending on the current conditions of the network input currents, the intermediate circuit current (4) always corresponding to the largest network input current (4, 4 or 4).
  • FIG. 3 Voltage profiles of the mains input voltages (u a , U b and u c ) and the switched DC voltage Ubu in the third sector at time (ti) for the conventional control method.
  • the course of the switched DC voltage U bu has four voltage intervals, starting with a freewheeling interval (cc), ie a total of four switchovers are necessary during a switching period.
  • Figure 4 Exemplary voltage profiles of the mains input voltages (u a , b and u c ) and the switched DC voltage bu in the third sector at time (ti) for control methods according to the invention.
  • the course of the switched DC voltage bu has only two voltage intervals, ie the number of switchovers can be reduced to two during a switching period.
  • Figure 5 (composed of Figures 5a and 5b) circuitry implementation of the
  • the load converter is a bridge branch and positive output voltage terminal arranged in the connecting line between the positive output voltage rail of the three-phase bridge Full-bridge switch cell, which actively forms the DC link current via appropriate clocking.
  • Figure 7 Alternative version of the load converter as a pulse converter, in particular as
  • Pulse inverter for direct feeding of a machine M.
  • the current intermediate circuit profile (k) is thus regulated in the current intermediate circuit in such a way that a six-pulse current is established over a network period, the instantaneous value of which corresponds in each case to the largest phase current in terms of amount (see FIG. 2).
  • the bridge branch belonging to the largest phase current can be switched through within an input current sector depending on the polarity of the largest phase current, i.e. for a positive phase current, the upper switch remains and for a negative phase current, the lower switch remains on within an input current sector, and this bridge branch can be switched to freewheeling, i. simultaneous activation of the upper and lower switches can be prevented.
  • the degree of freedom with regard to the choice of the intermediate circuit current curve is thus used, the largest phase current being formed by changing the current in the intermediate circuit instead of pulse width modulation of all phases.
  • the method according to the invention thus avoids clocking the bridge branch with the largest phase current in terms of amount, which advantageously ensures lower switching losses and a lower drive power requirement (cf. FIG. 2).
  • the sum of the phase currents is forced to zero or the largest phase current flows back as a sum over the other two phases. Since the bridge branch with the largest phase current is not clocked in the method according to the invention, the largest phase current is divided between the other two Phases exclusively by switching between the two remaining bridge branches instead of switching between all bridge branches, whereby the lead time of the two switching intervals can be divided in such a way that a sinusoidal phase current curve is also achieved in the other two phases (see Figure 2).
  • the switched DC voltage at the output of the three-phase bridge circuit only consists of two instead of three concatenated input voltages, i.e. two instead of three voltage levels, which means that the number of switchovers in one clock period can be halved from four to two compared to the conventional method.
  • the greatest voltage jump can advantageously be avoided both times, as a result of which lower switching losses and a lower switching frequency ripple in the intermediate circuit inductance are achieved (cf. FIGS. 3 and 4).
  • the switched DC voltage (u bu ) is then composed of four voltage intervals, assuming a symmetrical pulse pattern, the upper and lower switches of the phase switch starting with a free-running interval (u cc ) (c) associated bridge branch are switched on and then only the upper switch of phase (b) or (o) is switched on for the two active linked voltages (u bc ) and (u ac ).
  • the intermediate circuit current (/ L ) is regulated according to the method according to the invention to a six-pulse course corresponding to the largest phase current in terms of magnitude, then, for example, for the same point in time (ti) shown in FIG (u ⁇ x) are omitted, the switched DC voltage (ubu) then assuming a symmetrical pulse pattern is composed of only two voltage intervals, ie the two active linked voltages (u bc ) and (u ac ), only the upper switch of each Phase (b) or (a) is switched on and the lower switch of phase (c) remains switched on continuously.
  • the lower switch of the bridge branch belonging to phase (c) also remains switched on permanently, but the upper switches each change between all three phases during a switching period, while in the method according to the invention the upper switches only change between two phases and thus the intermediate circuit current (/ L ) flows back through phase (c) during the entire interval shown in FIG. 4.
  • the pulse pattern and the switched voltage for the other sectors are carried out by cyclically interchanging the phases.
  • the load converter can be used to set the voltage (u bo ) at the load converter-side connection of the intermediate circuit inductance such that, in cooperation with the voltage at the other connection on the bridge circuit side (towards the bridge circuit, i.e. the switched DC voltage U bu ), the desired intermediate circuit current ( 4) results.
  • the switched DC voltage U bu is predetermined by the mains voltages and the switching states of the three-phase bridge circuit (2).
  • the desired intermediate circuit current (4) should - since it is sectionally equal to one of the phase currents - follow the six-pulse course mentioned.
  • the two-stage three-phase step-up converter pulse rectifier system is not limited to a pure step-up converter function, but can also be operated as a step-down converter.
  • the upper switch of the step-up converter on the output side must be switched through so that the output voltage (u Q ) corresponds to the mean value of the switched DC voltage at the output of the three-phase rectifier (U bu ); the active three-phase bridge is then operated as a conventional step-down converter Regulation of a constant intermediate circuit current or output voltage more than two bridge branches are to be clocked.
  • a method according to the prior art can then be used for the control.
  • a switch is made between the regulation according to the invention and a regulation according to the prior art.
  • the three-phase buck-boost converter pulse rectifier system to be controlled and regulated by the method according to the invention can have a known structure on the input side by means of a bidirectionally lockable three-phase two-point bridge circuit (2) (bridge circuit) with three AC phase inputs (a, b and c) and a positive one (p) and a negative voltage rail (s) are formed, both voltage rails being led to the input of an output-side converter stage (load converter) (3) which feeds a consumer, or a voltage (load voltage) (u D ) across the consumer. generated (see Figure 1).
  • phase inputs of the bridge circuit (o, b and c) are connected either directly or via an input filter to the associated phase terminals (a, b and c) of the three-phase AC mains (mains) (1), but at least each phase input the bridge circuit (o, b and c) is led to a filter capacitor (line filter capacitors) (C), which are connected either in a star or in a triangle, in order to keep the pulsating input currents in the three-phase bridge circuit that occur during operation away from the mains or the supply voltage independently to define the internal impedance of the network.
  • a filter capacitor line filter capacitors
  • the bridge circuit (2) generally has three bridge branches, each bridge branch having an upper electronic switch with bipolar blocking capability connected to the positive voltage rail (p) and a lower one connected to the negative voltage rail (s), and the phase output of the bridge branch (a, b and c) is formed by the junction of the free ends of the upper and lower switches.
  • the bipolar lockable switches (21 and 22) can each be implemented, for example, by the antiseries circuit of a unidirectionally blocking component and a diode, the antiseries circuit of two unidirectionally blocking components, or by a bidirectionally blocking component.
  • the load converter (3) can be designed as a simple bidirectional step-up converter, which has a switched two-point bridge branch (step-up converter bridge branch) (31), the upper switch (311) of which has the positive DC output voltage rail (d) and the lower switch (312) is connected to the negative DC output voltage rail (s), a buffer capacitor (output capacitor) (C Q ) being arranged between these two DC output voltage rails in order to support the output voltage (u 0 ) for the load present above it.
  • step-up converter bridge branch 31
  • the upper switch (311) of which has the positive DC output voltage rail (d) and the lower switch (312) is connected to the negative DC output voltage rail (s)
  • a buffer capacitor (output capacitor) (C Q ) being arranged between these two DC output voltage rails in order to support the output voltage (u 0 ) for the load present above it.
  • the switching point (ß) of the two-point bridge arm (31), i.e. the connection point of the upper switch (311) and lower switch (312) is connected to a first connection of a step-up converter inductance (L), the second connection of which is connected on the input side to the positive voltage rail (p) of the input-side three-phase bridge circuit (2) and thus as an intermediate circuit inductance acts.
  • the negative output voltage terminal of the two-point bridge branch (31) is connected directly to the negative voltage rail (s).
  • the system is modulated depending on the size relationships of the line phase currents (in other words: depending on the relations of the absolute values of the line phase currents), for a symmetrical three-phase network in sections within 1/6 of the network period, i.e. within a 60 ° wide sector or section a complete oscillation period of the mains voltages corresponding to 360 °, in each case the bridge branch belonging to the largest phase current is switched through depending on the polarity of the largest phase current, the upper switch (21) for a positive phase current and the lower switch (22) for a negative phase current Input power sector remains on. This results in a course of six pulses over a network period for the intermediate circuit current (/ L ), the instantaneous value of which corresponds in each case to the largest phase current (see FIG. 2).
  • FIG. 2 shows, for example, in a first time segment, corresponding to a sector 1: the largest current in terms of amount, it flows in the second phase (b) and has a negative sign.
  • the lower switch S b, L of the second (or selected) phase is switched through and carries the intermediate circuit current (/ L ) for the duration of this section.
  • ES remain the upper switch S b, H of this selected phase and the two other lower switches s a , L ; S C , L open.
  • the other two upper switches s a , H, S C , H can can be switched in order to divide the returning intermediate circuit current (/ L ) into the corresponding (not selected) two phases.
  • the largest phase current or the intermediate circuit current (k) must flow back as a sum over the other two phases. If the greatest phase current flows via the lower switch (22), the upper switches (21) of the bridge branches belonging to the other two phases must be switched on alternately, the lower switch (22) of which remains permanently switched off.
  • the switching period is divided between the two switching intervals in such a way that a sinusoidal phase occurs in all phases
  • Phase current curve is reached. If the phase with the largest phase current is passed through an upper switch (21), the lower switches (22) of the other two phases must be switched on alternately, the upper switch (22) then remaining switched off permanently (see FIG. 2 ).
  • a clock frequency is typically at least ten times or twenty times or fifty times a basic frequency of the alternating voltages, in particular the mains voltages.
  • the modulation is also adjusted accordingly. This means that the phase with the largest phase current is assigned to a different physical phase in each sector (cf. FIG. 2).
  • the aim of the system control is to determine sinusoidal currents (4, and 4) in phase (for power consumption from the network or in phase opposition for power recovery) with the associated network phase voltage (u 3 , U b and u c ) ) to achieve, with all currents having the same amplitude for a symmetrical network and optionally generate a defined load voltage (u) corresponding to a predetermined setpoint (load voltage setpoint) (u 0 * ) at the output of the load converter (3), or generally deliver a defined output to the consumer (see FIG. 5).
  • the line phase currents can also have a phase shift with respect to the associated line phase voltages.
  • ohmic network behavior is assumed for the sake of clarity.
  • the system can be seen as a star connection of the same ohmic resistances (input equivalent resistances) or conductance values (input equivalent conductance values), the power of which is passed on directly to the output, ie to the consumer, with the ideal assumption of losslessness.
  • the sum of the squares of the input phase voltages (u a , U b and u c ) determines the input setpoint (G * ), which corresponds to a power consumption from the network equal to the output power setpoint (P a * ).
  • the duty cycle for the three-phase bridge circuit (c4, c4 and c4) can be calculated from the ratio of the respective input phase current setpoints (4 * , 4 * and 4 * ) and the intermediate circuit current setpoint (4 * ), from which the effective duty cycle (C4, H, C4, L, C, H, C4, L, C / C , H and d c , L) and the local mean value of the switched DC voltage at the output of the three-phase bridge circuit (U bu ).
  • the local mean value of the switched DC voltage (Ubu) is obtained either by multiplying the calculated duty cycles (c4, d b and c4) by the corresponding phase voltages (u a , U b and u c ) or alternatively (not shown) by dividing the output power setpoint ( P 0 * ) by the DC link current setpoint (4 * ).
  • the intermediate circuit current or inductance current (4) is then regulated with a subordinate current control by first determining the control deviation by subtracting the measured intermediate circuit current (4) from the intermediate circuit current setpoint (4 * ) and then supplying it to an intermediate circuit current controller (52), which is supplied at its output outputs the target value of the voltage (UL) to be formed on the associated DC link inductance (L) on average over a clock period. After subtracting this setpoint (U L * ) from the calculated local mean value of the switched DC voltage (u bu ), the local mean value of the voltage setpoint (u bo * ) results which is at the center of the Booster bridge branch is required.
  • the direct duty cycle (cf d ) or the effective switching signal (s d ) for the upper one results directly from a modulator (53) Switch (311) of the step-up converter bridging branch, the lower switch (312) of the bridging branch being switched off during the on period of the upper switch (311), ie both switches of the bridging branch operate in push-pull and the clock period preferably a constant length or the clock frequency preferably a constant Has value.
  • the intermediate circuit current actual value (4) is managed in accordance with the associated intermediate circuit current setpoint curve (4 * ). It should be noted that the clocking of the three-phase bridge circuit and the
  • the step-up converter bridge branch can be made, for example, with the same clock frequency and can be timed so that a minimal switching-frequency fluctuation of the intermediate circuit current (4) is achieved.
  • a full-bridge switch cell 6 with internal DC voltage, for implementing active shaping of the intermediate circuit current (cf. FIG. 6).
  • the intermediate circuit current is equal to the output current, which is clocked accordingly
  • Full bridge switch cell 6 is impressed. Since the cell only compensates for power pulsations with six times the mains frequency, no DC supply is required. In addition, switching elements with a relatively low reverse voltage can advantageously be used for the full-bridge switching cell 6 compared to the three-phase bridge circuit.
  • a three-phase pulse converter in particular pulse inverter 7 (see FIG. 7), or another voltage-stabilizing converter stage such as also galvanically isolated DC / DC converters.
  • the full bridge switch cell 6 has two connections. It can be formed by two bridge branches, which are parallel to one another and parallel to a switch cell capacitance are connected between a switch cell positive pole and a switch cell negative pole. The center taps of the bridge branches form the two connections.
  • the circuit breakers of the full-bridge switching cell 6 can be designed to switch unidirectionally, in each case with antiparallel freewheeling diodes. It should be noted that for the drawings and the above statements, ohmic network behavior is assumed to be desirable, but the control method according to the invention can also be used for a phase shift of the network phase currents with respect to the assigned network phase voltages, in which case the phase current setpoints (which according to the above statements for ohmic network behavior corresponding blind components are added.

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  • Engineering & Computer Science (AREA)
  • Power Engineering (AREA)
  • Physics & Mathematics (AREA)
  • Electromagnetism (AREA)
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  • Rectifiers (AREA)

Abstract

Un procédé est utilisé pour régler un système redresseur d'impulsions triphasé (1, 2, 3) comprenant un circuit en pont triphasé (2), une inductance de circuit intermédiaire (L) et un convertisseur de charge (3). Dans des sections temporellement successives, respectivement une section, - dans la branche de pont, ci-après dénommée branche de pont sélectionnée, du circuit en pont triphasé (2), à laquelle est associée la phase de réseau ayant le plus fort courant de phase en valeur absolue, ci-après dénommée phase de réseau sélectionnée, cette phase de réseau est reliée au choix en fonction de la polarité du courant de phase à un commutateur de la branche de pont sélectionnée, ci-après dénommé commutateur sélectionné, à un point de raccordement positif (p) ou un point de raccordement négatif (n) du circuit en pont triphasé (2), cette phase de réseau étant reliée au point de raccordement positif dans le cas d'un courant de phase positif et au point de raccordement négatif dans le cas d'un courant de phase négatif, et restant connectée pendant la durée de cette portion, - dans les deux autres branches de pont non sélectionnées, le point de raccordement positif ou négatif correspondant est relié alternativement aux deux phases non sélectionnées par cadencement des commutateurs opposés au commutateur sélectionné.
PCT/EP2020/052022 2019-02-01 2020-01-28 Procédé de réglage d'un système redresseur d'impulsions triphasé avec circuit intermédiaire Ceased WO2020157053A1 (fr)

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CH00127/19 2019-02-01
CH00127/19A CH715813B1 (de) 2019-02-01 2019-02-01 Verfahren zum Regeln eines Dreiphasen-Pulsgleichrichtersystems mit Stromzwischenkreis.

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Cited By (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN115001294A (zh) * 2022-06-07 2022-09-02 中国人民解放军海军工程大学 一种循环脉冲大功率消磁主电源系统
CN118842002A (zh) * 2024-09-24 2024-10-25 湖南大学 基于svg相间有功功率交换的新能源配电网故障电压支撑方法
CN120301216A (zh) * 2025-06-16 2025-07-11 湖南科技大学 基于Buck-Boost逆变电路的低纹波可调直流稳压电源主电路参数稳定域确定方法

Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP1760870A2 (fr) * 2005-09-06 2007-03-07 Siemens Aktiengesellschaft Redresseur haute tension
EP3217522A1 (fr) * 2016-03-08 2017-09-13 Siemens Aktiengesellschaft Systeme redresseur ayant une capacite de retour
WO2017153366A1 (fr) * 2016-03-09 2017-09-14 Renault S.A.S Procédé et dispositif de commande en monophasé d'un chargeur de véhicules à traction électrique ou hybride embarqué sans isolation galvanique

Patent Citations (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
EP1760870A2 (fr) * 2005-09-06 2007-03-07 Siemens Aktiengesellschaft Redresseur haute tension
EP3217522A1 (fr) * 2016-03-08 2017-09-13 Siemens Aktiengesellschaft Systeme redresseur ayant une capacite de retour
WO2017153366A1 (fr) * 2016-03-09 2017-09-14 Renault S.A.S Procédé et dispositif de commande en monophasé d'un chargeur de véhicules à traction électrique ou hybride embarqué sans isolation galvanique

Cited By (3)

* Cited by examiner, † Cited by third party
Publication number Priority date Publication date Assignee Title
CN115001294A (zh) * 2022-06-07 2022-09-02 中国人民解放军海军工程大学 一种循环脉冲大功率消磁主电源系统
CN118842002A (zh) * 2024-09-24 2024-10-25 湖南大学 基于svg相间有功功率交换的新能源配电网故障电压支撑方法
CN120301216A (zh) * 2025-06-16 2025-07-11 湖南科技大学 基于Buck-Boost逆变电路的低纹波可调直流稳压电源主电路参数稳定域确定方法

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CH715813B1 (de) 2022-05-31

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