Classifications

• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
  • H03H9/00—Networks comprising electromechanical or electro-acoustic elements; Electromechanical resonators
  • H03H9/46—Filters
  • H03H9/462—Microelectro-mechanical filters
  • H03H9/465—Microelectro-mechanical filters in combination with other electronic elements
• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
  • H03H9/00—Networks comprising electromechanical or electro-acoustic elements; Electromechanical resonators
  • H03H9/02—Details
  • H03H9/02244—Details of microelectro-mechanical resonators
  • H03H9/02393—Post-fabrication trimming of parameters, e.g. resonance frequency, Q factor
  • H03H9/02409—Post-fabrication trimming of parameters, e.g. resonance frequency, Q factor by application of a DC-bias voltage
• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
  • H03H9/00—Networks comprising electromechanical or electro-acoustic elements; Electromechanical resonators
  • H03H9/46—Filters
  • H03H9/462—Microelectro-mechanical filters
• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
  • H03H9/00—Networks comprising electromechanical or electro-acoustic elements; Electromechanical resonators
  • H03H9/46—Filters
  • H03H9/462—Microelectro-mechanical filters
  • H03H9/467—Post-fabrication trimming of parameters, e.g. center frequency
• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
  • H03H9/00—Networks comprising electromechanical or electro-acoustic elements; Electromechanical resonators
  • H03H9/46—Filters
  • H03H9/48—Coupling means therefor
  • H03H9/50—Mechanical coupling means
  • H03H9/505—Mechanical coupling means for microelectro-mechanical filters
• • H—ELECTRICITY
  • H01—ELECTRIC ELEMENTS
  • H01H—ELECTRIC SWITCHES; RELAYS; SELECTORS; EMERGENCY PROTECTIVE DEVICES
  • H01H59/00—Electrostatic relays; Electro-adhesion relays
  • H01H59/0009—Electrostatic relays; Electro-adhesion relays making use of micromechanics
• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
  • H03H9/00—Networks comprising electromechanical or electro-acoustic elements; Electromechanical resonators
  • H03H9/02—Details
  • H03H9/02244—Details of microelectro-mechanical resonators
  • H03H2009/02488—Vibration modes
  • H03H2009/02511—Vertical, i.e. perpendicular to the substrate plane
• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
  • H03H9/00—Networks comprising electromechanical or electro-acoustic elements; Electromechanical resonators
  • H03H9/02—Details
  • H03H9/02244—Details of microelectro-mechanical resonators
  • H03H2009/02488—Vibration modes
  • H03H2009/02519—Torsional
• • H—ELECTRICITY
  • H03—ELECTRONIC CIRCUITRY
  • H03H—IMPEDANCE NETWORKS, e.g. RESONANT CIRCUITS; RESONATORS
  • H03H9/00—Networks comprising electromechanical or electro-acoustic elements; Electromechanical resonators
  • H03H9/02—Details
  • H03H9/02244—Details of microelectro-mechanical resonators
  • H03H2009/02488—Vibration modes
  • H03H2009/02527—Combined

Definitions

• This invention relates to methods and apparatus for generating a signal having at least one desired output frequency utilizing a bank of vibrating micromechanical devices.
• Figure 2 illustrates a comparison of MEMS and SAW technologies wherein MEMS offers the same or better high- ⁇ 2 frequency selectivity with orders of magnitude smaller size. Indeed, reductions in size and board-level packaging complexity, as well as the desire for the high performance attainable by superheterodyne architectures, are principal drivers for this technology. ⁇ Although size reduction is certainly an advantage of this technology
• the stability of the oscillator signals used for frequency translation, synchronization, or sampling is of utmost importance.
• Oscillator frequencies must be stable against variations in temperature against aging, and against any phenomena, such as noise or microphonics, that cause instantaneous fluctuations in phase and frequency.
• the single most important parameter that dictates oscillator stability is the Q of the frequency-setting tank (or of the effective tank for the case of ring oscillators). For a given application, and assuming a finite power budget, adequate long- and short-term stability of the oscillation frequency is insured only when the tank Q exceeds a certain threshold value.
• Tank Q also greatly influences the ability to implement extremely selective IF and RF filters with small percent bandwidth, small shape factor, and low insertion loss. As tank Q decreases, insertion loss increases very quickly, too much even for IF filters, and quite unacceptable for RF filters. As with oscillators, high- ⁇ 2 tanks are required for RF and IF filters alike, although more so for the latter, since channel selection is done predominantly at the IF in super-heterodyne receivers. In general, the more selective the filter, the higher the resonator Q required to achieve a given level of insertion loss.
• U.S. Patent No. 6,049,702 to Tham et al. discloses an integrated passive transceiver section wherein microelectromechanical (MEM) device fabrication techniques are used to provide low loss, high performance switches. Utilizing the MEM devices also makes possible the fabrication and use of several circuits comprising passive components, thereby enhancing the performance characteristics of the transceiver.
• MEM microelectromechanical
• U.S. Patent No. 5,872,489 to Chang et al. discloses an integrated tunable inductance network and method.
• the network utilizes a plurality of MEM switches which selectively interconnect inductance devices thereby providing a selective inductance for a particular circuit.
• U.S. Patent No. 5,963,857 to Greywall discloses an article comprising a micromachined filter.
• the micromachined filters are assembled as part of a radio to miniaturize the size of the radio.
• U.S. Patent Nos. 5,976,994 and 6,169,321 to Nguyen et al. disclose a batch-compatible, post-fabrication annealing method and system to trim the resonance frequency and enhance the quality factor of micromechanical structures.
• U.S. Patent Nos. 5,455,547; 5,589,082 and 5,537,083 to Lin et al. disclose microelectromechanical signal processors.
• the signal processors include many individual microelectromechanical resonators which enable the processor to function as a multi-channel signal processor or a spectrum analyzer.
• U.S. Patent No. 5,640,133 to MacDonald et al. discloses a capacitance-based, tunable, micromechanical resonator.
• the resonators may be selectively tuned and used in mechanical oscillators, accelero meters, electromechanical filters and other electronic devices.
• U.S. Patent Nos. 5,578,976 to Yao, 5,619,061 to Goldsmith et al. and 6,016,092 to Qiu et al. disclose various micromechanical and microelectromechanical switches used in communication apparatus.
• U.S. Patent No. 5,839,062 to Nguyen et al. disclose a MEMS-based receiver including parallel banks of microelectromechanical filters.
• U.S. Patent No. 5,783,973 to Weinberg et al. discloses a micromechanical, thermally insensitive silicon resonator and oscillator.
• the local oscillator (LO) synthesizer is among the most difficult functional blocks to implement in a transceiver, and among the largest power consumers in the total subsystem.
• two oscillators are utilized: a high-frequency (0.4-4 GHz) voltage-controlled oscillator (VCO) referenced to a medium Q (Q ⁇ 40) off-chip LC tank; and a lower frequency ( ⁇ 10 MHz) reference oscillator using a quartz crystal with a loaded Q ⁇ 10,000.
• the VCO In the synthesizer, the VCO generates the LO output, and is able to achieve good far- from-carrier phase noise, but must be locked via a phase-locked loop to the reference oscillator to attain adequate thermal stability and sufficiently low close-to- carrier phase noise.
• the power consumption in such synthesizers is often dominated by the requirement for low phase noise ( ⁇ 30 mW needed in the VCO for short- range applications), and by the need for phase-locking and frequency division when locking the two oscillators ( ⁇ 70 mW needed). If oven-control is required for stability against thermal variations, an additional 1-10 W may be required, not to mention a much larger size.
• An object of the present invention is to provide a method and apparatus for generating a signal having at least one desired output frequency in a system in response to a tuning voltage and without the need for a phase-locking circuit by utilizing a bank of vibrating micromechanical devices.
• a method for generating a signal having at least one desired output frequency in a system without the need for a phase-locking circuit.
• the method includes providing a bank of micromechanical devices and vibrating one or more of the devices so that each vibrating device generates a signal having an output frequency.
• the method also includes controllably selecting one or more of the vibrating devices to generate a signal having the at least one desired output frequency in response to the tuning voltage.
• Each of the devices may have a Q in excess of 100 and, preferably, more than 5000.
• the number of devices may be more than 10, and may even be more than 1000.
• the desired output frequency may be more than 300 MHz.
• Each of the devices may have an adjustable Q.
• the method may further include lowering the Q of the at least one vibrating device having the at least one desired output frequency initially during the step of vibrating and then raising the Q during the step of vibrating.
• the step of lowering may include controUably coupling at least one circuit element to the at least one vibrating device having the at least one desired frequency and the step of raising may include the step of controUably decoupling the at least one circuit element from the at least one vibrating device having the at least one desired frequency.
• a local oscillator synthesizer apparatus for generating a signal having at least one desired output frequency is provided for use in a system in response to a tuning voltage and without the need for a phase-locking circuit.
• the apparatus includes a parallel bank of micromechanical resonators. Each of the resonators corresponds to a frequency.
• a controller is provided for selecting at least one desired resonator of the bank of resonators to thereby provide frequency selection.
• the apparatus further includes an oscillator sustaining circuit wherein each of the resonators is switchable into or out of the oscillator sustaining circuit by the controller.
• the controller may include a decoder for controlling application of an appropriate bias voltage and to the at least one desired resonator.
• the resonators may be interlinked and wherein each of the resonators may have an input connected to a common input and also may have an output connected to a common output.
• the frequency of each of the resonator devices is not only switchable but also tunable (i.e. , via dc-biasing) to obtain a bank of switchable, tunable resonator devices.
• Each of the resonators has a predetermined Q and the controller may selectively switch at least one circuit element to the at least one desired resonator to lower the Q of the at least one desired resonator from its predetermined Q and selectively switches the at least one circuit element from the at least one desired resonator to raise the Q of the at least one desired resonator to its predetermined Q.
• FIGURE 1 is a prior art, schematic view of the front-end of a transceiver including off-chip, board-level implementation of SAW, ceramic and crystal resonators in a schematic perspective view;
• FIGURE 2a is a prior art, schematic perspective view of a SAW resonator and a number of MEMS resonators formed on a silicon die to compare the two approaches;
• FIGURE 2b is a prior art, enlarged schematic perspective view of one of the MEMS resonators as indicated at 2b in Figure 2a;
• FIGURE 3 is a system level schematic block diagram of the front-end design for a typical wireless transceiver showing off-chip, high-Q, passive components targeted for replacement via micromechanical versions of the present invention
• FIGURE 4 is a graph of transmission [dB] versus frequency illustrating desired filter characteristics
• FIGURE 5a is a perspective schematic view of a symmetrical two- resonator VHF / ⁇ mechanical filter with typical bias, excitation and signal conditioning electronics;
• FIGURE 5b is an electrical equivalent circuit for the filter of Figure 5a;
• FIGURE 6 is a system level block diagram of an RF front-end receiver including an RF channel-select receiver architecture utilizing large numbers of micromechanical resonators in banks and schematically and perspectively illustrating a typical micromechanical filter of Figure 5a;
• FIGURE 7 is a system/circuit diagram for an RF channel-select micromechanical filter bank
• FIGURE 8 is a system level block diagram of the RF front-end receiver of Figure 6 and schematically and perspectively illustrating a micromechanical switch thereof;
• FIGURE 9 is a system level block diagram of the RF front-end receiver of Figures 6 and 8 and schematically and perspectively illustrating a mixer- filter-gain stage thereof based on the filter of Figures 6 and 5a;
• FIGURE 10 is a system level block diagram of the RF front-end receiver of Figures 6, 8 and 9 and schematically and perspectively illustrating a micromechanical resonator oscillator thereof;
• FIGURE 11 is a system/circuit diagram for a switchable ⁇ mechanical resonator synthesizer
• FIGURE 12 is a system level block diagram of the RF front-end receiver of Figures 6, 8, 9 and 10 wherein the LNA is shown eliminated by phantom lines due to the low loss channel selector, the T/R switch and the mixer- filter-gain stage;
• FIGURE 13 is a system block diagram architecture showing the receive path of a communication device
• FIGURE 14 is a system block diagram of an RF channel-select transmitter architecture utilizing high-power /imechanical resonators
• FIGURE 15 is a schematic top perspective view of a 92 MHz (VHF) free-free beam polysilicon / ⁇ mechanical resonator wherein support beams isolate the resonator beam element from a substrate thereby allowing higher Q operation;
• VHF 92 MHz
• FIGURE 16 is a graph illustrating a measured frequency characteristic for the resonator of Figure 15;
• FIGURE 17a is a schematic perspective view of a UHF ⁇ mechanical filter utilizing free-free beam /xmechanical resonators designed to operate at a second mode
• FIGURE 17b is a partial equivalent circuit for the filter of Figure 17a, identifying the circuit functions of individual beam elements.
• Figure 3 presents a system level schematic block diagram for a typical super-heterodyne wireless transceiver.
• a small box is positioned in the corner of each box to represent a component that can be replaced with a micromechanical (MEMS) version.
• MEMS micromechanical
• several of the constituent components can already be miniaturized using integrated circuit transistor technologies. These include the low noise amplifiers (LNA's) in the receive path, the solid-state power amplifier (SSPA) in the transmit path, synthesizer phase-locked loop (PLL) electronics, mixers, and lower frequency digital circuits for baseband signal demodulation.
• LNA's low noise amplifiers
• SSPA solid-state power amplifier
• PLL synthesizer phase-locked loop
• the SSPA (and sometimes the LNA's) are often implemented using compound semiconductor technologies (i.e. , GaAs). Thus, they often occupy their own chips, separate from the other mentioned transistor-based components, which are normally realized using silicon-based bipolar and CMOS technologies. However, given the rate of improvement of silicon technologies (silicon-germanium included), all of the above functions may be integrated onto a single-chip.
• clamped-clamped and free-free flexural-mode beams with ⁇ 2's on the order of 10,000 (in vacuum) and temperature coefficients on the order of -12 ppm/°C, are available for the VHF range, while thin-film bulk acoustic resonators ( ⁇ 2 ⁇ 1,000) have so far addressed the UHF range.
• Anchor loss mechanisms can be greatly alleviated by using "anchorless" resonator designs, such as shown in the above-noted patent application Serial No. 09/482,670 and as illustrated by the view of Figure 15.
• This device utilizes a free-free beam (i.e. , xylophone) 60 suspended above a ground plane and sense electrode 61 and a drive electrode 63 by four torsional support beams 62 attached at flexural node points. The beams 62, in turn, are supported at anchors 64.
• UHF frequency is obtained via use of free- free beam resonators specifically designed to operate at higher modes.
• the desired mode is selected (while suppressing other modes), by strategic placement and excitation of electrodes, and by the use of dimples under the structure 60 to force node locations corresponding to the desired mode.
• the mode can also be specified by placing the support beams at nodal locations, as in Figure 17a.
• Figure 16 shows a frequency characteristic for a 92.25 MHz version of this ⁇ mechanical resonator with a Q of nearly 8,000 — still plenty for channel- select RF applications.
• Table 1 presents expected resonance frequencies for various beam dimensions, modes, and structural materials, showing a wide range of attainable frequencies, from VHF to UHF.
• Timoshenko methods Determined for free-free beams using Timoshenko methods that include the effects of finite h and W r .
• ⁇ mechanical circuits for communications are those implementing low-loss bandpass filters, capable of achieving frequency characteristics as shown in Figure 4 where a broader frequency passband than achievable by a single resonator beam is shown, with a sharper roll-off to the stopband (i.e. , smaller shape factor).
• a number of micromechanical resonators may be coupled together by soft coupling springs.
• a coupled resonator system is achieved that exhibits several modes of vibration.
• the frequency of each vibration mode corresponds to a distinct peak in the force-to-displacement frequency characteristic, and to a distinct, physical mode shape of the coupled mechanical resonator system.
• each resonator in the lowest frequency mode, all resonators vibrate in phase; in the middle frequency mode, the center resonator ideally remains motionless, while the end resonators vibrate 180° out of phase; and finally, in the highest frequency mode, each resonator is phase- shifted 180° from its adjacent neighbor.
• the complete mechanical filter exhibits a jagged passband.
• termination resistors designed to lower the ⁇ 2's of the input and output resonators by specific amounts are required to flatten the passband and achieve a more recognizable filter characteristic, such as in Figure 4.
• the filters use a number of high- ⁇ micromechanical beam elements connected in a network that achieves the specified bandpass frequency response. If effect, a micromechanical filter is another example of a micromechanical circuit, similar to that of Figure 15, but in this case using a plurality of beam elements to achieve a frequency shaping response not achievable by a single beam element.
• the constituent resonators in ⁇ mechanical filters are normally designed to be identical, with identical dimensions and resonance frequencies.
• the center frequency of the overall filter is equal to the resonance frequency f 0 of the resonators, while the filter passband (i.e. , the bandwidth) is determined by the spacings between the mode peaks.
• the relative placement of the vibration peaks in the frequency characteristic — and thus, the passband of the eventual filter — is determined primarily by the stiffnesses of the coupling springs (k si j) and of the constituent resonators at the coupling locations (k r ). Specifically, for a filter with center frequency f 0 and bandwidth B, these stiffnesses must satisfy the expression: where k y is a normalized coupling coefficient found in filter cookbooks.
• the filter bandwidth is not dependent on the absolute values of resonator and coupling beam stiffness; rather, their ratio k siJ /k r dictates bandwidth.
• the procedure for designing a mechanical filter involves two main steps (not necessarily in this order): first, design of a mechanical resonator with resonance frequency f 0 and adjustable stiffness k r ; and second, design of coupling springs with appropriate values of stiffness k ⁇ j to enable a desired bandwidth within the adjustment range of resonator k r 's.
• Figure 5 a shows a perspective view schematic of a practical two- resonator micromechanical filter capable of operation in the HF to VHF range.
• the filter consists of two ⁇ mechanical clamped-clamped beam resonators with anchors 18 at their opposite ends, coupled mechanically by a soft coupling spring or beam 19, all suspended above a substrate (not shown).
• Conductive (polysilicon) strips 20, 22, 24, and 26 underlie each resonator by approximately 1000 A (as also in Figures 6, 9 and 17a), a center one 20 serving as a capacitive transducer input electrode positioned to induce resonator vibration in a direction perpendicular to the substrate, a center one 24 serving as an output electrode and the flanking ones 22 and 26 serving as tuning or frequency pulling electrodes capable of voltage-controlled tuning of resonator frequencies.
• the resonator-to-electrode gaps are determined by the thickness of a sacrificial oxide spacer during fabrication and can thus be made quite small (e.g. , 0.1 ⁇ m or less) to maximize electromechanical coupling.
• the filter is excited with a DC-bias voltage V P applied to the conductive mechanical network, and an AC signal applied to the input electrode, but this time through an appropriately valued source resistance R ⁇ that loads the Q of the input resonator to flatten the passband.
• the output resonator of the filter must also see a matched impedance to avoid passband distortion, and the output voltage v Q is generally taken across this impedance.
• the required value of I/O port termination resistance can be tailored for different applications, and this can be advantageous when designing low noise transistor circuits succeeding the filter, since such circuits can then be driven by optimum values of source resistance to minimize noise.
• An electrical input signal is applied to the input port and converted to an input force by the electromechanical transducer (which for the case of Figure 5a is capacitive) that can then induce mechanical vibration in the x direction;
• Mechanical vibration comprises a mechanical signal that is processed in the mechanical domain — specifically, the signal is rejected if outside the passband of the filter, and passed if within the passband;
• micromechanical signal processor clearly suits this device. Details of the design procedure for micromechanical filters now follow.
• the network topologies for the mechanical filters of this work differ very little from those of their purely electronic counte ⁇ arts, and in principal, can be designed at the system-level via a procedure derived from well-known, coupled resonator ladder filter synthesis techniques.
• a procedure derived from well-known, coupled resonator ladder filter synthesis techniques Given the equivalent LCR element values for a prototype ⁇ mechanical resonator, it is possible to synthesize a mechanical filter entirely in the electrical domain, converting to the mechanical domain only as the last step.
• such a procedure is not recommended, since knowledge and ease of design in both electrical and mechanical domains can greatly reduce the effort required.
• the coupling beam is recognized as an acoustic transmission line that can be made transparent to the filter when designed with quarter-wavelength dimensions as described in the prior art.
• the filter bandwidth B is determined not by absolute values of stiffness, but rather by a ratio of stiffnesses (k sl2 /k rc ), where the subscript c denotes the value at the coupling location; and second, the value of resonator stiffness k rc varies with location (in particular, with location velocity) and so can be set to a desired value by simply choosing an appropriate coupling beam attachment point.
• the location is easily determined as described in the prior art.
• Figure 5b presents the equivalent circuit for the two-resonator micromechanical filter of Figure 5a.
• Each of the outside resonators are modeled via circuits.
• the coupling beam actually operates as an acoustic transmission line, and thus, is modeled by a -network of energy storage elements.
• Transformers are used between the resonator and coupling beam circuits of Figure 5b to model the velocity transformations that arise when attaching the coupling beams at locations offset from the center of the resonator beam.
• the whole circuit structure of Figure 5b can be recognized as that of the LC ladder network for a bandpass filter.
• Figure 17a is similar to Figure 5a and presents the schematic of a filter structure, along with a partial equivalent electrical circuit of Figure 17b (obtained via electromechanical analogy) that identifies the mechanical network as a bandpass filter.
• the filter structure is seen to be comprised of a number of mechanical resonators (modeled by LCR tanks) connected by acoustic transmission lines (modeled by T-networks of energy storage elements) — a structure similar to other resonator-based filters, but using micromechanical elements with orders of magnitude higher Q, giving it the ability to perform with much lower insertion loss than other technologies. (Not to mention orders of magnitude smaller size.)
• UHF frequency is obtained via use of a second mode free-free beam resonator including beams 70 and 72 coupled together and to an output beam 74 by coupling beams 73.
• the resonator is specifically designed to operate at higher modes.
• the desired mode is selected (while suppressing other modes), by strategic placement and excitation of balanced input electrodes 76 and 78, and by the use of dimples 80 under the beams 70, 72 and 74 to force node locations corresponding to the desired mode.
• Balanced output electrodes 81 are positioned under the output beam 74.
• the beams 70 and 74 are supported by non-intrusive supports or beams 82 and 84, respectively, which, in turn, are supported by anchors 86 and 88, respectively.
• the filter of Figure 17a constitutes the first attempt of its kind to implement filter circuits using free-free beam micromechanical resonators. Unlike previous filters using clamped-clamped beams (that could only attain VHF frequency), both transversal and torsional motions are considered for the coupling beams in Figure 17a.
• Figure 9 presents the schematic for a symmetrical ⁇ mechanical mixer-filter, showing the bias and input scheme required for down-conversion with a gain stage.
• this device provides filtering as part of its function, the overall mechanical structure is exactly that of a ⁇ mechanical filter. The only differences are the applied inputs and the use of a non-conductive coupling beam to isolate the IF port from the LO. If the source providing V P to the second resonator is ideal (with zero source resistance) and the series resistance in the second resonator is small, LO signals feeding across the coupling beam capacitance are shunted to AC ground before reaching the IF port.. In reality, finite resistivity in the resonator material allows some amount of LO-to-IF leakage.
• the mixer conversion gain/loss in this device is determined primarily by the relative magnitudes of the DC-bias V P applied to the resonator and the local oscillator amplitude V LO .
• the mixer-filter device described above is one example of a ⁇ mechanical circuit that harnesses non-linear device properties to provide a useful function.
• Another very useful mode of operation that further utilizes the non-linear nature of the device is a ⁇ mechanical switch.
• Figure 8 presents an operational schematic for a ⁇ mechanical switch.
• a conductive or actuation plate 30 is suspended above a pair of actuation electrodes 32 by suspension beams 34 having anchors 36.
• a switch conductor portion 38 of the plate 30 is suspended over a pair of grounds 40 and a sense electrode or conductor 42. When the switch is in the "on-state" here, the conductor 42 is shorted to the grounds 40.
• switches In general, to minimize insertion loss, the majority of switches use metals as their structural materials. It is their metal construction that makes ⁇ mechanical switches so attractive, allowing them to achieve "on- state" insertion losses down to 0.1 dB — much lower than FET transistor counte ⁇ arts, which normally exhibit ⁇ 2 dB of insertion loss. In addition to exhibiting such low insertion loss, ⁇ mechanical switches are extremely linear, with IIP3's greater than 66 dBm, and can be designed to consume no DC power (as opposed to FET switches, which sink a finite current when activated).
• Figure 3 shows the use of other M ⁇ MS-based passive components, such as medium- ⁇ 2 micromachined inductors and tunable capacitors used in VCO's and matching networks, as well as low-loss ( ⁇ 0.1 dB) ⁇ mechanical switches that not only provide enhanced antenna diversity, but that can also yield power savings by making TDD (rather than FDD) more practical in future transceivers.
• ⁇ mechanical circuits offer the same system complexity advantages over off-chip discrete components that planar IC circuits offer over discrete transistor circuits.
• ⁇ mechanical circuits should be utilized on a massive scale, or at least as much as possible.
• Figure 6 presents the system-level block diagram for a possible receiver front-end architecture that takes full advantage of the complexity achievable via ⁇ mechanical circuits, such as the micromechanical filter of Figure 5a.
• the main driving force behind this architecture is power reduction, attained in several of the blocks by trading power for high selectivity (i.e. , high- ⁇ ).
• the key power saving blocks in Figure 6 are now described.
• channel selection (rather than pre-selection) were possible at RF frequencies (rather than just an IF)
• succeeding electronic blocks in the receive path e.g. , LNA, mixer
• LNA local oscillator
• Figure 7 presents one fairly simple rendition of the key system block that realizes the desired RF channel selection.
• this block consists of a bank of ⁇ mechanical filters with all filter inputs connected to a common block input and all outputs to a common block output, and where each filter passband corresponds to a single channel in the standard of interest.
• each filter passband corresponds to a single channel in the standard of interest.
• it does not have to be limited to a single channel. It could also be several channels as well (e.g. , 3 channels) and this could still be very advantageous, depending upon the communications standard.
• a given filter is switched on (with all others off) by decoder-controlled application of an appropriate DC-bias voltage to the desired filter.
• the desired force input and output current are generated in a ⁇ mechanical resonator only when a DC-bias V P is applied (i.e. , without V P , the input and output electrodes are effectively open-circuited).
• this RF channel selector can be quantified by assessing its impact on the LNA linearity specification imposed by the IS-98-A interim standard for CDMA cellular mobile stations.
• the required IIP3 of the LNA is set mainly to avoid desensitization in the presence of a single tone (generated by AMPS) spaced 900 kHz away from the CDMA signal center frequency.
• AMPS AMPS-generated single tone spaced 900 kHz away from the CDMA signal center frequency.
• reciprocal mixing of the local oscillator phase noise with the 900 kHz offset single tone and cross-modulation of the single tone with leaked transmitter power outputs dictate that the LNA IIP3 exceeds +7.6 dBm.
• the power requirement in the sustaining amplifier might be dictated solely by loop gain needs (rather than by phase noise needs), which for a ⁇ mechanical resonator-based VCO with R X ⁇ 40 ⁇ , L x ⁇ 84 ⁇ H, and C X ⁇ 0.5JF, might be less than 1 mW.
• a switchable bank similar to that of Figure 7 but using ⁇ mechanical resonators 46, not filters, each corresponding to one of the needed LO frequencies, and each switchable into or out of an oscillator sustaining circuit 48 by transistor switches 49 at their electrodes 51 is illustrated in Figure 11 and is preferred over the tuning-fork- resonator oscillator 50 illustrated in Figure 10. Because ⁇ mechanical resonators are now used in this implementation, the Q and thermal stability of the oscillator may now be sufficient to operate without the need for locking to a lower frequency crystal reference. The power savings attained upon removing the PLL and prescaler electronics needed in past synthesizers can obviously be quite substantial.
• synthesizer power consumption can be reduced from the -90 mW dissipated by present-day implementations using medium-Q L and C components, to something in the range of only 1-4 mW. Again, all this is attained using a circuit topology that would seem unreasonable if only macroscopic high- ⁇ resonators were available, but that becomes plausible in the micromechanical arena.
• Figure 11 presents the basic topology of the LO synthesizer.
• a bank of numerous switchable high- ⁇ 2 (Q> 5,000) micromechanical resonators is utilized, where each resonator corresponds to one of the needed frequencies in a given communications network.
• no phase-locking circuit is required, since the local oscillator merely switches the appropriate resonator into the oscillator feedback loop to generate the needed output frequency.
• the Q of the micromechanical resonators are orders of magnitude higher than before ( ⁇ 2 ⁇ 5,000, as opposed to 40), the phase noise performance of this oscillator should be orders of magnitude better than present-day VCO's.
• the l// 2 -to-white noise corner frequency of the standard phase noise vs. frequency plot may be so close to the carrier, that l// 2 noise may no longer be a major consideration, and only white noise is present at the important frequency offsets.
• power could be traded for Q to such an extent that it may become possible to operate the oscillator with less than 1 mW of power consumption. Combined with the fact that phase-locking is no longer required in the scheme of Figure 11 , this constitutes more than 90 mW of power savings, while improving the performance of the oscillator.
• the controller of Figure 11 preferably switches in a loss mechanism
• the controller switches out the loss so the oscillation can have high Q characteristics (e.g. , lower phase noise, low temperature dependence.)
• the resistance is typically a Q-loading resister, R Q , for fast start-up.
• R Q Q-loading resister
• an R Q should be switched in at first (i.e. , for low Q - fast start-up). Then, remove the R Q when steady state is recorded, for high Q and good phase noise.
• a single-chip LO synthesizer, using the architecture of Figure 11 can achieve impressive phase noise performance at UHF frequencies (-800 MHz). By using higher-mode, free-free beam resonator designs previously described, UHF (and possibly S-Band or higher) frequency synthesizers are feasible. K- or Ka-Band applications are not unreasonable, and should benefit from the architecture of Figure 11 either directly as shown, or indirectly, since a UHF high-Q reference oscillator should greatly improve the stability of K- or Ka-band synthesizers.
• the use of a ⁇ mechanical mixer-filter in the receive path as illustrated in Figure 9 eliminates the DC power consumption associated with the active mixer normally used in present-day receive architectures.
• the mixer-filter-gain stage includes a pair of n-type resonators 21 ,23 coupled together by a p-type or undoped beam or spring 19 similar to filters of Figures 5a and 6.
• FIG. 13 depicts a receive path comprised of a relatively wideband image reject ⁇ mechanical RF filter followed immediately by a narrowband IF mixer-filter that then feeds subsequent IF electronics.
• the only active electronics operating at RF in this system are those associated with the local oscillator, which if it uses a bank of ⁇ mechanical resonators as previously described, may be able to operate at less than 1 mW.
• Figure 14 depicts one rendition, in which an RF channel selector is placed after the power amplifier (PA) in the transmit path.
• This channel selector uses a similar circuit as that of Figure 7, but using ⁇ mechanical resonators with sufficient power handling capability.
• This transmit topology provides enormous power savings.
• the high- ⁇ , high-power filter with less than 1 dB of insertion loss follows the PA, cleaning all spurious outputs, including those arising from spectral regrowth. Consequently, more efficient PA designs can be utilized, despite their non-linearity.
• a PA previously restricted by linearity considerations to 30% efficiency in present-day transmitter architectures may now be operable closer to its maximum efficiency, perhaps 50% . For a typical transmit power of 600 mW, this efficiency increase corresponds to 800 mW of power savings.
• a more efficient PA topology could be used, such as Class E, with theoretical efficiencies approaching 100%, the power savings could be much larger.
• the architecture of Figure 14 also features a micromechanical upconverter that uses a mixer-filter device, such as previously described, to upconvert and filter the information signal before directing it to the power amplifier.
• the baseband signal at frequency ⁇ IF to be upconverted is applied to the input electrode 20 and the upconverting carrier signal at frequency ⁇ LO is applied to the input resonator device 21.
• Upconversion occurs through mixing around the nonlinear capacitive transducer between the input electrode 20 and the resonator 21.
• the electrical baseband information signal at frequency ⁇ IF is upconverted to a force at frequency ⁇ LO + ⁇ IF .
• This force is then filtered by the filter structure of Figure 9, which is now designed to have a passband around ⁇ L0 + ⁇ IF .
• this passband is made small enough, channel selection to the point of removing not only distortion harmonics, but also spectrally regrown components, is possible.
• V p dc-bias V p
• the displacement of this resonator is converted to an electrical output voltage or current, depending upon the output load. It should be noted, also, that gain is possible in upconverting the baseband signal to the RF signal, so this stage also serves as a gain stage, as well.
• the second method for upconversion involves filtering the based band signal first, then upconverting.
• the baseband signal is again applied to the input electrode 20, but the dc-bias is applied to the input resonator, and the carrier signal to the output resonator.
• the baseband signal is first filtered by the structure, then upconverted at the output via electromechanical amplitude modulation. Again, gain is possible in this configuration.
• High power handling micromechanical resonators may use alternative geometries (e.g. , no longer flexural mode) and the use of alternative transduction (e.g. , piezoelectric, magneto strictive).
• Vibrating ⁇ mechanical resonators constitute the building blocks for a new integrated mechanical circuit technology in which high Q serves as a principal design parameter that enables more complex circuits.
• high Q serves as a principal design parameter that enables more complex circuits.

Landscapes

• Physics & Mathematics (AREA)
• Acoustics & Sound (AREA)
• Piezo-Electric Or Mechanical Vibrators, Or Delay Or Filter Circuits (AREA)
• Micromachines (AREA)
• Transceivers (AREA)
• Transmitters (AREA)
• Oscillators With Electromechanical Resonators (AREA)
• Filters And Equalizers (AREA)
• Networks Using Active Elements (AREA)
• Measurement Of Velocity Or Position Using Acoustic Or Ultrasonic Waves (AREA)
• Input Circuits Of Receivers And Coupling Of Receivers And Audio Equipment (AREA)
• Amplifiers (AREA)

Applications Claiming Priority (3)

Publications (1)

Family

ID=22736053

Family Applications (4)

Family Applications After (3)

Country Status (5)

Families Citing this family (32)

Family Cites Families (4)

• 2001
  • 2001-04-20 EP EP01929080A patent/EP1285491A2/de not_active Ceased
  • 2001-04-20 AU AU2001255868A patent/AU2001255868A1/en not_active Abandoned
  • 2001-04-20 WO PCT/US2001/040565 patent/WO2001082478A2/en not_active Ceased
  • 2001-04-20 JP JP2001579448A patent/JP2003532322A/ja active Pending
  • 2001-04-20 CA CA002406543A patent/CA2406543A1/en not_active Abandoned
  • 2001-04-20 JP JP2001579439A patent/JP2003532320A/ja active Pending
  • 2001-04-20 JP JP2001579450A patent/JP2004515089A/ja active Pending
  • 2001-04-20 CA CA002406223A patent/CA2406223A1/en not_active Abandoned
  • 2001-04-20 CA CA002406518A patent/CA2406518A1/en not_active Abandoned
  • 2001-04-20 EP EP01931148A patent/EP1277277A2/de not_active Ceased
  • 2001-04-20 WO PCT/US2001/040566 patent/WO2001082479A2/en not_active Ceased
  • 2001-04-20 WO PCT/US2001/012806 patent/WO2001082475A2/en not_active Ceased
  • 2001-04-20 WO PCT/US2001/040564 patent/WO2001082477A2/en not_active Ceased
  • 2001-04-20 AU AU2001255865A patent/AU2001255865A1/en not_active Abandoned
  • 2001-04-20 AU AU2001257612A patent/AU2001257612A1/en not_active Abandoned
  • 2001-04-20 AU AU2001261036A patent/AU2001261036A1/en not_active Abandoned
  • 2001-04-20 AU AU2001255867A patent/AU2001255867A1/en not_active Abandoned
  • 2001-04-20 JP JP2001579451A patent/JP2003532323A/ja active Pending
  • 2001-04-20 EP EP01929079A patent/EP1290788A2/de not_active Ceased
  • 2001-04-20 AU AU2001255866A patent/AU2001255866A1/en not_active Abandoned
  • 2001-04-20 WO PCT/US2001/040563 patent/WO2001082467A2/en not_active Ceased
  • 2001-04-20 WO PCT/US2001/040562 patent/WO2001082476A2/en not_active Ceased
  • 2001-04-20 CA CA002406176A patent/CA2406176A1/en not_active Abandoned
  • 2001-04-20 EP EP01929082A patent/EP1275201A2/de not_active Ceased

Non-Patent Citations (1)

Also Published As

Similar Documents

Legal Events