JPH0136567B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a knock control device using a piezoelectric knock sensor, and more particularly to a shot detection circuit that detects shot when a piezoelectric knock sensor shoots.

Knocks that occur in an engine are accompanied by a knocking sound that reduces running performance, and also causes a reduction in engine output due to the generation of reverse torque or damage to the engine due to overheating. This knock is closely related to the ignition timing, and it is known that engine output can be maximized by setting the ignition timing, that is, the ignition advance, just before the knock due to the characteristics of the engine. Therefore, reducing the ignition advance angle in order to avoid the occurrence of a knock will conversely reduce the engine output, so it is necessary to control the ignition timing immediately before the occurrence of a knock. Especially in turbocharged engines,
High compression ratios require optimal ignition timing to maintain maximum efficiency.

Further, it is necessary that the knock sensor for detecting the knock condition is operating normally, and if an abnormality occurs in the knock sensor, especially a shot, it is necessary to promptly detect this.

The shot detection of the conventional piezoelectric knock sensor is
The determination is made based on whether or not an output signal is output from the piezoelectric knock sensor. That is, if the output from the piezoelectric knock sensor is zero, it is detected that a short is occurring. Therefore, for example,
As shown in No. 169232, conventionally, the output signal from a piezoelectric knock sensor is amplified and integrated, and when the output voltage of the integrator is smaller than a predetermined value, it is determined that there is a shot.

However, the shot detection means of the conventional piezoelectric knock sensor greatly amplifies the output to determine whether or not the piezoelectric knock sensor is outputting, and uses the amplified value as the basis. Since the output signal from this piezoelectric knock sensor is originally small, noise is detected in the low rotation range (usually 1500 to 2000 rotations or less), so it has the disadvantage that the threshold level cannot be set low and cannot be detected. Ta.

SUMMARY OF THE INVENTION An object of the present invention is to provide a shot detection circuit for a piezoelectric knock sensor that can detect shot of a piezoelectric knock sensor in any engine rotation range.

The present invention biases a piezoelectric knock sensor with high resistance and transmits a signal through capacitor coupling.
By detecting the shot of the piezoelectric knock sensor when the transmitted signal drops to near zero potential, the shot of the piezoelectric knock sensor is detected in any engine rotation range.

Examples of the present invention will be described below.

FIG. 1 shows the entire knock control system in accordance with one embodiment of the present invention.

In the figure, the knock control device includes a knock sensor 100 for detecting a knock signal, a knock control device 101 for outputting a control signal for controlling the ignition timing of the ignition coil 135 based on the knock signal input from the knock sensor 100, Pickup coil 105 and Pickup coil 1 for detecting spark timing of ignition coil 135
05 and the output from the knock control device 101, the ignition coil is ignited, and the knock control device 1
01 and a non-contact ignition device 103 for sending a feedback signal to the ignition device 101.

The knock control device 101 includes a knock sensor 100.
The detection signal of the non-contact ignition device 103 and the output signal of the non-contact ignition device 103 are taken in, and the non-contact ignition device 1
03 to perform advance or retard angle control.

The knock control device 101 includes a knock sensor 100.
a fail-safe device 102 that detects a failure of the engine and sends a signal to forcibly retard the ignition timing;
AC coupling circuit 112 for removing DC components;
An ignition noise cut circuit 113 having a gate for cutting ignition noise in synchronization with spark timing, a bandpass filter (BPF) 114 for bandpassing the knock signal, and BPF 11
an automatic gain control circuit (AGC circuit) 160 that controls the gain of its own amplifier in proportion to the input signal ratio by the output of 4; a mask circuit 150 that masks the AGC output at a predetermined timing;
a background level (BGL) detection circuit 119 for obtaining the average value of the knock signal from the input from the AGC circuit 160 via the mask circuit 150;
Amplifier 16 for amplifying the output of the BGL detection circuit 119 and providing feedback to the AGC circuit 160
1. Amplifier 1 that amplifies the output of the mask circuit 150
51, a comparator 118 that compares the output voltage of the BGL detection circuit 119 with the output signal of the amplifier 151 and generates a retard signal proportional to knocking; comparator 1;
a mask circuit 153 that masks the output of 18 at a predetermined timing and outputs the mask circuit 153;
an integrator 154 that integrates the output;
A retard signal setting circuit 126 for setting a retard signal proportional to knocking by the output of the ignition coil 135 by a signal from the non-contact ignition device 103.
A monostable circuit 128 that generates a signal with a constant pulse width in synchronization with the interruption of the power transistor 134 (that is, in synchronization with the base current of the power transistor 134), the monostable circuit 1
An advance signal setting circuit 12 outputs a signal for advancing the ignition timing of the ignition coil 135 at a constant voltage/period ratio while the output pulse 28 is being output.
7. Generates a DC voltage proportional to the retard signal set in the retard signal setting circuit 126 and the advance signal set in the advance signal setting circuit 127, and ignites the ignition coil 135 by the output from the failsafe device 102. It consists of an integrating circuit 125 and a reference voltage generating circuit 120, which send out an output for advancing or forcibly retarding the timing. Furthermore, this embodiment includes a circuit number detection circuit 156 and fixed time generation circuits 157 and 158 for setting the mask timing of the mask circuits 150 and 153. The fixed time generation circuit 157 also controls the integration circuit 125.

The reference voltage generation circuit 120 includes an operational amplifier OP3,
Capacitor C5, resistor R15, R11, R12,
Consists of R14. By appropriately selecting resistors R14 and R15, the output RV of the operational amplifier OP3 is fixed to the reference voltage. This reference voltage RV is, for example, 3V, and is used for various reference voltages.

The failsafe device 102 is a knock sensor 1.
It has a fail-safe function that detects various failures or abnormal modes of the 00 and performs retard control according to the mode. This device 102 consists of a sensor abnormality detection circuit 102A having a sensor short detection circuit 108, and outputs from each circuit are sent to an integrator circuit 102A.
25 and performs a fail-safe function. The BGL detection circuit 119 includes a half-wave rectifier 116,
It consists of an integrating circuit 117 and an amplifier 117A, and rectifies and integrates the output signal of the AGC circuit 160 sent through the mask circuit 150 to obtain an averaged signal of the entire signal. This averaged signal becomes a BGL signal. Comparison circuit 118 has at least two comparators 118A and 118B. The comparator 118A compares the BGL output with the output of the amplifier 151, and the comparator 118B compares the output of the amplifier 151 with a reference voltage.

The non-contact ignition device 103 includes an amplifier 131 that shapes the waveform of the output signal of the pickup coil 105;
A retard circuit 132 that controls the ignition timing according to the output voltage of the knock control circuit 101, and an ignition coil 1.
The power transistor 134 generates a high voltage on the secondary side of the power transistor 35.

Next, the operation of the above configuration will be explained based on the waveforms shown in FIG. FIG. 2 1 shows an ignition timing waveform, and this waveform signal is actually the base signal of the power transistor 134 of the non-contact ignition device 103, which will be described later. Power transistor 1 at H level
34 is on (ON), and when the power transistor 134 is at L level, the power transistor 134 is turned off (OFF). Sparks in the ignition coil 135 are generated during the process of switching from ON to OFF. 2. Signal S2 in FIG. 2 is a constant width pulse output signal of a monostable circuit 128 which receives the base signal as input and generates a constant width (t ₁ ) pulse signal by being triggered when it changes from ON to OFF.

Figure 2.3 shows the level-up knock sensor 1.
00 output is shown. The signal detected by the knock sensor 100 is a signal that swings positive and negative with the DC zero (0) level as a reference. The detection signal includes a knock signal. On the other hand, a detection circuit 108 is provided to enable fail-safety. In the failure mode of these detection circuits, the mode cannot be detected if the DC zero level signal remains. As a countermeasure against this, a high resistance R1 is provided. By superimposing a constant voltage from the high resistance R1 on the output of the sensor 100, a constant bias in the positive direction is provided. This resistor R1 plays an important role. That is, the output of the knock sensor 100 is ±5 mV.
Varies between ~±600mV. The range of change is 120 times. When attempting to attenuate the knock sensor output using a general bias means using resistive voltage division for a change range of 120 times, for example, if the knock sensor output is close to ±5 mV and the attenuation rate of the knock sensor output by the bias means is 1. /20,
The output after biasing will drop to 0.25mV. This obviously leads to a decrease in detection accuracy, which in turn makes it impossible to detect the failure mode. The present invention improves knock detection accuracy and facilitates failure mode detection by providing a high resistance R1. FIG. 2 shows the sensor output biased by resistor R1.

Now, when the knob sensor 100 shoots,
The voltage at the point is approximately 0V, which is lower than the voltage at the point.
The output voltage of the comparator OP2 becomes approximately 0V as shown in FIG. 24. When the output of the comparator OP2 becomes LOW, the transistor T80 is turned off. As a result, as shown in FIG. 2, V ₀
A fail-safe voltage is output from the terminal and is automatically retarded to the maximum retard amount or intermediate retard amount. Note that the driver can be easily notified by taking out the output of the comparator OP2 and inputting it to a warning display device (not shown).

AC coupling circuit 112 is provided to remove DC components. Furthermore, it may play a role in causing resonance in the knocking frequency region. By removing the DC bias in the AC coupling circuit 112, it makes a significant contribution to improving the noise cutting performance of the ignition noise cutting circuit 113 in the next stage.
This is because ignition noise cannot be cut unless the ignition noise is cut after removing the DC bias. The ignition noise cut circuit 113 is controlled by the pulse output of the monostable circuit 128 as shown in FIG. The noise cut circuit 113 is a type of AND gate, and inputs the output pulse of the monostable circuit 128. Therefore, only in the _t1 period when the output pulse of the monostable circuit 128 is "1", the sensor output obtained via the AC coupling circuit 112 is almost unchanged from the ignition noise cut circuit 1.
The output will be 13. This output is shown in FIG. 26. This cuts ignition noise.

The BPF 114 emphasizes the knock signal (attenuates other signals) and outputs it, and has a characteristic of slightly attenuating at frequencies higher than the knock signal of knocking. AGC circuit 160 is amplifier 1
In response to a feedback signal from the BGL detection circuit 119 via 61, the gain of the BGL detection circuit 119 is changed in inverse proportion to the feedback signal, that is, the BGL output. The mask circuit 150 masks the output of the AGC circuit 160 at predetermined timing. This masking is done by the pulse signal S2 of FIG. In response to the output of this mask circuit 150, the BGL detection circuit 119 detects BGL.
The comparison circuit 118 detects the BGL of the BGL detection circuit 119.
Comparator 1 compares the output (voltage) with the output of amplifier 151.
18A, and the comparator 118B compares the output of the amplifier 151 with the reference voltage. The state of both signals at this time is shown in FIG. 7. The comparator 118A of the comparison circuit 118 has an amplifier 1 larger than the BGL output.
Only the output of 51 is formatted and output. Further, comparator 118B removes the output of amplifier 151 at a level higher than the reference voltage applied to the negative terminal (this voltage is a reference value for limiting the abnormal amplitude of the knock signal). This clamps any abnormally high voltages in the knock signal. This comparator 1
The output of 18 is input to the mask circuit 153 and the retard signal output circuit 126 and converted to a DC level, and the DC level signal becomes the knock signal to be obtained. Here, the mask circuit 153 has the role of removing noise similar to a knock signal generated from the intake cylinder during high speed rotation. This mask is performed by S2. The output S6 of the comparator 118 is in the form of a single pulse, and the retard signal output circuit 126 has a function of averaging this single pulse.

FIG. 28 shows the knock signal detected by comparator 118. FIG. 29 is an enlarged view of a portion thereof. The integration circuit 125 is connected to the retard signal output circuit 12.
A predetermined integration is performed using the output signal of No. 6 as input. Therefore, an integral value corresponding to the number of knock pulses is output from the integrating circuit 125. The retard angle control of the retard circuit 132 is performed using this integral output. The above operation is a general operation except for the roles and functions of the signals S4 and S10. New operations based on signals S4, G10, etc. will be described later.

Figure 3 shows BPF114 from knock sensor 100.
A specific circuit example of the above configuration is shown. The knock sensor 100 is a capacitive sensor using a piezoelectric element, and equivalently constitutes a parallel circuit of a capacitor C and a constant current source.

The power supplied to the resistor 1 is supplied from the power supply section 120. The power supply section 120 includes resistors R11 and R1.
2, R14, R15, capacitor C5, and operational amplifier OP3. Operational amplifier OP3 has the role of buffer. The B power supply is applied to the positive end of the operational amplifier OP3 after being divided by resistors R14 and R15. The output of operational amplifier OP3 is set to 3 (V). Therefore, the DC bias voltage at point a is approximately 6 (V), and the output signal of the knock sensor 100 is superimposed on the 6 (V) voltage.

On the other hand, if the input impedance of the knock control device is increased, disturbance noise is likely to be superimposed. A typical example of disturbance noise is ignition noise (Ig noise) that occurs in synchronization with ignition timing.

The ignition noise of this device will be explained below.

Base control of the power transistor is performed by pulses as shown in FIG. The pulse is H
When the level is on, the power transistor is on (ON)
and turns off when it is at L level. This ON
In the process of switching from OFF to OFF, or at the time of OFF, the secondary voltage of the ignition coil rises rapidly, generating primary noise. Furthermore, this increase in secondary voltage breaks down the insulation of the air layer between the plugs, causing ignition. Secondary noise occurs during this ignition. The second-order noise includes noise due to a capacitive discharge current flowing at the initial stage of ignition, and noise due to an induced discharge current flowing at a subsequent stage. Among the second-order noises, the former noise is a major noise source. When the input impedance is increased, primary noise and secondary noise (the former noise) are superimposed on the knock sensor output as disturbance noise that adversely affects knock signal identification.

It is necessary to remove such disturbance noise. This disturbance noise continues for a time of about 50 to 60 μsec. Therefore, the knock sensor output may be masked during this time. In order to achieve this purpose, the AC coupling circuit 112 and the noise cut circuit 11
There are 3. However, the actual mask interval is sufficiently larger than the above noise duration time, e.g.
It is set to about 0.8msec.

The AC coupling circuit 112 includes a capacitor C1 and a resistor R.
It consists of 3. The noise cut circuit 113 is a resistor R
4, R5, R6, R8, R101, capacitor C
2. Consists of transistor T2 and operational amplifier OP1. The AC coupling circuit 112 is provided as a means for extracting a knocking signal from the knock sensor output signal, and by passing the knocking signal through this circuit 112, the DC bias voltage riding on the knock sensor output signal is removed. It's forcing me. If an attempt is made to extract only the knock signal from the knock sensor output obtained by superimposing a DC bias component, and if an attempt is made to apply the above-mentioned noise mask, the processing becomes extremely complicated. Although the idea of cutting off direct current is simple, it is an extremely practical technical means for correctly classifying knock signals.

The noise cut circuit 113 performs Ig noise cut mainly by the function of the transistor T2. Transistor T2 is turned on and off by output S2 of monostable circuit 128. monostable circuit 1
28 is triggered by the fall of the base signal of the power transistor shown in FIG. 2, and generates a pulse having the width of the mask section. FIG. 2 shows the output S2 of this monostable circuit 128, and the time width _t1 is the mask interval width. The transistor T2 is turned on only during the _t1 period when the output S2 of the monostable circuit 128 becomes "1". As a result, in this _t1 interval, the knock sensor output is shorted to ground, and the operational amplifier OP1
The input to is eliminated, creating a masking effect that masks Ig noise. Incidentally, as the load impedance of the sensor 100, the resistors R3, R4, R5 and the capacitors C1, C2 can be considered, but by setting the resistor R5 to a high resistance, for example, 1 MΩ, the resistors R4,
R5 and capacitor C2 can be ignored as loads. Therefore, the load on the sensor 100 can be considered to be only the capacitor C1 and the resistor R3.

BPF114 consists of resistors R7, R9, R10, capacitors C3, C4, C22, and operational amplifier OP2.
It consists of The non-inverting terminal of the operational amplifier OP2 is
Operational amplifier of power supply section 120 which plays the role of buffer
Connected to OP3 output RV (3V reference voltage).
Therefore, the BPF 114 output is DC biased to the reference voltage. this
The BPF 114 has a filter function to increase the level difference between the knock signal and the non-knock signal.
This facilitates identification by level.
The output S1 from the operational amplifier OP2 of the BPF 114 is input to a resistor R16 of an AGC circuit 160, which will be described later.

Next, the operation of the knock sensor failure detection circuit 102A will be explained. The sensor short detection circuit 109 is
It consists of a resistor R55 and a comparator CO2. Furthermore, it is assumed that the potential at point c of the comparator CO2 is set to approximately 0.2 (V).

Now, if the knock sensor 100 is shot, the potential at point a becomes approximately 0 (V) as shown in FIG. 2. Therefore, the potential at point a becomes lower than the potential at point c, and the output of comparator CO2 becomes approximately 0 (V). During normal operation when the lock sensor 100 does not shoot,
Since the potential at point a is set higher than the potential at point c due to the DC bias, the output of comparator CO2 is approximately at the potential at point a (meaning it is greater than 0 (V)).
It's getting old. In other words, when you shoot,
The g-point voltage becomes 0 (V). These results become the signal S3 from the transistor T80, and the integrator 12
The output to 5 becomes 0 (V). Thus, the comparator
Shoot failure can be detected by CO2.

One of the features of this device is the circuit configuration between AGC circuit 160 and comparison circuit 118. BPF1
The output S ₁ of No. 14 is input to the AGC circuit 160.
The output of the AGC circuit 160 is divided into two systems via the mask circuit 150. The first system is a system consisting of an amplifier 151 that amplifies the knock signal and inputs it to one input terminal of the comparison circuit 118.
The second system includes a half-wave rectifier circuit 116, a maximum clamp circuit 116A, an integration circuit 117, and an amplifier 117.
This is a BGL circuit 119 consisting of A. Amplifier 11
The output of 7A is input to the other input terminal of the comparison circuit 118. The output of amplifier 117A is output from amplifier 16
1 and is negatively fed back to the AGC circuit 160.

Knock sensor output is ±5 (mV) to 600 (mV)
The range is . In other words, the sensor output will fluctuate over a range of 120 times. When this output is simply amplified (eg, 100 times), it becomes ±0.5 (V) to ±60 (V). However, in an automobile, the maximum battery voltage is approximately 12 (V), and a value of 60 (V) is impossible.
Therefore, in the past, either a small gain was used to avoid saturation, or a process was performed with the expectation that saturation would occur. The former has the disadvantage that it is less sensitive to small human forces, and the latter has a disadvantage that it is less sensitive to large-amplitude human forces. The configuration of this embodiment is characterized in that an AGC circuit 160 is provided, and further, this AGC circuit 160 is provided on the output side of the BPF 114. With this configuration, the level difference between the knock signal and the non-knock signal is increased in the BPF 114, and this increased level difference is input to the AGC circuit 160 as it is, thereby obtaining an output with a good S/N ratio. be able to.

As is clear from the input/output characteristics shown in Figure 4,
The AGC circuit 160 can control the output V ₀ to be substantially constant except for low level output (±V _L ) and high level input (V _H ).

A specific example of the circuit from the AGC circuit 160 to the comparison circuit 118 is shown in FIG. The AGC circuit 160 is
Resistors R16, R17, R19, R18, R92,
It consists of R102, operational amplifier OP3, and FET T17. The half-wave rectifier circuit 116 includes a capacitor C
6. Resistor R20, R21, R22, diode
It consists of D ₁ , D ₂ and operational amplifier OP4.

The clamp circuit 116A includes resistors R25 and R2.
6, R27, R29, operational amplifier OP6, comparator
It consists of CO5, capacitor C8, and diode D10. Integrator 117 consists of resistor R29 and capacitor C9. The amplifier 117A is a resistor R3.
0, R31, R32, R33, R34, R35,
Consists of operational amplifier OP7. The amplifier 161 is
Operational amplifier OP8, resistors R93, R37, R38,
It consists of R36, R40, R39, and capacitor C10.

The operation of the circuit shown in FIG. 5 will be explained.
Output S1 filtered by BPF 114 is input to OP3 of AGC circuit 160 via resistor R16. A FET (T1) whose gain is controlled via an amplifier 161 is connected to the negative end of OP3.
7) is provided. As a result, AGC circuit 1
The gain of 60 is changed according to the output of OP7 of amplifier 117A. The output of the AGC circuit 160 is masked at a predetermined timing by the mask circuit 150, and is passed through C6 and R20 to the half-wave rectifier 1.
16. By the action of diodes D1 and D2, only the positive direction component is half-wave rectified and input to the maximum clamp circuit 116A. In this clamp circuit 116A, the voltage is clamped to a predetermined maximum value (corresponding to the voltage applied to the positive terminal via R27, for example, 5V) by the action of the comparator CO5. The output of this clamp circuit 116A is
The signal is integrated and smoothed by an integrating circuit 117 formed by a resistor R29 and a capacitor C9, and further amplified by an amplifier 117A and output. Next, it will be explained in detail.

The input voltage of the AGC circuit 160 (i.e. BPF 114
) is V ₁ , the output voltage is V ₀ , and EFT
If the output impedance of (T17) is Z _F , then
Since the value of the resistor R19 is set to a high resistance, the amplification rate of the operational amplifier OP3 is determined by Z _F and the resistor R19. That is, the gain G is as follows: G=V ₀ /V ₁ ≒ (1+R ₁₉ /Z _F )...(3'). The important point in equation (3′) is the impedance
Z _F is the gate-source voltage V _GS of FET (T15)
It changes depending on the situation. For example, V _GS is 0
(V) to -2(V), the amplifier 16
Due to the output feedback of 1, Z _F becomes large and the gain G becomes small. The opposite is true when it gets bigger. As a result, the BGL output voltage output from the amplifier 117A of the AGC circuit 160 remains constant regardless of fluctuations in the input voltage to the AGC circuit 160, and its SN ratio becomes nearly constant.

AGC works well in the range of V _GS = ±200 (mV). Note that the resistor R19 is for protection in the event of a disconnection failure of the FET (T15), and is set to have a high resistance. Furthermore, Z _F
The value is approximately 200Ω to 2KΩ.

The mask circuit 150 consists of a transistor T15 and a resistor R97. Transistor T15 is rendered conductive by signal S4 or S2 applied to its base.
AGC circuit 16 is activated by conduction of transistor T15.
The zero output drops to ground potential and is masked.
Signal S4 is the output of constant time width generation circuit 158, and signal S2 is the output of monostable circuit 128.

The resistor R95 of the amplifier 151 is an operational amplifier OP.
Perform 360 (mV) DC correction on the output of 9.
Furthermore, resistor R35 and capacitor C10 are BGL
It has ripple prevention function. Furthermore, through the overall diagram, the operational amplifier (OP) is Hitachi's HA17902,
The comparator (CO) uses Hitachi's HA17901.

Resistors R40, R39, R38 of amplifier 161
is, when there is no BGL input, the terminal of operational amplifier OP8 is connected to the output terminal (3V) of operational amplifier OP3, so the terminal input becomes 3 (V), and at this time, the output _V2 of operational amplifier OP8 is As shown in Figure 8, it has a DC correction function that sets the voltage to 4 (V). Furthermore, the operational amplifier OP8 is an inverting amplifier, and when the input V ₁ increases, the output V ₂ increases.
will be in the direction of decrease. Since this output V ₂ is the gate input of the FET (T15) of the AGC circuit 160, when BGL increases, V ₁ increases, V ₂ goes down, and the gain G also goes down, causing the AGC to operate. Here, the AGC circuit 160 starts working from V ₂ = 3 (V) and reaches 4 (V).
~3(V), Z _F remains constant.

Operational amplifier OP6 with maximum clamp circuit 116A
is a buffer provided for impedance conversion, and when the non-inverting terminal voltage _V1 attempts to exceed 5 (V), the comparator CO5 becomes conductive and the diode D
10. The current is discharged through the resistor R28, so that the terminal voltage of the non-inverting terminal of the operational amplifier OP6 is
Clamped to a maximum of 5 (V). By this,
The inverted output voltage (background voltage level) of the comparator CO5 does not change depending on the intensity of knocking, and there is no possibility that BGL will rise during knocking, making it difficult to detect knocking.

The comparison circuit 118 includes comparators 118A, 118
B, resistors R43 and R59, and diode D3. Comparator CO4 of comparison element 118A is amplifier 1
A comparison is made between the signal including the knock signal which is the output of 51 and the reference voltage divided by resistors R41 and R42. The reference voltage is set to 4 (V), for example. On the other hand, comparator CO3 is connected to BGL output and amplifier 1
A comparison is made with the output of 51. As a result, from the output of the comparator circuit 118, the voltage is higher than the reference voltage.
Output S6 of amplifier 151 larger than BGL output
outputs. Note that the diode D3 has a role of transmitting this output, and the resistor R59 has a role of a discharging resistor between it and a capacitor C13 of a retard amount setting circuit 126, which will be described later. The output S6 of the comparison circuit 118 is input to the mask circuit 153 at the next stage.

A specific circuit configuration from the mask circuit 153 to the integration circuit 125 is shown in FIG.

The lead angle signal setting circuit 127 includes a fixed lead angle setting circuit 127A and a variable lead angle setting circuit 127B. Fixed lead angle setting circuit 127A is resistor R56, R
68, R70, R98, R99, R100, and transistor T11. Variable advance angle setting circuit 12
7B consists of resistors R65, R69 and transistor T8. The advance angle output signal of the fixed advance angle setting circuit is determined by the power supply voltage D for advance angle at the time of starting. The lead angle output signal of the variable lead angle setting circuit 127B is determined by the output S9 of the monostable circuit 128. The output S9 is a signal with a pulse width proportional to the rotation speed, and therefore,
The advance angle output signal of the circuit 127B becomes an advance angle signal proportional to the rotation speed. The lead angle output signal of the lead angle signal setting circuit 127 is input to the integrator 140.

The output of the comparator 118 may be applied directly to the integration circuit 125, or may be applied to the integration circuit 125 via the retard signal output circuit 126. Further, by providing a mask circuit 153 between the retard signal setting circuit 126 and the comparator 118, a more accurate retard signal can be generated. This example discloses this case. The mask circuit 153 includes a resistor R5.
7, R58, and transistor T7. The retard signal output circuit 126 is the comparator 1 that has escaped the mask.
18 output pulses are smoothed and integrated, and the corresponding retard control is performed. The retard signal setting circuit 126 includes a capacitor C13, resistors R63, R64, R66,
It consists of R67 and transistors T9 and T10. Transistor T7 is turned on by signal S2 or S10, and the output S6 of comparator 118 at this time flows to ground via transistor T7 and is masked. When transistor T7 is off, signal S6 is accumulated in capacitor C13 and drives transistor T10 via resistor R63. Transistor T10 is also driven by signal S3 via resistor R64. The signal S2 is the output of the monostable circuit 128, and the signal S10 is the output of the fixed time generation circuit 157. Signal S3 is the output of abnormality detection circuit 102A. The power supply voltage D applied to the base of the transistor T9 is provided by a power supply circuit (described later). When starting the engine, the battery voltage drops below a predetermined minimum allowable voltage. The same applies when the battery capacity becomes low. When this abnormal voltage drop occurs, the voltage D becomes a high voltage, and when the voltage is normal, it becomes a low voltage. When the voltage D is high, the transistor T9 is turned on, and the transistor is connected to the resistor R63,
It remains off regardless of the signal applied through R64. On the other hand, when the voltage D is low, the transistor T9 is turned off, and as a result, the transistor T10 is turned on and off depending on the voltage value via the resistors R63 and R64.

The integration circuit 125 includes an integrator 140, a maximum voltage clamp circuit 141, and a minimum voltage clamp circuit 142.
Consists of. The integrator 140 is an operational amplifier OP15,
It consists of capacitors C14 and C15 and resistor R200. The maximum voltage clamp circuit 141 is an operational amplifier
OP16, resistors R71, R74, R75, R76,
It consists of R77, R78, diode D7, and transistor T20. Minimum voltage clamp circuit 142
is operational amplifier OP17, resistor R60, R61,
It consists of a diode D6.

Next, the circuit operation of the integrating circuit 125 will be explained.
Now, in response to the knock signal which is the output of the retard signal setting circuit 126 which receives the output of the comparator 118, the transistor T10 is turned on in synchronization with the knock signal. Therefore, as shown in FIG. 2, the transistor T10 is conductive during the pulse width _t0 (approximately 40 to 70 μsec) of the knock signal, and the current _i1 flows through the operational amplifier OP1.
From 5, capacitors C14 and C15, resistor R67,
It flows to ground via transistor T10.
Further, the output voltage of the operational amplifier OP15 at this time is 3 (V).

Therefore, 1 of the operational amplifier OP15 at this time
Voltage rise rate per pulse (voltage rise/1 pulse)
ΔV ₁ is as follows.

i ₁ = 3/R ₆₇ (4), ΔV ₁ = i ₁ /Ct ₀ (5) where capacitance C is the series capacitance value of capacitors C14 and C15. As is clear from this equation (5),
The output voltage of the operational amplifier OP15 increases in proportion to the number of knocking pulses.

On the other hand, every cycle, the inverted output S9 of the monostable circuit 128 changes from the transistor T6 to the transistor T.
8 and for a constant mask time t ₁ ,
Turn off transistor T8. Therefore, during this time,
Current _i2 flows from the power supply V ₊ to the operational amplifier OP via resistors R98 and R100 and capacitors C14 and C15.
Flows to 15. The Zener voltage of the Zener diode ZD4 is 6 (V). Also, op amp
The terminal of OP15 is set to -3 volts. Therefore, monostable circuit 12 is added to operational amplifier OP15.
Operational amplifier OP1 every time 1 pulse is input from 8.
The output voltage of No. 5 will fall according to the voltage fall rate (falling voltage value/cycle) ΔV ₂ below.

i ₂ = 6-3/R ₉₈ + R ₁₀₀ ……(6) Therefore, ΔV ₂ = i ₂ /Ct ₁ ……(7) This voltage drop rate ΔV ₂ takes into consideration the power performance such as engine torque and horsepower. The voltage increase rate ΔV ₁ is set to approximately 1/50. The maximum value of the output of the integrator is clamped by the clamp voltage of the maximum clamp circuit 141, and the minimum value is clamped by the clamp voltage of the maximum clamp circuit 14.
It is clamped by a clamp voltage of 2.

The integrating circuit 125 is configured to have a specific advance angle characteristic (advanced angle value) by turning on the transistor T11 due to the voltage D when the engine is started. This advance angle characteristic is determined by the integrator circuit 125 issuing a command and the retard circuit 132 determining the actual advance angle (retard angle).
Take control. This retard circuit 132 is, for example,
The following document (US Patent application, Ser. No.
80202, by Noboru Sugiura, filed october 1,
1979 and assigned to the assignee of this
application“Ignition timing control system
for internal combustion engine”) will be used.

The operation of the retard circuit 132 will now be explained.

In general, ignition timing characteristics are relative;
It is determined according to a certain operating mode determined by the distributor and the ignition system used. Furthermore, a maximum retardation characteristic is given at the time of knocking, and this characteristic is used when the engine is knocked. Figure 8 shows the lead angle and retard angle characteristics, where the solid line shows the minimum retard angle (i.e., minimum clamp voltage) characteristic in a certain driving mode, and the dotted line shows the maximum retard angle (i.e., maximum clamp voltage) characteristic when knocking. ing. At low speeds, e.g.
Below 200 rpm, control is performed to achieve the maximum advance characteristic determined by the ignition timing characteristics. The reason for adopting such characteristics is to ensure starting at startup. That is, at the time of starting, if the ignition timing is delayed, the engine generates reverse rotation torque, and the load on the starter becomes extremely large. As a result, the driving current of the starter becomes abnormally large, making it impossible for the starter to turn the engine, resulting in a so-called starting failure. In order to eliminate such startup failures, at startup, e.g.
Below 200 rpm, the maximum advance characteristic is determined by the ignition timing characteristic.

FIG. 9 shows the characteristics of the retard circuit 132 that should achieve the above characteristics. As shown in the figure, integrating circuit 1
25, that is, the output voltage of the integrator 140, has a retard characteristic so as to have a constant angle slope characteristic. Therefore, the advance angle is a constant angle every cycle. That is, the ignition timing is configured to advance by a constant angle every cycle while being retarded in accordance with the number of knocking pulses.

Next, the operation of the integrating circuit 125 that controls the retard circuit 132 will be described, and in particular, a countermeasure for starting the engine by advancing the starting angle will be described. Since this start-up countermeasure is related to the power supply circuit, the power supply circuit shown in FIG. 10 will be explained first.

This power supply circuit consists of a start detection power supply device 50 and an actual power supply device 51. It consists of a power supply device 50, resistors R86, R87, R88, R89, a Zener diode ZD3, a capacitor C19, and a diode D9. The power supply device 51 includes resistors 90, 9
1. Consists of capacitors C20 and C21, and Zener diodes ZD4 and ZD5. Battery power supply is BW
The Zener diode ZD5 cuts off voltages above a predetermined voltage (6.2V).
=6.2V is output. Voltages A and D are voltages that reflect starting detection. That is, the battery voltage decreases at the time of starting. When the amount of decrease exceeds the reference value, the transistor T14 is turned off, and the voltages A and D have the same value. The same operation occurs when the power capacity of the battery decreases. If the battery power supply voltage is normal, the transistor T14 is on,
The D voltage is approximately at ground potential, and the A voltage is at the resistor R8.
This corresponds to the drop voltage of 6. Resistor R86 is set to a relatively high resistance (22KΩ). This D
A voltage is applied to the base of the transistor T9 and the base of the transistor T11 to set a predetermined advance angle characteristic at the time of starting.

Further, a signal from the water temperature switch is inputted from the T terminal. That is, when the engine cooling water temperature is below a predetermined value (for example, 70° C.), a Low signal is input. Therefore, from the BW terminal, the resistor R221 is passed through the resistor R88 and the diode D12.
Current flows to the T terminal through the diode D12
The potential on the anode side of is lowered. Then, no current flows through the Zener diode ZD3, and the transistor T14 is cut off. This transistor T1
When transistor T4 is turned off, current flows through the base of transistor T9 connected to the collector side of transistor T14, turning it on. When the transistor T9 is turned on, the transistor T10 is turned off, so that the knock control is not activated. In other words, the engine is not retarded during warm-up operation.

Next, we will discuss the operation of the integrating circuit 125 that controls the retard circuit 132, and in particular, the countermeasures against starting by advancing the starting angle. Zener diode ZD
No. 3 has a Zener voltage of about 6 (V), and when the power supply voltage (V ₊ ) is low, that is, when the engine is started with the starter on, the midpoint voltage between resistors R88 and R89 makes it impossible to turn on the Zener diode ZD3. Therefore, transistor T14 is turned off, and transistors T9 and T10 are turned on. At this time, transistor T10 is turned off. Also, the transistor T11
When turned on, current flows from the power supply through resistor R70.
The current flows in the same direction as L ₂ , and the operational amplifier OP15
The output will be reduced to the same voltage as the k-point voltage and will be clamped. This k-point voltage corresponds to the minimum clamp voltage of 1.5 (V) shown in FIG. This clamped output sets the maximum retardation characteristic at the time of starting as shown by the dotted line in FIG. By this,
The retard circuit 132 is controlled and set to the maximum retard characteristic.

Next, the integration circuit 1 when the sensor 100 fails
The operation of 25 will be explained. The outputs from the sensor short detector 108 and the sensor open detector 109 shown in FIG. 3 turn off the transistor T7 shown in FIG. 7 and turn on the transistors T10 and T20. transistor T
When OP10 is turned on, the current _i1 continues to flow through the capacitors C14 and C15, as in the retarding operation described above, and therefore the output voltage of the operational amplifier OP15 becomes h.
It is clamped to the same voltage of 6V (maximum voltage) as the point voltage. Furthermore, as a result of transistor T20 being turned on, the voltage at point h is controlled to a fail-safe voltage of 5V, which is lower than the normal voltage. As a result, appropriate retardation characteristics can be obtained even in abnormal situations. Note that the fail-safe clamp voltage when the transistor T20 is omitted is 6V as shown in FIG.

FIG. 11 shows an operating waveform diagram when an abnormal voltage is superimposed on point a in FIG. 3. This abnormality detection is performed by the detection circuits 108 and 109 operating in common. 11 shows the base signal of the power transistor. When the knock sensor 100 becomes an abnormal signal for some reason, as shown in FIG. 11 2, a voltage higher than the voltage at point b is continuously generated. The output of the comparator continuously drops to 0 (V) as shown in FIG. ) is clamped.
Therefore, it is possible to adequately deal with abnormal voltage.

FIG. 12 shows a timing signal generation circuit that generates various timing signals. monostable circuit 128
are resistors R44, R45, R46, R47, R4
8, R49, R50, R51, R52, R53,
R54, capacitors C11, C12, transistors T3, T4, T5, T6, capacitors D4, D
Consists of 5. The rotation speed detection circuit 156 is a resistor R7
9, R80, R81, R83, R84, R96,
It consists of R103, R201, R202, R203, operational amplifier OP10, comparator CO6, capacitor C18, diode D8, and transistor T13.

The operation of monostable circuit 128 will be explained. A base signal of the power transistor shown in FIG. 2 is applied to the terminal P. When the base signal is high, the transistor T3 is turned on and the transistor T4 is turned off. By turning off the transistor T4, the capacitor C12 is connected to the power supply B → resistor R48 → R50 → C1.
2→A path to the base of transistor T5 is formed. On the other hand, when the base signal is L, the transistor T3
is off, transistor T4 is on, and power supply B
→ Resistor R51 → Capacitor C12 → Resistor R50 →
A path from D5 to transistor T4 to R49 to ground is formed. These two paths are capacitor C1
2, and a pulse S2 synchronized with the spark timing having a time width _t1 as shown in FIG. 2 is generated at the collector end of the transistor T5. Also, the transistor T6 is the transistor T5.
Since the pulse S has an opposite phase to the signal S2,
Outputs 9. This signal S2 is applied to the base of the transistor T2 of the ignition noise cut circuit 113 to serve as an ignition noise cut signal, and is also applied to the base of the transistor T7 of the mask circuit 153 to play the role of ignition noise cut. Signal 9 is the transistor T8 of the advance angle signal output circuit 127.
is applied to the base of the motor and is used for advance angle control.
Further, the signal S2 serves as an input signal to the rotation detection circuit 156.

The operation of the rotation speed detection circuit 156 will be explained. The transistor T13 is turned on when two conditions are met: the signal S2 is H and the transistor T3 is off.
As a result, it turns on with the pulse width _t1 shown in FIG. Since the period of this pulse is proportional to the rotation speed,
In the end, transistor T13 is driven according to the rotation speed. The positive terminal of the operational amplifier OP10 has the k-point voltage (approximately 1.7V) of the lowest voltage clamp circuit 142.
is applied. When the transistor T13 is turned on, a path from the output side of the operational amplifier OP10 to the capacitor C18→R80→T13→ground is created, and the capacitor C18 is charged. When transistor T13 is off, the charge in capacitor C18 flows to resistor R81. The operational amplifier OP10 generates an output corresponding to the deviation of the voltage applied to the positive terminal and the negative terminal, and the output is applied to the negative terminal of the comparator CO6. Also comparator CO
A constant voltage (3.0V) divided by resistors R84 and R83 is applied to the positive terminal of 6. A voltage of 1.7V or more and corresponding to the rotation speed is applied to the negative terminal of comparator CO6, and a constant voltage of 3V is applied.
compared to Comparator CO6 when 3V or more
The output becomes L, and becomes H when the voltage is 3V or less. The reference voltage of 3V is the voltage for high-speed rotation.
Specifically, the rotation speed corresponding to this voltage of 3V is
It is set to 3000rpm. Therefore, the output of comparator CO6 becomes H only when the rpm is below 3000 rpm.
When the rotation is below 3000 rpm, the transistor T20 of the clamp circuit 141 is turned on. transistor T
By turning on transistor T20, the voltage applied to the negative terminal of operational amplifier OP16 becomes lower than when transistor T20 is off. In addition, diode D8, resistor R10
3 has hysteresis characteristics,
It takes time for this circuit to respond to 3000rpm, and during this time the rotational speed may rise slightly, so the output is adjusted to account for this increase.

Therefore, according to this embodiment, with a simple circuit,
It is possible to detect the shot of the knock sensor regardless of the rotation speed.

As described above, according to the present invention, the shot of the piezoelectric knock sensor can be detected in any engine rotation range.

[Brief explanation of drawings]

Figure 1 is an overall configuration diagram of the present invention, Figure 2 1 to 10
are time charts, Figures 3, 5, 7, 10, and 12 are specific circuit examples, and Figures 4, 6, 8, and 9 are characteristic diagrams. , Figs. 11-4 and Figs. 13 1-7 are time charts. 100... Knock sensor, 101... Knock control device.

Claims (1)

[Claims] 1 In a device that controls engine ignition timing based on an output signal from a knock detection circuit that detects the occurrence of a knock based on a signal output from a piezoelectric knock sensor and outputs a signal, the piezoelectric knock sensor is biased with a high resistance. A shot of a piezoelectric knock sensor, characterized in that the piezoelectric knock sensor is configured to transmit signals through capacitor coupling, and detects the shot of the piezoelectric knock sensor when the transmitted signal drops to near zero potential. detection circuit.