JPH0152993B2 - - Google Patents

Description

[Detailed Description of the Invention] [Technical Field of the Invention] The present invention is capable of improving the power factor and achieving polyphase balancing for a multiphase unbalanced low power factor load, as well as compensating harmonic ripple of the load. The present invention relates to a power adjustment device.

[Technical background of the invention and its problems]

For a multiphase AC power supply, the current values of each phase are equally balanced with respect to the load, there is no harmonic ripple, and at the same time the phase difference between the current and voltage is zero, that is, the power factor is 1. This is desirable. If the above requirements are not met, this will lead to a significant increase in the power equipment capacity and a decrease in the power utilization rate.

However, multi-phase unbalanced low power factor loads such as arc furnace equipment and stationary Leonard equipment using thyristors consume a large amount of reactive power, and due to its fluctuations, it is It has problems that have a negative impact. Furthermore, in loads such as cycloconverter devices, high frequency components other than the power supply frequency are generated, causing induction disturbances such as radio noise, and increasing harmonic loss in the system.

Conventionally, phase advance capacitors and phase adjusters have been mainly used for the purpose of power factor improvement, but these have had problems such as increased power loss, poor response to load fluctuations, and compensation accuracy reaching its limit. Ta. This greatly limits the conditions for its use.

On the other hand, technology aimed at balancing three-phase power is
It is introduced in Japanese Patent Application Laid-Open No. 52-87650. The "power adjustment device" shown here uses an automatic converter to compensate for reactive power, including harmonics, and also shows that it can perform three-phase balancing of the current supplied from the power supply side. . However, automatic converters require complicated circuits for forced commutation, making them expensive devices. Additionally, there is a possibility that commutation may fail due to ignition noise, etc., resulting in a lack of reliability as a system.

In addition, recently, a power active filter using a separately excited converter (1983 National Conference of the Institute of Electrical Engineers of Japan, Paper No.
608) was announced, showing that it is possible to compensate for reactive power including harmonics. However, the converter used here always requires a positive group converter and a negative group converter to operate in pairs, and in order to supply compensation current to a three-phase power supply,
At least six separately excited converters must be prepared. Therefore, the structure of the device becomes complicated and expensive.

[Purpose of the invention]

The present invention has been made in view of the above, and is capable of balancing multiphase power and compensating reactive power including harmonic components using only separately excited converters, and moreover, reduces the number of separately excited converters compared to the conventional device described above. The purpose is to provide a power adjustment device that can be implemented with half the number of devices.

[Summary of the invention]

In order to achieve this object, the present invention provides a power adjustment device that includes a multiphase AC power source, a high frequency reactive power source composed of a capacitor or a capacitor and an AC reactor, and an AC power source connected to the high frequency reactive power source through an isolation transformer. A converter consisting of at least three AC/DC power converters having side terminals connected to each other and configured so that a unidirectional circulating current flows through the DC side terminals via a DC reactor, and a DC power supply between the multiphase AC power source and the converter. It is composed of an AC reactor provided between the side terminals, and provides a command value for the multiphase balanced current supplied from the multiphase AC power supply according to the deviation between the voltage detection value of the high frequency reactive power source and the command value. It is characterized by the following.

[Embodiments of the invention]

FIG. 1 is a main circuit configuration diagram showing an embodiment of the power adjustment device of the present invention. In the figure, U, V, W are the electric lines of the 3-phase AC power supply, Load is the 3-phase unbalanced load, TR _S
is a power transformer, AL ₁ , AL ₂ , AL ₃ are AC reactors, L ₁ , L ₂ , L ₃ are DC reactors, SS1, SS
2, SS3 is an AC/DC power converter (separately excited converter),
TR _C is an isolation transformer, CAP is a phase advance capacitor that serves as a source of high frequency reactive power, CT ₁ to CT ₆ are current transformers,
_PTS and _PTO are transformers, and D is a rectifier circuit.

FIG. 2 is a block diagram showing an embodiment of the control circuit of the apparatus shown in FIG. In the figure, C _O , C _U , C _V are comparators, G _O (S), G _U (S), G _V (S) are control compensation circuits,
ML _U and ML _V are multipliers, A ₁ , A ₂ , A ₃ are adders,
K ₁ and K ₂ are inverting amplifiers, PH-1, PH-2, PH
-3 indicates a phase control circuit, and PTG indicates a three-phase sine wave generator.

The three-phase load Load in Figure 1 is the three-phase unbalanced current I _LU ,
It is assumed that I _LV and L _LW are taken, and the magnitude of each phase current and the phase angle with respect to the voltage are different. However, since it is a 3-phase-3-wire system, I _LU + I _LV + I _LW =
It is assumed that O is satisfied. FIG. 3a shows a voltage-current vector diagram at that time. V _U , V _V V _W are the voltages of each phase of the power supply, whereas the load currents I _LU , I _LV ,
When I _LW flows, the compensation currents I _CU , I _CV ,
It is shown that by supplying I _CW , three-phase balanced currents I _SU and I _SV are supplied from the power supply.

Hereinafter, a method of controlling the compensation current using the power adjustment device of the present invention will be explained with reference to the embodiments shown in FIGS. 1 and 2.

Separately excited converters SS1, SS2, SS3 and DC reactors L ₁ , L ₂ , L ₃ in Figure 1 constitute a triangularly connected circulating current type cycloconverter, which is located between the DC reactors L ₁ , L ₂ , and L ₃ . From the tap, AC reactors AL ₁ , AL ₂ , AL ₃ and power transformer
Compensation currents I _CU , I _CV and I _CW are supplied to the three-phase AC power supply via TR _S.

In addition, separately excited converters SS1, SS2, and SS
The AC side terminal of No. 3 is connected to a phase advancing capacitor CAP, which is a source of high frequency reactive power, via an isolation transformer _TRC .

The frequency of the reactive power source is determined by the firing control frequency of the triangularly connected cycloconverter, and the cycloconverter circulates so that the leading reactive power taken by the phase advancing converter CAP and the lagging reactive power taken by the cycloconverter cancel each other out. Current flows.

As mentioned above, the voltage of the reactive power source can be established by first taking in active current from the AC power source, and after that, the active power corresponding to the loss can be supplied from the AC power source so that the voltage becomes almost constant. .

The AC voltage of the reactive power source is detected by the transformer _PTO ,
It is made into a DC voltage V _O through a rectifier D. The detected voltage V _O is input to the comparator C _O shown in FIG. 2 and compared with the set voltage value V _O ^* . The deviation ε _O =V _O ^* −V _O becomes the peak value command I _n of the effective current via the control compensation circuit G _O (S). The control compensation circuit G _O (S) is usually composed of an integral element, and controls so that the steady-state component of the deviation ε _O becomes zero.

Above, the peak value command I _n is given by the following multipliers M _U and
It is input to ML _V and multiplied by unit voltage sine waves V _U ′ and V _{V ′} synchronized with AC power supply voltages V _U and V _V . Note that the unit voltage sine waves V _U ′ and V _V ′ are obtained by a transformer _PTS connected to the power supply terminal.

That is, the outputs of the multipliers M _U and M _V are as shown in the following equations.

I _PU = I _n ×V _U ′ = I _n・sinωst I _PV = I _n ×V _V ′=I _n・sin (ωst−2〓/3) Next, adder A ₁ calculates the U-phase load current. Detected value
Add I _LU and the inverted value of the output I _PU of the multiplier ML _U - I _PU to obtain the command value of the U-phase compensation current I ^* _CU = I _LU - I _PU .

Similarly, adder A ₂ adds the V-phase angular load current detection value I _LV and the inverted value of the output I _PV of the multiplier ML _V - I _PV to obtain the V-phase compensation current command value I _*CV = Find I _LV − I _PV .

The U-phase compensation current _ICU is controlled as follows.

First, the U-phase compensation current detection value _ICU and _the command value I ^* _CU are input to the comparator CU, and the deviation _εU =I ^* _CU − _ICU is determined. The deviation ε _U is input to the next control compensation circuit G _U (S), and ratio sequence amplification or integration is performed. One of the outputs e _U of the control compensation circuit G _U (S) is input to the phase control circuit PH-1 via the inverting amplifier K ₁ and the adder A ₃ , and the other is input to the direct phase control circuit PH- 2 is input. For convenience of explanation, the discussion will proceed assuming that the V phase and the phase input signal _eV from the compensation current control circuit are zero.

When the command value I ^* _CU is larger than the detected value / I _CU , the deviation ε _U = I ^* _CU − I _CU becomes a positive value, and the phase input signal e _U
Increase the value of Then, converter SS-2,
The output voltage V ₂ of the converter SS-1 generates a positive voltage proportional to the input e _U , and the output voltage V ₁ of the converter SS-1 is proportional to the input -
e Generates a negative voltage proportional to _U. Note that the output voltage V ₃ of the converter SS-3 generates zero voltage because the input e _V =0. Therefore, V ₁ = −V ₂ , V ₂
= 0, and the compensation current I _CU =-I _CW is determined by the voltage V ₂ as follows: SS2 → L ₁ → AL ₁ → TR _S → AL ₃ → L ₃ → SS2
increases along the path of , and settles down to become equal to the command value I ^* _CU . At this time, the V-phase compensation current I _CV becomes I _CV =0 because the output voltage V ₁ of the converter SS-1 and the output voltage V ₂ of the converter SS-2 are balanced.
That is, it does not affect the V-phase current control system.

When the command value I ^* _CU becomes smaller than the detected value I _CU , control is performed in the same way, and finally I ^* _CU = I _CU .

Next, the control operation of the V-phase compensation current _ICV will be explained as follows.

Input the V-phase compensation current detection value I _CV and its command value I ^* _CU to comparative C _V and find the deviation ε _V = I ^* _CU − I _CV . The deviation ε _V is input to the next control compensation circuit G _V (S), and proportional amplification or integration is performed. Control compensation circuit G _V (S)
One of the outputs _eV is input to the phase control circuit PH-3 via the inverting amplifier _K2 , and the other is input to the adder.
It is input to the phase control circuit PH-1 via _A3 .
The discussion will proceed with the initial condition being that the U-phase compensation current _ICU is controlled to be equal to its command value I ^* _CU .

If the command value I ^* _CU is larger than the detected value I _CV , the deviation ε _V
becomes a positive value and increases the value of the phase input signal _eV . Then, the output voltage V 1 of converter SS- ₁ generates a positive voltage in proportion to the input e _V , and the output voltage V 3 of converter SS- ₃ generates a negative voltage in proportion to the input -e _V. do. Therefore, depending on the voltage V ₁ , the current I _CV becomes SS−1 → L ₂ → AL ₂ → TR _S → AL ₁ → L ₁ →
It begins to flow along the path of SS1, and as a result, I ^* _CU > I _CU , increasing V ₂ in the positive direction and decreasing V ₁ . As a result, the W-phase compensation current I _CW flows in the direction opposite to the arrow, and finally I _CU = I ^* _CU , I _CV = I ^* _CV ,
It settles down as I _CW = −I _CU −I _CV .

When I ^* _CV < I _CV , the control is performed in the same way, and finally I ^* _CV = I _CV and I ^* _CU = I _CU .

Since the W-phase compensation current I _CW is uniquely determined as a result of the above-mentioned U-phase and current control, there is no need to configure a control loop.

The output voltages of the converters SS1, SS2, and SS3 are balanced as V ₁ +V ₂ +V ₃ =0, except for transient cases.

When V ₁ +V ₂ +V ₃ >0, the circulating current I _O of the triangular connected cycloconverter increases, and conversely, V ₁ +V ₂
When +V ₃ <0, the circulating current I _O decreases.

The delayed reactive power Q _CC of the cycloconverter seen from the high-frequency power supply side is proportional to the magnitude of the above-mentioned circulating current I _O.

FIG. 4 shows an equivalent circuit on the high frequency power supply side, and the part surrounded by the broken line is the cycloconverter. When the circulating current I _O increases, the delayed reactive power Q _CC increases, causing the value of the variable inductance L _CC to decrease. Therefore, the resonance frequency, that is, the oscillation frequency _O due to the capacitor CAP and the inductance L _CC increases. Conversely, when I _O decreases, Q _CC also decreases, which acts to increase L _CC and reduces the oscillation frequency _O.

In fact, the circulating current _{I O is automatically determined so that the oscillation frequency O} matches the output frequency _O ^* of the three-phase sine wave generator PTG in FIG. The principle is as follows.

The three-phase sine wave generator PTG in Fig. 2 provides a phase reference to the phase control circuits PH-1, PH-2, and PH-3, and its output frequency _O ^* is, for example, 1k.
Hz is constant. Here, to simplify the explanation, it is assumed that the phase control input signals e _U and e _V are constant. Therefore converters SS-1, SS-2 and SS-3
The firing phase angles α ₁ , α ₂ and α ₃ of are constant and the output voltage
It generates V ₁ , V ₂ and V ₃ .

Figure 5 shows the output waveform and firing timing for one phase of the three-phase sine wave generator PTG, and the phase angle α ₁
is controlled by. As shown by the broken line, when the oscillation frequency _O of the high frequency power supply becomes lower than the output frequency _O ^* of the three-phase sine wave generator PTG, the broken line (actual voltage)
The firing phase angle α ₁ ′ for the solid line is the firing phase angle α _{1 ′} for the solid line.
The output voltage V ₁ of the converter SS-1 increases in the positive direction. Similarly, converter SS-2,
The output voltages V ₂ and V ₃ of SS-3 also increase, and V ₁ +V ₂ +
V ₃ >0. Therefore, the circulating current I _O increases, the variable inductance L _CC in FIG. 4 decreases, and the oscillation frequency _O increases. Eventually, a circulating current I _O flows such that _O ^* = _O and settles down. When _O becomes higher than _O ^* , V ₁ +V ₂ +V ₃ <0, I _O decreases, _O becomes lower, and _O = _O ^* , which settles down.

That is, the oscillation frequency _O of the high frequency power source is controlled to be equal to the output frequency _O ^* of the three-phase sine wave generator PTG shown in FIG.

Next, the operation of voltage control of the high frequency power supply will be explained.

First, the case where the load currents I _LU , I _LV and I _LW are zero will be described. Initially, a command value V _O ^* >0 is given for V _O =0. Then, the deviation ε _O =V _O ^* becomes a positive value and increases the peak value I _n of the effective current command value. Therefore, U
The active current commands I _PU and I _PV of the phase and V phase are given as follows.

I _PU = I _n・sinωt I _PV = I _n・sin (ωt−2π/3) Since the load currents I _LU and I _LV are both zero, the compensation current command values are respectively I _*CU = −I _PU , I _*CV = -I _PV , and the effective currents I _PU and I _PV are taken from the power supply. At this time, the W-phase compensation current I _CW naturally becomes -(I _CU +I _CV ). Of course, the power supply current I _SU = I _PU , I _SV = I _PV , I _SW =
I _PW becomes a three-balanced current and the reactive power is zero.

As a result of taking effective current from the power supply, the energy is transferred to the phase advance capacitor CPA, which is a high frequency power supply.
is stored in , increasing its voltage _VO . At this time, the frequency _O is a three-phase sine wave generator, as mentioned earlier.
Matches the PTG output frequency _O ^* .

When the voltage V _O increases and becomes higher than the command value V _O ^* , the deviation ε _O becomes a negative value, and the peak value I _n decreases, furthermore, I _n <0. Therefore, I _PU = −I _n・sinωt I _PV = −I _n・sin (ωt − 2π/3) I _PW = (I _PU + I _PV ) = −I _n sin (ωt + 2π/3), and the effective current Return to AC power. Therefore, energy returns from the high frequency power source to the AC power source,
Voltage _VO decreases. In this way, V _O ^* = V _O
It calms down. If the command value V _O ^* is changed, the voltage V _O will follow accordingly.

Next, the load current shown in the vector diagram in Figure 3
The control operation when I _LU , I _LV , and I _LW flows will be explained.

Initially, V _O ^* = V _O , so the deviation ε _O = V _O ^*
−V _O becomes zero. Therefore, I _PU =I _PV =0, and the compensation currents are controlled to I ^* _CU = I _LU and I ^* _CV = I _LV , respectively. In other words, the entire load current is compensated. As a result, the energy of the high frequency power supply is reduced and the voltage
V _O also decreases. Then, ε _O =V _O ^* −V _O becomes a positive value, increases _In , and increases the effective current commands I _PU and I _PV . Therefore, the compensation currents are I ^* _CU = I _LU − I _PU , I ^* _CV =
Controlled by I _LV − I _PV , from AC power supply, I _SU = I _PU ,
I _SV , I _SW = -(I _PU + I _PW ) are now supplied,
The voltage V _O of the high frequency power supply also increases, and V _O = V _O ^*
I get used to it and calm down.

Finally, from the AC power supply, the three-phase balanced effective current I _SU ,
I _SV and I _SW are supplied, and the compensation currents are I _CU , I _CV , and I _CW . At this time, as shown in Figure 3b, I _CU + I _CV
+I _CW =0 is satisfied.

When the load currents I _LU , I _LV , _and I _LW change, they are controlled in ^the _same way, and 3
A phase balanced current is supplied.

Figure 6 shows each partial waveform when the load current I _LU includes harmonic components, where V _U is the U-phase power supply voltage, I _LU is the U-phase load current, and I ^* _CU is the phase compensation current command value. ,
_ISU represents the U-phase power supply current.

When I _PU = I _SU is given as the effective current command, the compensation current command value becomes I ^* _CU = I _LU - I _PU , and a command is given to compensate for the harmonic components of the load current as well. Therefore, by controlling the actual compensation current I _CU to be equal to the above command value I ^* _CU , the power supply current I _SU
is only the effective current component without harmonics. V
The same applies to the phase and W phase.

The above explained the 3 AC phase power supply, but 2
Needless to say, the present invention can be similarly applied to phase or six-phase power supplies.

Furthermore, although only the phase advancing capacitor CAP was connected to the high frequency power supply, the same operation is possible by connecting an AC reactor in parallel or series to the capacitor. For parallel reactants, the circulating current _IO decreases, and for series reactants, _IO increases.

〔Effect of the invention〕

As described above, the power adjustment device of the present invention can balance a multiphase AC power supply, compensate for load current harmonics, and set the power source power factor to 1 using only a separately excited converter. Furthermore, compared to conventional devices, the number of separately excited converters can be halved, making it possible to reduce the size and weight of the system.

Further, since the command value of the three-phase balanced effective current is determined according to the voltage deviation of the high-frequency power source, a special power calculation device is not required, and a power adjustment device with excellent quick response can be provided.

[Brief explanation of drawings]

FIG. 1 is a main circuit configuration diagram showing an embodiment of the power adjustment device of the present invention, FIG. 2 is a block diagram showing a control circuit of the device in FIG. 1, and FIGS. 4 is an equivalent circuit diagram for explaining the operation of the device in FIG. 1, FIG. 5 is an ignition timing diagram, and FIG. 6 is a diagram for explaining the operation of the device. FIG. 2 is a voltage and current waveform diagram for explaining the operation of the device shown in FIG. 1. FIG. U, V, W...3-phase AC power supply, Load...3-phase unbalanced low power factor load, TR _S ...Power transformer,
AL ₁ , AL ₂ , AL ₃ ... AC reactor, L ₁ , L ₂ ,
L ₃ ...DC reactor, SS1, SS2, SS3...
Separately excited converter, TR _C ...Isolation transformer, CAP
... Phase advance capacitor, PT _S , P _O ... Transformer,
CT ₁ to CT ₆ ... AC generator, D ... Rectifier circuit, C _O ,
C _U , C _V ... Comparator, A ₁ , A ₂ , A ₃ ... Adder, G _O
(S), G _U (S), G _V (S)...control compensation circuit,
ML _U , ML _V ...multiplier, _K1 , _K2 ...inverting amplifier, PH-1, PH-2, PH-3...phase control circuit, PTG...3-phase sine wave generator.

Claims (1)

[Claims] 1. A high-frequency reactive power source consisting of a polyphase AC power supply, a capacitor or a capacitor and an AC reactor, an AC side terminal connected to the high frequency reactive power source via an isolation transformer, and a DC side terminal connected via a DC reactor. a converter consisting of at least three AC/DC power converters configured to allow a unidirectional circulating current to flow; and an AC reactor interposed between the multiphase AC power supply and the DC side terminal of the converter, A power adjustment device characterized in that a command value of a multiphase balanced current supplied from the multiphase AC power source is given according to a deviation between a voltage detection value of a reactive power source and a command value.