JPH0198005A - Power saving controller for inductive load - Google Patents

Abstract

PURPOSE:To control the electric power for inductive load and to attain the high energy saving effect by using a auto-transformer type voltage regulator, a DC exciting power supply, a semiconductor switch, a load detecting circuit, and a switch control circuit. CONSTITUTION:An auto-transformer type voltage regulator 24 is set between an AC power supply 12 and an inductive load 18 and an output wiring is formed with use of a serial wiring 46, a serial wiring 50 having the adverse polarity to the wiring 46 and a shunt wiring 48. A DC exciting power supply 28 is set at the input side of the load 18 and an AC reactor 82 takes out the component dependent on the output voltage. While the component dependent on the inductive load current is taken out by a current transformer 80. These components undergo the vector synthesization and rectification and the DC exciting current of a control wiring 26 of a voltage controller 24 is supplied via a semiconductor switch circuit 30. At the same time, an inductive load detecting circuit 32 produces an output signal in accordance with the load state of the load 18. Then a control circuit 34 controls the conduction rate of a switch element 88 of the circuit 30 in response to said output signal. In such a constitution, the load voltage is instantaneously controlled at an optimum level in response to the due state. Thus the load 18 is always driven at the maximum power factor and the high energy saving effect is ensured.

Description

DETAILED DESCRIPTION OF THE INVENTION [Object of the Invention] The present invention relates to an AC power control device, and particularly to a power saving control device for an inductive load such as an AC induction motor.

[Prior art]

Conventionally, for the purpose of energy saving of AC induction motors and other inductive loads, U.S. Patent No. 4,052,6
No. 48 and No. 4,337,640 propose changing the input voltage of an induction motor by phase control to improve the power factor.

In these power control devices, the phase of the AC voltage supplied to the load is directly controlled by the thyristor, so the load current contains many harmonic components, and this harmonic current flows through the power capacitor and linear reactor of the power control device. Inflow,
These devices caused problems such as abnormal noise, vibration, overheating, and damage. Moreover, the waveform of the received power supply voltage is distorted by the harmonic current, causing great trouble to information devices such as computers and other control devices. The thyristor is fired in synchronization with the voltage in every cycle, but the synchronization signal for firing the thyristor is taken from the power supply voltage.

The synchronization signal sometimes fluctuates due to this waveform distortion. Therefore, depending on the state of the load, control becomes unstable, or in some cases becomes uncontrollable, resulting in safety and reliability problems. In order to solve this problem, U.S. Patent No. 4,602,200
The issue proposed installing a harmonic filter, but this device required a large number of large-capacity capacitors, reactors, and resistors, making the entire device large and extremely expensive to manufacture. Next, when starting an induction motor or induction coil, a large starting current that is more than 6 times the rated current of the motor flows, so 1
1! The capacity of the power semiconductor element is 2 to 2 of the rated capacity of the inductive load.
It is necessary to select one that is four times as large as the conventional one, and as a result, the semiconductor element becomes expensive, and the control circuit therefor inevitably becomes larger and more complicated, resulting in poor responsiveness.

Furthermore, a control device that also uses a thyristor as a semiconductor element has the disadvantage that a large internally generated loss occurs in the main circuit portion, resulting in increased power consumption in the main circuit portion. In particular, the main circuit requires a forced commutation circuit consisting of a commutation reactor and a commutation capacitor, and losses occur due to the energy transferred each time the commutation occurs within the commutation circuit.

In addition, power consumption was large due to losses in the main circuit snubber circuit (resistance, diode, etc.), losses in the smoothing reactor, AC reactor, etc. (iron loss, copper loss, etc.), and internal loss of the capacitor. As described above, in conventional power control devices, the power consumption of the device itself is large, so that the energy saving effect of AC induction motors and other inductive loads has been small.

[Purpose of the invention]

SUMMARY OF THE INVENTION An object of the present invention is to solve the above-mentioned problems and provide a power-saving control device for an inductive load that has a high energy-saving effect.

Another object of the present invention is to provide a power-saving control device for an inductive load that is small, lightweight, and inexpensive.

Another object of the present invention is to provide a power-saving control device for an inductive load that can quickly respond to fluctuations in an inductive load such as an AC induction motor.

Another object of the present invention is to provide a power-saving control device for an inductive load that prevents distortion to a sinusoidal AC waveform.

Another object of the present invention is to provide a power saving control device for an inductive load that can automatically drive the inductive load at the highest power factor in response to the load condition of an AC inductor motor.

Another object of the present invention is to provide a power-saving control device for an inductive load that is small, lightweight, and low cost.

Another object of the present invention is to provide a power-saving control device for an inductive load that has a large overload capacity, high stability and reliability, and does not require maintenance or inspection.

[Structure of the invention]

The power saving control device of the present invention includes an output winding connected between an AC power supply and an inductive load, and a control winding for adjusting the output voltage of the output winding. an AC reactor connected to the input side of the inductive load to take out a component dependent on the output voltage, a current transformer to take out a component dependent on the current of the inductive load, and a current that is a vector combination of both components. A DC excitation power supply comprising a rectifier that rectifies and supplies DC excitation current to the control winding.

connected between the control winding and the DC excitation W1g,
a semiconductor switch that controls the DC excitation current supplied to the control winding; a load detection circuit that generates an output signal corresponding to the load state of the load; and a load detection circuit that controls the conduction of the semiconductor switch in response to the output signal. The present invention is characterized by comprising a control circuit that controls the flow rate.

〔Example〕

The present invention will be described in detail below with reference to one drawing.

In FIG. 1, a power-saving control bag for inductive load according to a preferred embodiment of the present invention is shown! 10 is an input end 14.16 connected to the AC power supply 12, an output end 20.22 connected to the inductive load 18, and a control winding that variably adjusts the output voltage supplied to the 1M conductive negative load 18 according to the load state. an autotransformer type voltage regulator 24 having a wire 26; and a DC excitation power supply 28 that supplies a DC excitation current to the control winding 26.

A semiconductor switch circuit 30 that is connected between the control winding 26 and the DC excitation current g28 and that varies the DC excitation current supplied to the control winding 26, and a load detection circuit 32 that generates an output signal corresponding to the load state. and a control circuit 34 that controls the conduction rate of the semiconductor switch circuit 30 in response to the output signal and adjusts the output voltage in response to the load state.

In Figures 1 to 5, autotransformer type voltage control! 1 pseudo 24 is a first saturable iron core that constitutes the main magnetic flux loop path, and a magnetic shunt iron core 44 for bypassing a part of the main magnetic flux loop path.
The first saturable core 42 is made of a wound core. 1st
The saturable core 42 has a first series winding 46 and a shunt winding 48.
and an output winding consisting of a second series winding 50. 1st
The series winding 46 is connected between the high voltage terminal connected to the output end 20 and the medium voltage terminal connected to the input end 14, and the first shunt winding 48 is connected in series with the first series winding 46 with the same polarity. The lower end of the zero shunt winding 48 is connected to a neutral point connected to the input terminal 16. The second series winding 50 is connected between the lower end of the shunt 948 and the output end 22 with opposite polarity to the first series winding 46 . A second saturable iron ll152 comprising a wound iron core for controlling the magnetic flux density of the magnetic shunt iron core 44 by changing the magnetic saturation state of at least a part of the main magnetic flux loop path.
is controlled by the control winding 26 as described below.

In FIGS. 2-3, the first saturable core 42 includes a center leg 54 and outer curved legs 56, 58 that constitute a main magnetic flux loop path. The center leg 54 includes a first core portion 54a and a second core portion 54b separated by the magnetic shunt core 44. Furthermore, the center leg 54 includes an extension portion extending outwardly from the outer legs 56, 58, ie, a third core portion 54c. The center leg 54 is disposed above the first saturable core 42 and the fixture 6
0.62 and are fixed to each other and integrated. Section 2.3.
As is clear from FIG. 5, the magnetic shunt core 44 is made up of a C-shaped cross-sectional core made by laminating a large number of silicon steel plates. ′&
& 1l144a of 44 is center leg 5
4 and magnetically coupled with each other. End portion 44b of magnetic shunt core 44.

44c is an interlayer 64 having a required thickness corresponding to a constant air gap between the first coil block of the first series winding 46 and the shunt winding 48 and the second coil block of the second series winding 50. , 66 on the outer legs 56, 5B of the first saturable core 42, and is fixed to the outer legs 56, 58 by fixtures 68, 70, so that each core is integrated. A magnetic shunt core 64 shunts a portion of the magnetic flux in the main flux loop path through high reluctance gaps (formed by intersperses 64, 66) to the outer legs 56, 58 to provide an output voltage. It also functions to attenuate harmonics and reduce waveform distortion of the output voltage. The second saturable core 52 is the magnetic shunt core 5
4, i.e. the first series winding 46 and the first coil block of the shunt winding 48 and the second series winding 50.
The cores are arranged between the second coil block of the center leg 54 on the upper part of the second core part 54b and the lower end of the third core part 54c, and the respective cores are integrated by fixing devices 72 and 74. magnetically coupled. In this way, the second saturable core 52 is arranged so as to overlap the lower half of the first saturable core 42, and magnetically saturates a part of the first saturable core to increase the magnetic flux of the second series winding LA50. First series winding 4
6 and the shunt winding 48, and functions to shift the magnetic flux of the first series winding I@46 and the shunt winding 48 to the magnetic shunt core 44.

First series winding 1@46 and shunt winding 48. The second series winding 50 and the control winding 26 are wound on the first to third core portions 54a, 54b, 54a of the center leg 54, respectively, and are arranged substantially in the same plane. Further, the upper surface and the lower surface of each winding are arranged so as to be aligned with the upper surface of the second saturable iron core 52 and the lower surface of the first saturable iron core 42, respectively. That is, a second coil block consisting of a coil block of the first series winding 46, a shunt winding 48, a second series winding 50, a third coil block of the control winding 26, a center leg 54, It is arranged substantially within the thickness of the first and second saturable cores 42.52. 3rd of center leg 54
The core portion 54c extends outside the first saturable core 42, and the control winding 26 is attached to the lower end 54c of the center leg 54.
wrapped on top. The second saturable iron core 52 is the second T1. The second coil block of column winding 50 and the third coil block of control winding 26
It surrounds the coil block. In FIGS. 2 and 3, the upper and lower parts of the second saturable core 52 are respectively provided with fixtures 72.
74 constitutes an auxiliary magnetic flux loop path together with the center leg 54, and functions as a path for the magnetic flux of the control winding 26 when the control winding 26 is supplied with a DC excitation current. That is, the magnetic flux of the control winding 26 partially magnetically saturates the second core portion 54b of the center leg 54, thereby causing the first core portion 54b to become partially magnetically saturated.
The magnetic flux of the series winding 46 and the shunt winding 48 is routed from the main flux loop through the magnetic shunt core 44 to the outer leg 56,
Shift to 58.

The first series winding 46 and the shunt winding 48 are connected to the center leg 5.
The second series winding 50 is wound on the second core 54b with the opposite polarity to the first series winding 46 to form a so-called autotransformer. Differentially coupled.

In the above configuration, when the input terminals 14 and 16 are connected to the AC power supply 12 and the output terminals 20 and 22 are connected to the inductive load 18, a large current flows through the first and second series windings. A difference current between the input current and the output current flows through the first shunt winding 48 .

1 and 2, when the DC excitation current is not supplied to the controller $9126, the first series winding 46 and the shunt winding 4
8 and the second series winding 5o differentially coupled to the shunt winding 48 passes from the center leg 54 through the outer legs 56, 58 and circulates back to the center leg 54.

At this time, since the magnetic flux generated by the first series winding 46 and the shunt winding 48 and the magnetic flux generated by the second series winding 50 are in opposite directions, the mutual magnetic flux as a whole acts as a difference. . Therefore, the output voltage at this time is the minimum.

Next, when the DC excitation current is supplied to the control winding 26,
Since the second saturable iron core 52 is magnetically saturated together with the second and third core portions 54b and 54c of the center leg 54,
The magnetic flux produced by first series winding 46 and shunt winding 48 is shifted to magnetic shunt core 44 . At this time, the magnetic flux circulates through the first core portion 54a, the outer legs 56, 58, and the magnetic shunt core 44, and the output voltage at the output ends 20, 22 becomes maximum. When the DC excitation current supplied to the control winding 26 is reduced, the output voltage at the output end of the output winding is reduced accordingly. Thus, the second . By variably controlling the magnetic saturation state of the third core portions 54b and 54C, the differential coupling state of the second series winding 5o to the output winding consisting of the first series winding 46 and the shunt winding 48 is changed. The magnetic fluxes of the first series winding 46 and the shunt winding 48 shifted to the magnetic shunt core 44 can be controlled, and the output voltage at the output end can be variably controlled.

Returning to FIG. 1, the DC excitation power supply 28 includes a current transformer 80 connected to the output side of the autotransformer thermal type voltage regulator 24 and an AC reactor 82 connected via a transformer 81. The current transformer 80 functions as a current component circuit for extracting a component dependent on the current of the inductive load 18. The AC reactor 82 functions as a voltage component circuit for extracting a component dependent on the output voltage of the voltage regulator 24 from the voltage transformed from high voltage to low voltage via the transformer 81. Both components are the AC input of the rectifier 84. Vector composition is performed on the side. The DC output current of the rectifier 84 corresponds to a rectified composite current of open components, is smoothed by a capacitor 86, and is used as the DC excitation current I of the control winding 26. Due to the current-dependent component and voltage-dependent component included in the DC output current of the rectifier 84, it is possible to compensate for load fluctuations with a high-speed response by changing the DC output current when the load is turned on or off, or when the load suddenly fluctuates. .

The semiconductor switch circuit 30 includes a semiconductor switch 88,
This semiconductor switch 88 is connected between the DC output terminals of the rectifier 84 to control the DC exciting current. As the semiconductor switch 88, a transistor or a thyristor can be used.

In FIG. 1, semiconductor switch 88 includes first and second control transistors 88a and 88b forming an inverted Darlington circuit.

Here, the inverted Darlington circuit is a PNP
In other words, in order to control the base current of the first control transistor 88a, the second control transistor 88b is connected in an inverted Darlington manner. It forms the Doderlington circuit. A current absorption circuit 90 is connected in parallel to the control winding 26 that is supplied with a DC excitation current. 1! ! A capacitor is used as the current absorption circuit 90. This current absorption circuit 90
acts to absorb the difference current between the DC output current of the rectifier 84 and the DC excitation current when the semiconductor switch 88 is off. A voltage limiting element 92 is connected in parallel with the current absorbing circuit 88 . This voltage control element 92 becomes conductive when the excitation voltage reaches the voltage limited by the voltage limiting element 92, and is provided to prevent overvoltage from being applied to the semiconductor switch 88 and the current absorption circuit 90. An embodiment in which a constant voltage diode is used as the voltage limiting element 92 is shown in FIG. In FIG. 1, a backflow prevention diode 94 is inserted between a capacitor serving as a current absorption circuit 90 and a semiconductor switch 88. Diode 94 prevents discharge current from capacitor 90 from flowing through semiconductor switch 88 when semiconductor switch 88 is on. This prevents the risk of destruction of elements such as the illustrated transistor used as the semiconductor switch 88.

6 and 7, in the load detection circuit 32, a sine wave voltage signal (a) from a transformer (not shown) is supplied to an amplifier 11100 consisting of an operational amplifier, and similarly a current transformer (not shown) The sinusoidal current signal (b) from ) is supplied to an amplifier 102 which also comprises an operational amplifier. Amplifiers 100 and 102 have large amplification factors, convert signals (a) and (b) into rectangular waves, respectively, and output signals ((1) and (d). Then, signals (c) and (d) are output. d
) is supplied to the NOR circuit 104, and the signals (c) and (
A pulse (e) equal to the phase difference (0) of d) is output.

This pulse (e) is converted into a DC signal (f) via a low-pass filter 106 consisting of a resistor and a capacitor. This DC signal (f) is supplied to the control circuit 34.

In FIG. 1, the control circuit 34 includes a transistor 108,
The output signal of the triangular wave oscillator 110 and the load detection circuit 32 (
f) and the triangular waveform output g of the triangular wave oscillator 110, and outputs drive pulses with different pulse widths to the base of the transistor 108. The collector of the transistor 108 is connected to the collector side of the transistor 88a via resistors R1 and R2, and the conduction rate of the semiconductor switch 80 is controlled by turning on/off the transistor 88b. This adjusts the excitation current of the control winding 26. In this case, the conduction rate is controlled by the control circuit 34 so as to eliminate the phase difference between the load voltage and the load current.

Next, the operation will be explained with reference to examples of voltage and current waveforms of each part shown in FIG.

The circuit constants between the DC output terminals of the rectifier 84 are selected in such a way that they are greater than the required value of the excitation current I' of the control winding 26 in any case. When the semiconductor switch 88 is on, the DC output current 2 of the rectifier 84 is shunted by the semiconductor switch 88, and the exciting current ′ decreases.
Next, when the semiconductor switch 88 is turned off, the rectifier output current flows into the control winding 26 from an increasing point.

Due to the inductance of the control winding 26, the number of excitation currents is
can only increase gradually, so the difference current I-I'i±
The current flows into the current absorbing capacitor 90. In this way,
The excitation electric current fiI't± is controlled by the instantaneous value of 5LZ to be maintained at the target value by the base signal of the semiconductor switch 88.

Time 0, when the output signal f proportional to the phase difference between the load voltage and load current applied to the negative input terminal of the amplifier 112 is compared with the triangular waveform signal g applied to the positive input terminal, and an output pulse h is generated. At t1, the amplifier 1121 or 1
``signal is output, and at one time t2, an rzo'' signal is output. When a "1" signal is output from the amplifier 112, the transistor 108 is turned on.

Transistors 88a and 88b are turned on.

The conduction rate α of the semiconductor switch 88 at a certain moment can be expressed as on α=−−−−−−−, where the on time is Ton and the period is T, and the average value of the exciting current ■′ is 1′ av
L, if the average value of the rectifier output is av, then the relationship is I'av=aIav. That is, when viewed as an average value, it can be seen that out of the rectifier output type i4I, only αIav flows for excitation, and the remaining (1-α)Iav flows in the semiconductor switch B8. In this manner, the semiconductor switch 88 is turned on and off in response to the load condition detected by the load detection circuit 32.

The control circuit 34 controls the phase difference between the load voltage and the load current so that it always approaches zero level. That is, when the phase difference θ between the load voltage and the load current is large, the power factor of the inductive load is extremely low, and the output f of the load detection circuit 32 becomes high. At this time, as is clear from FIG. 8, since the pulse width of the output j of the transistor 108 becomes large, the conduction rate of the semiconductor switch 88 becomes large, and the amount of divided excitation current becomes large. Therefore, the control current I' supplied to the control winding NlA26 decreases, and the second core portion 54b of the center leg S4 of the autotransformer thermal type voltage regulator 24 decreases.
magnetic saturation decreases. At this time, the magnetic fluxes of the first series winding 46 and the shunt winding 48 in FIG. descend. Next, when the inductive load increases and the phase difference between the load voltage and the load current becomes smaller, the output f of the load detection circuit 32 becomes lower. At this time, since the pulse width of the output of the amplifier 112 becomes smaller, the conduction rate of the semiconductor switch 88 becomes smaller, the excitation current 1' increases, and the output voltage of the voltage regulator 24 increases. In this way, the control circuit 34 controls the excitation current generator' by controlling the conduction rate of the semiconductor switch 88 in response to the output signal f of the load detection circuit 32, and also controls the inductive current from the voltage regulator 24. The output voltage supplied to the load 18 is adjusted so that the power factor becomes unity. As a result.

For example, if a 3-phase 200V rated 2.2KW induction motor is driven with an input voltage of 200V in a no-load state, 4.
When a current of 8A flows, the power factor is 0.56° and the power consumption is 537.
．． 6 watts, but if the input voltage is lowered to 50 V so that the power factor is 1, the current consumption will be only 1.158, and the power consumption will be about 1710, or 59.5 watts.

Next, if the load factor of this motor is about 50% and the input voltage is 200V, the current consumption will be 5. OA has a power factor of 0.
85. While the power consumption is 850 watts, if the input voltage is lowered to a level where the power factor is 1, that is, 130 volts, the current consumption becomes 4.88 and the power consumption becomes 624 watts, a reduction of 27.6%. It saves energy.

Although a single-phase embodiment of the present invention has been described above, the magnetic control layer voltage WRmg1 described above can also be connected to a three-phase AC power source by three-phase wiring. The load detection circuit 32 is a phase detection circuit disclosed in, for example, U.S. Pat. No. 3,588,710 and U.S. Pat.
, No. 258 may be used.

[Effects of the Invention] As is clear from the above, the power saving control device according to the present invention brings about the following effects.

(1) The load voltage is instantaneously controlled to the optimum level according to the load condition, that is, the load voltage is reduced to the optimum level in proportion to the decrease in the load factor, so the inductive load is always driven at the highest power factor, resulting in a significant Energy saving effect can be obtained.

(2) The load voltage is controlled by controlling the excitation current flowing through the control winding of the autotransformer thermal type voltage regulator, and there is no direct phase control of the AC voltage in the power supply line, so the load current is caused by harmonics. It does not contain any components and does not distort the AC voltage waveform, so it does not cause any damage to computers or other control devices.

(3) Since the load current does not include harmonic components, a large and expensive high-capacity harmonic filter can be omitted, improving reliability and safety, as well as significantly reducing size and weight.

(4) The semiconductor switch does not directly control the AC voltage of the power line, but rather controls the low voltage and low excitation current of the control winding of the autotransformer lead layer voltage regulator. Significantly smaller capacity and significantly lower costs can be achieved.

Also, circuit design becomes easier.

(5) Power-saving control device with large load capacity is 1/100th
Since it can be controlled by the following self-capacitance single-turn transformer lead layer voltage regulator, the entire device is smaller and lighter, does not generate large electromagnetic noise, and is highly reliable, making it suitable for use in spacecraft such as shuttles. It is possible.

(6) High-voltage, large-capacity voltage control is possible by combining a low-voltage, small-capacity semiconductor switch with the control winding of an autotransformer lead-layer voltage regulator, making it safe, reliable, and easy to use.
This has great practical effects because it is possible to control large amounts of power, which was previously impossible, using extremely inexpensive electronic components.

(7) Enables the adoption of an autotransformer lead-layer voltage regulator with a small self-capacity and a low-power control circuit for a large load capacity, minimizing energy loss, resulting in significantly higher efficiency.

Fig. 1 is a wiring diagram of a preferred embodiment of the power-saving control device according to the present invention, Fig. 2 is a plan view of the autotransformer type voltage regulator M shown in Fig. 1, and Fig. 3 is a voltage regulator shown in Fig. 2. Figure 4 is a bottom view of the voltage regulator in Figure 2, Figure 5 is a side view of the voltage regulator in Figure 2, and Figure 5 is a side view of the voltage regulator in Figure 2.
6 is a circuit diagram showing an example of the load detection circuit of FIG. 1; FIG. 6 is a cross-sectional view taken along line V;

7 shows a waveform diagram of the circuit shown in FIG. 6, and FIG. 8 shows a current voltage waveform diagram of the circuit shown in FIG. 1.

24... Autotransformer lead layer voltage regulator 28.
......DC excitation power supply 3α...Semiconductor switch circuit 32...
...Load detection circuit 34... Control circuit Patent applicant Alex Electronics Co., Ltd. Figure 4) Figure 5 Book Otsu Diagram L Book J Figure 7

Claims (1)

[Claims] 1. (a) An autotransformer comprising an output winding connected between an AC power source and an inductive load, and a control winding for adjusting the output voltage of the output winding. (b) an AC reactor that is connected to the input side of the inductive load and takes out a component that depends on the output voltage; and a current transformer that takes out a component that depends on the current of the inductive load; (c) a DC excitation power supply comprising: a rectifier that rectifies a vector-combined current to supply a DC excitation current to the control winding; (c) a DC excitation power supply connected between the control winding and the DC excitation power supply; a semiconductor switch that controls the DC excitation current supplied to the control winding; (d) a load detection circuit that generates an output signal corresponding to the load state of the inductive load; , a control circuit for controlling the conduction rate of the semiconductor switch, and a power saving control device for an inductive load. 2. The inductive load according to claim 1, wherein the semiconductor switch is connected to a DC output terminal of the DC excitation power source, and a part of the DC excitation current is shunted to the semiconductor switch. Power saving control device for use. 3. The power saving control device for an inductive load according to claim 2, characterized in that a current absorption circuit is connected in parallel to the semiconductor switch. 4. The power saving control device for an inductive load according to claim 3, wherein a voltage limiting element is connected in parallel to the semiconductor switch. 5. The power-saving control device for an inductive load according to claim 1 or 2, wherein the load detection circuit includes a phase difference detection circuit that detects a phase difference between a load voltage and a load current. . 6. The control circuit includes an amplifier that generates an output pulse with a pulse width in response to the output signal, and a transistor that controls the conduction rate of the semiconductor switch in response to the output of the amplifier. A power saving control device for an inductive load according to claim 1 or 2. 7. The autotransformer type voltage regulator includes a first series winding, a shunt winding connected in series to the first series winding, and a shunt winding having a polarity different from that of the shunt winding. a main flux loop path comprising a first saturable iron core having a second series winding connected in series with the line; and a main flux loop path disposed between the shunt winding and the second series winding to at least one magnetic shunt core with an air gap for bypassing a portion of the loop path; and a portion of the first saturable core between the magnetic shunt core and the second series winding. and an auxiliary magnetic flux loop path consisting of a second saturable core magnetically coupled to the control winding, the control winding magnetically saturating the part of the first saturable core via the second saturable core. 3. The power saving control device for an inductive load according to claim 1, wherein the magnetic flux of the first series winding and the shunt winding is shifted to the magnetic shunt core. 8. The first saturable core comprises a first-turn core having a center leg and an outer leg, and the auxiliary flux loop path surrounds the second series winding and the control winding on the center leg. a second-volume core arranged as shown in FIG. A power saving control device for an inductive load according to claim 7. 9. Further comprising a first fixture for fixing the first volume core and the center leg, and a second fixture for fixing the second volume core and the center leg. A power saving control device for an inductive load according to claim 8. 10. Claims 8 to 9, characterized in that the first volume core is arranged on one side of the center leg, and the second volume core is arranged on the other side of the center leg. 10. The power saving control device for inductive load according to claim 9. 11. Claim 11, characterized in that the first series winding, the shunt winding, the second series winding, and the control winding are arranged in substantially the same plane. The described power saving control device for inductive loads.