JPH0213980B2 - - Google Patents

Description

[Detailed description of the invention]

In a PM transmission system, the present invention provides communication confidentiality against interception or leakage by inverting the frequency of the transmitted signal on the PM transmission path (wireless section, etc.) with respect to the spectrum of the input signal of the system. and does not perform spectrum inversion to improve speech quality due to fading noise generated in the PM transmission line system.
This relates to a spectrum inversion communication system that allows transmission to be maintained at the same level as the PM transmission system. Conventional systems of this type consist of a group of functions shown in Figure 1, and each function generates a signal with a spectrum as shown in Figure 2, which is then transmitted as a PM, which reduces the fading noise suppression effect. The first drawback is that it is not very promising, and the second drawback is that different processing is required on the transmitting side and the receiving side, making it difficult to expect the effects of mass production of devices. The drawings will be explained in detail below. In FIG. 1, 1 is a spectrum inversion circuit;
2 is a pre-emphasis circuit, 3 is a PM transmission circuit,
4 is a transmitting antenna, 4' is a receiving antenna, 5 is a
PM receiving circuit, 6 is due diligence circuit, 7 is
The PM transmission path formed by the PM transmitting/receiving circuit, the antenna, and the propagation space, and points a to f are observation points. Second
The diagram shows the signal spectrum at each observation point in FIG. 1, with the horizontal axis representing frequency and the vertical axis representing the absolute value of amplitude, both on a logarithmic scale. Second
In the figure, f ₁ is the lower limit frequency of the band, f ₂ is the upper limit frequency of the band, the area above the horizontal axis represents the signal component, and the shaded area below represents fading noise, which doubles the f ₁ component of the original signal. Represented by a line. Figure 3 shows the power spectrum of the male voice, and Figure 4 the statistically averaged power spectrum of the female voice spectrum. As can be seen from both figures, the band [0.3, 3] (f ₁ = 0.3, f ₂
= 3kHz), the spectrum envelope can be approximated as f ^- 〓(α1), as shown by the dotted line. FIG. 5 shows the statistical average of the power spectrum of the reception noise in the PM transmission system, and it can be observed that the reception noise always has a downward-sloping envelope to the right, regardless of the reception level. This is because fading noise, which is the dominant term in reception noise, is impulsive noise and can be considered white noise, but during PM demodulation, it is configured with a function that integrates FM demodulation, so white noise is integrated and becomes downward slope to the right. This can be easily explained from this point of view. In other words, it can be inferred from FIG. 5 that the power spectrum of the reception noise in the FM transmission line has a flat characteristic. Next, the operation will be explained in order. The input terminal 8 has the spectrum shown in Figure 2 a.
Input the original signal G(f) of the form f ^- 〓(α1). Spectrum inversion circuit 1 inverts the spectrum and
The output G _I (f) shown in Figure b is found. G _I (f) = S [G (f)] = G (f ₀ - f), where f ₀ = f ₁ + f ₂ ... (1) (= 3.3kHz) S [*] has a spectrum inversion function. represent Next, the pre-emphasis circuit 2 performs pre-emphasis to obtain the input signal G _T (f) to the PM transmitting circuit 3. G _T (f)=H _p (f)・G _I (f) ……(2) Here, H _p (f) is the pre-emphasis characteristic,
In the conventional method, most of the characteristics had an integral characteristic or had a characteristic with a gentler slope than an integral characteristic. For example, as an example of H _p (f), H _p (f)=1+(f ₀ -f) ² /f _C ² /1+f ² /f ² / _C , f _C =0.
8kHz...(3) is given. Its characteristics are shown in FIG. The input signal G _I (f) of the pre-emphasis filter is
The spectrum is converted by pre-emphasis into the spectrum shown in FIG. 2c. When G _T (f) is transmitted through the PM transmission path, inverted triangular reception noise is mixed in at observation point d, as shown in Fig. 2d. Let G _R (f) be the signal at observation point d. G _R (f) is de-emphasized by a de-emphasis circuit 6. G _D the output of de-emphasis circuit 6
(f), G _D (f) is given as follows. G _D (f)=H _d (f) G _R (f) ... (4) G _D (f) is then spectrum inverted to obtain the reproduced signal G _O (f). G _O (f)=S [G _D (f)] = H _d (f _{0 −} f) G _R (f ₀ − f) ……(5) Here H _d (f)=1+f ² /f _C ² /1 + (f ₀ - f) ² /f _C ² If the PM transmission path is distortion-free and noise-free, then G _R (f)≡G _T (f)
It is. Therefore, the reproduced signal G _O (f) is G _O (f)=H _d (f ₀ − f) G _T (f _{0 −} f)=H _d (f ₀ − f) H _p (f
₀ − f) G _I (f ₀ − f) = H _d (f ₀ − f) H _p (f ₀ − f) G(f) ……(6). If the condition H _p (f)・H _d (f)=1 is satisfied, then
G _O (f)=G(f), and it is known that the original signal is not distorted by pre-emphasis and de-emphasis. just
Since the received noise mixed in the PM transmission path is converted into a substantially flat signal by de-emphasis, the noise spectrum after spectrum inversion also becomes flat. This relationship is shown in Figure 2 e and f. As a second example of this type of conventional method, shown in Figure 7.
A principle inversion method consisting only of a PM transmission path and a spectrum inversion function is shown. In Figure 7, 1, 3, 4, 4', 5 are the first
These are the same spectrum inversion circuit, PM transmission line, transmitting antenna, receiving antenna, and PM receiving circuit as shown in the figure, g, h, i, and j are observation points, and 7 is the PM transmission line. If the original signal G(f) is input to the input terminal 8 in FIG. 7, the spectrum at observation points g and h will have the same spectrum envelope as in FIG. 2. When G _I (f) is directly transmitted as a PM, noise having an inverted triangular spectrum envelope is mixed in as shown in FIG. 8i.
When this is reversed, it can be seen that noise having a triangular envelope shown in FIG. 8j is mixed into the reproduced signal. Here, the inverted triangular shape mixed in when receiving PM, FM
We quantitatively determine the degree of interference caused to the signal by flat noise mixed in during reception and triangular noise mixed in by principle inversion. The flat shape during FM reception is the conventional H _p (f),
The shape is very similar to the noise of the emphasis inversion method using H _d (f). These inverted triangular shapes will be referred to as type A, and the flat spectra will be referred to as type B.
Furthermore, the triangular shape is called C shape. Many people are studying methods for quantitatively evaluating noise that forms a spectrum. Here, the input signal is frequency-divided into critical bandwidths faithfully based on the perceptual model shown in Figure 9, and the influence of masking is corrected by taking into account the amount of masking and audiogram due to neighboring band components. Determine the total size by finding the sum of the sizes of E.
Use Zwicker's method. E. Zwicker's method is edited by the Acoustical Society of Japan, published by Corona Publishing, Acoustic Engineering Course 6 "Auditory and Psychoacoustics" (written by Hisao Sakai and Tsuyoshi Nakayama), Chapter 11, 11, 6.
1, but here we will briefly describe the calculation process using the same means as in the above document. FIG. 9 is a model of the auditory organ. In the figure, 10 indicates parts corresponding to the outer ear and middle ear of the human ear, 11 is a critical band filter in the auditory model, 12 is an effective value detector, and 13 is an auditory organ. The upper integrator, 14 is a loudness level correction circuit, 15 is a dB-sone converter that converts the loudness amount into a sone value,
16 is a masking effect correction circuit, and 17 is a loudness adder for corrected sone values in each critical band. Equivalent sound pressure is calculated using E. Zwicker's method. In Table 1, the auditory sound pressure in the case of having an inverted triangular integral characteristic is determined. First of all, band [0.3, 3]
Divide kHz into 13 critical bands No. 4 to No. 16. Next, the band level L _G is determined for each band.

[Table] 2

【table】

[Table] 2

[Table] 2
Since type A has an integral characteristic, assuming that the sound pressure per 1 Hz of 1 kHz is 40 dB, the band level L _G in each critical band is obtained by approximating the value at the center frequency f _G. Therefore L _G is 40+20log ₁₀ Δf _G −
Given as 20log ₁₀ f _G. The sound loudness level L _1kHz can be found by adding the loudness correction amount to L _G. E. Zwicker's correction amount is as shown in the table. If we find the sound loudness level L _1kHz , we can find the sound loudness N _G [sone] from this using Stevens' power law log ₁₀ N _G =0.03L _1kHz −1.2 ...(7). E. Zwicker has shown that the masking characteristics due to noise per critical bandwidth are given as shown in FIG. This masking is done by E.
As shown in Figure 11, Zwicker uses an excitatory pattern model of sensory organs, assuming that the perceptual amount is proportional to the area of the excitation pattern, and that excitation patterns with adjacent critical bandwidths overlap, but the overlapping part is not perceptual. It was assumed that the envelope area of the pattern was not involved. In other words, it is explained that the overlapping area represents the amount of masking of noise forming a spectrum. The area S _G of A in FIG. 11a is defined as follows. S _G =2N _G lo ² ...(8) where N _G is the noise level of the critical bandwidth, and lo ² is the unit area of the excitation pattern. From E. Zwicker's experiments, the spread of the lower base corresponding to the upper base of the trapezoid of the pattern becomes wider as the sound volume level (L _1kHz ) increases, and the characteristics shown in Figure 12 are required. The unit n on the vertical axis in FIG. 12 represents the spread of the lower base, and in the range of L _1kHz from 50 to 75 [phon], n=3. Therefore, the height h of the excitation pattern is obtained from the relational expression S _G =n+1/2hlo≡2N _G lo ² ...(9), h=4/n+1N _G lo=N _G lo｜ _o=3 ... (10) is known to be proportional to N _G. Therefore, 1/2 of the sum of the sound magnitudes N _G in each critical band becomes approximately the perceptual sound ratio level [dB]. At this time, if the protrusion of the pattern at the edge of the band is corrected, S _G can be determined correctly. In other words, the magnitude of the noise forming the spectrum N〓
is N〓=1/2 ₁₆ 〓 ⁱ⁼⁴ N _Gi +1/4 (N _G4 +N _G16 ) [dB] ...(11) However, α is the spectrum pattern name A, B, C
, and N _Gi is determined as the sound level of the critical band Noi. A-type noise N _A is as shown in Table 1.
It is determined by 30.1 [dB]. Next, the auditory magnitude of the B-type spectrum noise is determined. In the case of type A, its normalized power is ∫ ^f2 _f1 1/f ² df=1/f ₁ -1/f ₂ , f ₁ =0.3, f ₂ =3
.0kHz ……(12) is given. On the other hand, in the case of type B, since it has a flat characteristic, the power is ∫ ^f2 _f1 kdf=k(f ₂ − f ₁ ) (13). If the noise powers are not equal, the auditory loudness of the sound cannot be compared correctly. Therefore, the conditions for matching the powers of the flat B type and the integral A type are found from the value of k that makes equations (12) and (13) equal. That is, from the condition k(f ₂ −f ₁ )=f ₂ −f ₁ /f ₁ f ₂ , k=1/f ₁ f ₂ =1/0.9 (=0.5 dB) (14) is obtained. This is the sound pressure per 1Hz for type B.
It is necessary to increase it by 0.5dB. Therefore, as shown in Table 2, L _G =40+20log ₁₀ Δf _G +0.5 [dB] ...(15) Calculation of L _1kHz and _NG is the same as in Table 1. If we add the band edge correction amount (1/4 N _G4 + 1/4 N _G16 ) to 1/2 of the total sum of N _G , we can immediately find N _B [dB] as N _B =33.9 [dB] ... (16) . The noise magnitude N _C of the C-type differential characteristic is determined as shown in Table 3. The differential characteristic is the inversion of the integral, and when variable transformation x=f ₀ -f is applied, dx=-df, so equation (12) can be transformed as follows. ∫ ^f ₀ ^-f2 _f0-f1 1/(f ₀ -x) ² (-dx), where f ₀ = f ₁ + f
₂ =∫ ^f1 _f2 1/(f ₀ −x) ² (−dx)=∫ ^f2 _f1 1/(f ₀
-x) ² dx ...(17) Therefore, since the powers of the A-type and C-type spectra are equal, the L _G of the A-type in Table 3 is L _G =40+20log ₁₀ Δf _G -20log ₁₀ f L of the B-type _G is L _G = 40 + 20log ₁₀ Δf _G −20log ₁₀ (f ₀ − f)
...(18) is given. L _1kHz , N _G , and N _C can be calculated in the same way, and N _C =40.6 [dB] ... (19) is found. Since N _A , N _B , and N _C are the values that express the noise forming the spectrum in terms of the sound pressure [dB] of a 1kHz pure tone, the degree of interference is relative to the noise forming the A, B, and C patterns. The interference degree for each type A pattern I _A = 0 dB Type B pattern 〃 I _B = 3.8 dB Type C pattern 〃 I _C = 10.5 dB
...(20) is found. As already mentioned, the A, B, and C type noises refer to those whose envelope patterns are in the shape of an inverted triangle, flat, or triangular, as shown in a, b, and c of FIG. As explained earlier, the noise of the conventional emphasis inversion method shown in Figure 1 is type B, and the noise of the conventional principle inversion method shown in Figure 7 is type C, so compared to type A. This has the disadvantage that the degree of noise interference with the signal, in other words, the SN value deteriorates. In order to eliminate these drawbacks, the present invention provides a transmitting and receiving isomorphic emphasis characteristic H(f )
It uses the same emphasis on the transmitter and receiver sides, improves the effectiveness of mass production of devices, and improves economic efficiency.
The drawings will be explained in detail below. FIG. 14 shows an example of the configuration of a wireless communication system according to the present invention. In FIG. 14, 18 is the emphasis characteristic H
(f), 1 is a spectrum inversion circuit, 3 is a PM transmitting circuit, 4 is a transmitting antenna, 4'
is the receiving antenna, 5 is the PM receiving circuit, k, l,
m, n, o, p are observation points. FIG. 15 shows the spectrum of the signal at the observation points shown in FIG. 14 corresponding to each observation point. It will be shown that audio processing on the transmitting side and the receiving side can be realized by a group of identical functions in which the filter 18 and the spectrum inversion circuit 1 are connected in cascade. First, the emphasis on the sending side is H _T (f), and the emphasis on the receiving side is H _T (f).
Now let us compare the characteristics of H _T (f) and H _R (f). The original signal G(f) is input to the input terminal 8. The output G _E (f) of the filter 18 having the emphasis H _R (f) is G _E (f)=H _T (f)G(f) (21). Then, the spectrum is inverted, and the input signal G _T (f) to the transmitting circuit is determined. G _T (f) = S [G _E (f)] = H _T (f ₀ - f) G (f ₀ - f)
...(22) Assuming that the PM transmission path is distortion-free, the demodulated signal
G _R (f) matches G _T (f). Emphasis on the receiving side
After H _R (f) processing, the filter output G _D (f) is G _D (f)=H _R (f)G _R (f) = H _R (f)H _T (f ₀ −f)G( f ₀ − f) ...(23) The output of the receiving side emphasis filter 18 is spectrally inverted to produce a reproduced signal G _O (f). G _O (f)=S [G _D (f)] = H _R (f ₀ − f) H _T (f) G(f) ……(24) Therefore, H _R (f ₀ − f) H It is known that _T (f)=1 is a necessary and sufficient condition for the system to be distortion-free. Next, determine H _R (f) and H _T (f) specifically. The fact that the transmission modulation degree when applying the present invention is the same as that of the PM transmission system means that base stations and relay stations existing in the wireless section simply relay the voice without processing it, or connect to the telephone network. In doing so,
There is no need to change the equipment and it can be introduced economically. The conditions for keeping the modulation depth equal in PM modulation are given as follows. ∫ ^f2 _f1 f ² G ² (f)df＝∫ ^f2 _f1 f ² {S〔H _T (f)
G(f)]} ² df (25) The left side of equation (25) represents the modulation degree of the PM transmission system, and the right side represents the modulation degree in the system of the present invention. After expanding the right-hand side, if we apply variable transformation x=f ₀ −f, we get
Since dx = −df, the right side of equation (25) = ∫ ^f2 _f1 f ² H _T ² (f ₀ − f) G ² (f ₀
−f) df =∫ ^f ₀ ^-f2 _f0-f1 (f−x) ² H _T ² (x) G ² (x)
(−dx)=∫ ^f2 _f1 (f ₀ −x) ² H _T ² (x) G ² (x) dx……
(26) The condition that the left sides of equations (26) and (25) are equal is satisfied when the integrands are equal. Therefore, f ² G ² (f)=(f ₀ −f) ² H _T ² (f)G ² (f) ……(27) That is, H _T (f)=f(f ₀ −f) ⁻¹ ...(28) and H _T (f) are determined. By substituting equation (28) into the distortion-free transmission condition of the system, H _R (f) can be found. Restating the distortion-free transmission conditions, H _R (f ₀ −f) H _T (f)=1 (29). If we perform variable transformation on both sides of equation (29), we get
From the linearity of H _R and H _T , the conditions can be modified as follows. H _R (f)H _T (f ₀ −f)=1 (29)′ From equation (28), H _T (f ₀ −f) is given as follows. H _T (f ₀ - f) = (f ₀ - f) f ^-1 ... (30) Therefore, H _R (f) = 1/H _T (f ₀ - f) = f (f ₀ - f) ^-1 ……(31) is found. If we compare equations (28) and (31), it becomes immediately clear that H _T (f)≡H _R (f). Therefore, from now on, H _R (f) and H _T (f) will be simply written as H(f). Next, we will examine the characteristics of the system when the emphasis function H(f) is extended and made more general. That is, the generalized emphasis function is defined as follows. H(f)=f〓(f ₀ −f) ^- 〓, α1 ...(32) The transmission characteristics of the system when using such an emphasis function can be found in the same way as described above, and the transmission emphasis H′(f) is Between reception emphasis H″(f) H′(f ₀ − f) H″(f)=H′(f)H″(f ₀ − f)=1
If (33) holds true, there is no distortion. The fact H′(f ₀ −f)=H(f ₀ −f)=(f ₀ −f)〓f ⁻ 〓 H″(f)=H(f)=f〓(f ₀ −f) ⁻ 〓 and Given, the product of both sides H′(f ₀ − f)・H″(f)
It is known that the value becomes 1 and distortion-free transmission characteristics can be obtained. On the other hand, as can be seen from the envelope of the voice spectrum story shown in Figures 3 and 4, the statistical average G^ of G(f)
(f) can be approximated as G^(f)=f ^- 〓. The statistical average amount of the modulation degree according to the present invention is given by the following equation. ∫ ^f2 _f1 {S[H(f)G^(f)]} ² df...(34) Expand equation (34). Equation (34) = ∫ ^f2 _f1 f ² H ² (f ₀ − f) G^ ² (f ₀ − f) d
f＝∫ ^f2 _f1 f ² {(f ₀ − f)〓f ⁻ 〓} ² {(f ₀ − f) ⁻ 〓} ²
df =∫ ^f2 _f1 f ² f ^-2 〓df=∫ ^f2 _f1 f ² G^ ² (f)df……(3
5) The final right-hand side of equation (35) is nothing but the statistical average value of the modulation degree of the PM transmission system, and it is also known that the modulation degree is approximately preserved in the sense of a statistical average. As described above, it has been shown that the isomorphic emphasis reversal secret message transmission system performs isomorphic processing for send and receive, that the system can provide a secret function on the transmission path, and that it is distortion-free. The transmitting/receiving isomorphic emphasis function H(f) is H(f)=f〓(f ₀ − f) ^- 〓, where f ₀ = f ₁ + f ₂
,
α1, which is a realizable filter characteristic. In fact, since H(f) has almost a double differential characteristic, circuitization of H(f) can be easily inferred. On the other hand, the spectrum inversion function uses, for example, the frequency f ₀ = f ₁ + f ₂
This can be achieved by using a pure tone as a local signal, generating a modulation signal with the input signal using a modulator such as a mixer, and extracting only the lower sideband. Next, the noise characteristics according to the method of the present invention will be explained. FIG. 17 shows an equivalent circuit representation of the emphasis inversion shown in FIG. 16. FIG. 16 shows the audio processing section shown in FIG. 14, in which 19 is an emphasis circuit and 1 is a spectrum inversion circuit.
20 in Figure 17 is a filter with H ₁ (f) characteristic, 21
Let be a filter with H ₂ (f) characteristic, and let 1 be a spectrum inversion circuit. The condition that Figures 16 and 17 are equivalent is that the same output signal can be obtained for the same input signal G _R (f), and it is necessary that G _O (f)≡G _O ~ (f). This is a sufficient condition. G _O (f)=S[H(f)G _R (f)] =H(f ₀ −f)G _R (f ₀ −f) ……(36) G _O 〜(f)=H ₂ (f )S[H ₁ (f)G _R (f)] = H ₁ (f ₀ − f) H ₂ (f)G _R (f ₀ − f) ... (37) Therefore, H ₁ (f ₀ − f) H ₂ (f)≡H(f ₀ −f) is found. Alternatively, by finding the spectrum inversion form on both sides, the following formula: H ₁ (f)H ₂ (f ₀ − f)≡H(f)=f〓(f ₀ −f) ^- 〓
...(38) is obtained. By comparing both sides of equation (38), it is immediately determined that H ₁ (f)=f〓 H ₂ (f)=f ^- 〓, α1...(39). The function of FIG. 17 is now completely determined. In other words, if α=1, the filter 20
is a differential characteristic, and the filter 21 is an integral characteristic. This shows that the noise of the system of the present invention almost completely matches the noise of the PM transmission system. The noise in the PM transmission system has approximately integral characteristics as shown in FIG. When this is first differentiated, the noise has flat characteristics. When subsequently inverted, the envelope of the noise spectrum has a flat shape equal to the input. Finally, if we integrate a flat spectrum, it is clear that noise also has integration characteristics. In this way, the emphasis of the present invention H(f)=f〓(f ₀ −
f) ^- It is known that by using the characteristic, noise equivalent to that of the PM transmission system is mixed into the reproduced sound. As explained above, by differentiating, inverting, and integrating the audio on the transmitting and receiving sides, the frequency components can be inverted while keeping the spectral envelope of the signal on the transmission path equal to the envelope of the original signal. The first advantage is that it is possible to determine whether or not spectrum-inverted secret speech is being used from the spectrum envelope, and the speech confidentiality is higher than that of conventional methods. It is held equal to that, and compared to the conventional principle reversal secret story
The second advantage is that the SN value is 10.5 dB, and the SN value is 3.8 dB better than the conventional pre-emphasis/de-emphasis inversion method. The third advantage that can be expected to be improved is that when the present invention is applied to the press-talk transmission system, the fourth advantage is that audio processing only requires one circuit. 5. It is not necessary to change the system, and the secret code can be introduced economically by decoding the secret code only at the transmitting/receiving radio station.
There are advantages.

[Brief explanation of drawings]

Figure 1 shows an example of the configuration of conventional emphasis spectrum inversion, Figure 2 a-f shows the signal spectrum at the observation points in Figure 1, Figure 3 shows the statistically averaged male voice spectrum, and Figure 4 shows the signal spectrum at the observation points in Figure 1. Figure 5 is the statistical average of the noise spectrum of the PM transmission system, Figure 6 is an example of the pre-emphasis and de-emphasis characteristics of the conventional method, and Figure 7 is the configuration of the principle inverted secret speech method of the conventional method. For example, Fig. 8 g to j are the spectra of each signal at the observation points in Fig. 7, Fig. 9 is a model of the auditory organ used by E. Zwicker, Fig. 10 is a masking audiograph using critical band noise, Figure 11a
is a model of the excitation pattern (A), Fig. 11b is E. Zwicker's quantitative ratio model of masking (B), Fig. 12 is a graph showing the relationship between the sound pressure in the critical band and the base of the excitation pattern, and Fig. 13 The figure shows a typical pattern of transmission noise mixed into the reproduced signal, Figure 14 shows a configuration example of the present invention, Figure 15 k to p show each signal spectrum at the observation point in Figure 14, and Figure 16 shows the invention of the present invention. 17 is an equivalent circuit of FIG. 16. 1... Spectrum inversion circuit, 2... Pre-emphasis circuit, 3... PM transmitting circuit, 4... Transmitting antenna, 4'... Receiving antenna, 5... PM receiving circuit, 6... De-emphasis circuit, 7... PM
Transmission line, 8...Input terminal, 9...Output terminal, 10
...Outer ear/middle ear, 11...Critical band filter, 1
2... Effective value detector, 13... Integrator, 14...
Loudness level correction circuit, 15...dB-sone
Converter, 16... Masking correction circuit, 17...
Loudness adder, 18...Emphasis characteristic H
(f), 19... emphasis circuit, 20... filter having the characteristic H ₁ (f),
21...A filter having the characteristic H ₂ (f).

[Claims]

1. A digital signal relay device having a receiver that receives a carrier wave modulated by a digital signal, and a transmitter that modulates the carrier wave again using the digital signal demodulated by the receiver and transmits the modulated carrier wave. a first determining means for determining whether or not a signal is being received by detecting the intermediate frequency level of the receiver; and extracting a recovered clock frequency component obtained from the demodulated output of the receiver in synchronization with the digital signal. a second determining means that determines whether or not a signal is being received by level detection; and a second determining means that determines whether or not a signal is being received; A logic means for outputting a level, a time constant circuit for eliminating discontinuous responses outputted from the logic means, and a comparator for comparing the output of the time constant circuit with a reference value are provided. A transmission control method characterized by controlling the on/off of the transmitter's radio waves. 2. The transmission control system according to claim 1, wherein the time constant circuit has a quick response to the start of signal reception and a slow response to the end of signal reception. 3. Further, a third determining means is provided which takes a longer time than the second determining means to determine whether or not the recovered clock is at a sufficient level, and the output of the third determining means is used to determine whether the regenerated clock is at a sufficient level. The transmission control method according to claim 1 or 2, characterized in that the level of the reference value of is changed.