JPH02223384A - Resonant inverter - Google Patents

Abstract

PURPOSE:To prevent noise even upon variation of switching frequency due to variation of load by setting the switching frequency higher than the resonance frequency of resonance circuit. CONSTITUTION:During first half interval of switching period where a switching element Q1 is turned ON, DC current flows from a power source 1 through the element Q1 to a capacitor C6 and stored therein. During second half interval of the switching period, a switching element Q2 is turned ON to form a DC resonance circuit, and charges stored in the capacitor C6 flow alternately through a coil L1 and the capacitor C6 thus realizing resonance. Switching frequency is previously set higher than the resonance frequency. When the load is reduced, switching frequency increases to prevent occurrence of noise. Since the switching frequency is previously set not to drop below the resonance frequency, noise is not produced even under overload state.

Description

[Detailed Description of the Invention] (Industrial Application Field) The present invention relates to a resonant inverter that switches direct current, passes it through a resonant circuit, and then converts it to direct current and supplies it to a load, and is particularly applicable to load fluctuations. This invention relates to controlling the switching frequency to be changed accordingly.

[Conventional technology]

Conventionally, as this type of resonant inverter, one shown in FIG. 8 is known.

That is, in FIG. 8, the alternating current from the electric current iS1 is rectified by the diode bridge DB and the capacitor C1,
Smoothed and series connected switching cables Q1,Q
It is guided by 2. The switching elements Q1 and Q2 are alternately turned on and off at a switching frequency having a predetermined duty set by the oscillation section 2 and the duty determining section 3.

Capacitor C6 and coil L1 constitute a series resonant circuit,
Furthermore, a switching element Q2 and a primary winding T31 of a transformer T3 are inserted into the circuit.

This resonant circuit is supplied with current (power) from the power supply 1 when the switching element Q1 is turned on, and resonates when the switching element Q2 is turned on.

Due to the above resonance operation, magnetic flux is generated in the primary coil T31,
This magnetic flux causes an alternating current to flow through the secondary side of the transformer T3, which is rectified and smoothed by diodes D6, D7, capacitor C7, and further by choke coil L2 and capacitor C8.
Supplied to load 5.

In addition, the load 5 includes a resistor R8 for detecting the load current.
are connected in series, and a voltage proportional to the load current is fed back to the operational amplifier 6 via a resistor R9.

By the way, FIG. 7 shows the relationship between the on/off frequencies of the switching elements Q1 and Q2 and the resonant gain of the resonant circuit. In the figure, Ga is a characteristic curve representing the gain of the resonant circuit, that is, the amount of energy flowing through the resonant circuit.

Further, fwo is the resonant frequency of the resonant circuit.

The switching frequency is designed to vary within a certain range on the low frequency side outside the resonant frequency fwo region, so that sudden changes in load current do not occur despite slight fluctuations in load. .

That is, for example, when an overload condition occurs and the load current is insufficient and the feedback voltage from the resistor R9 decreases, the operational amplifier 6 increases the current of the phototransistor that constitutes the photocoupler P1, and the oscillation section 2 The oscillation frequency, that is, the switching frequency is increased, and the load current is increased.

On the other hand, when the load becomes lighter and the feedback voltage increases, the operational amplifier 6 suppresses the current of the phototransistor, lowers the switching frequency, and reduces the load current. As a result, an appropriate load current is supplied according to load fluctuations.

[Problem to be solved by the invention]

In the conventional resonant inverter described above, the switching frequency of the switching elements is designed to be lower than the resonant frequency of the resonant circuit, as shown in Figure 7, so load fluctuations, such as load fluctuations, are reduced. However, when the switching frequency decreases accordingly, it enters the audible range and generates noise.

The present invention has been made in view of the above, and provides a resonant inverter that does not generate noise even if the switching frequency changes in accordance with load fluctuations without changing the resonant frequency. The purpose is to

[Means to solve the problem]

In the present invention, a current is supplied during the ON period of one of two switching elements connected in series to a DC power supply, a resonant circuit formed during the ON period of the other resonates, and the output of this resonant circuit is extracted and rectified. In the resonant inverter, the switching frequency of the switching element is controlled to change according to fluctuations in the load, and the switching frequency is higher than the resonant frequency of the resonant circuit. This is what has been set.

[Effect]

According to the resonant inverter according to the present invention, even if the switching frequency of the switching element changes in accordance with changes in the load, the frequency does not drop below the resonant frequency and does not fall into the audible range.

〔Example〕

FIG. 1 is a circuit diagram showing an embodiment of a resonant inverter according to the present invention. In addition, in the figure, the same reference numerals as in FIG. 8 indicate the same parts.

In the figure, 1 is an AC power source such as a commercial power supply, DB is a diode bridge of the above type [1] that rectifies the AC current, and C
1 is a capacitor for smoothing the rectified output of the diode bridge DB, and R1 is a resistor for preventing rush current from flowing into the capacitor C1. Q, 1, and Q2 are switching elements such as MOS transistors, and their switching operations are controlled by an oscillation section 2, a duty determining section 3, and a distribution section 4 shown within the broken lines. That is, the oscillator 2 determines the switching frequencies of the switching elements Ql and Q2,
The distribution section 4 distributes the oscillation output of the oscillation section 2 into switching control pulses of opposite polarity and outputs them, and the duty determining section 3 determines the pulse duty of the switching control pulses.

The oscillation frequency of the oscillation section 2 is determined by a time constant determined by the combined resistance value of the resistor R4 and the conductivity of a phototransistor forming a photocoupler P1, which will be described later, and the capacitance of the capacitor C2.

The pulse duty of the duty determining section 3 is determined by the conductivity of the phototransistor constituting the photocoupler P1. Further, the distribution is made by the switching control pulses of opposite polarity from the section 4 to the switching element Q.
1 and Q2 are alternately turned on and off.

T1 and T2 are transformers that convert the output of the switching control pulses into operating levels of the switching elements Q1 and Q2. Zener diodes D2, D3 and D4, D5 connected in parallel with the secondary sides of the transformers T1, T2 and connected in series with mutually opposite polarities are used to protect the switching elements Q1, Q2 from overvoltage. Also,
Resistor R2, capacitor C4 and resistor R3, capacitor C5 connected to the gates of each switching element Q1, Q2
is an integrating circuit for noise absorption.

Capacitor C6 and coil L1 constitute a series resonant circuit,
Its resonant frequency f w o is f w o = 1 /
2 yr fTgo r[τ−·−(1).

This series resonant circuit also includes a switching cable Q2 and a secondary coil T31 of the transformer T3. are also inserted in series.

The series resonant circuit has switching elements Ql and Q2 switching IJi!, as will be explained below. Series resonance occurs in the latter half of the first period. That is, first, in the first half of the switching period when the switching element Q1 is turned on,
Direct current (power) from the power supply 1 side flows into the capacitor C6 via the switching element Q1 and is accumulated. Next, in the second half of the switching period, the switching cable Q
2 turns on, thereby forming a series resonant circuit, and the charge stored in capacitor C6 is transferred to coil L1, then to capacitor C6, and so on, 1/f w .

It resonates by vibrating and flowing alternately with a period of .

In the above, the switching frequency is the resonant frequency f w
It is set in advance to be higher than o.

Therefore, as shown in Fig. 7, the energy flowing in the series resonant circuit decreases as the switching frequency decreases to the resonant frequency f.
The closer it gets to fwo, the thicker it becomes (and, conversely, the more it rises and the farther away from the resonant frequency fwo, the smaller it becomes.

The transformer T3 separates the power supply side and the load side in terms of direct current, and transmits the energy flowing in the series resonant circuit to the load side. The secondary alternating current generated by the voltage induced in the secondary coils T32 and T33 of the transformer T3 is full-wave rectified by diodes D6 and D7, smoothed by a capacitor C7, and further smoothed by a choke coil L2 and a capacitor C8. and is supplied to the load 5. Note that the diode D8 functions as a flywheel diode that releases the energy accumulated in the choke coil L2 via the load 5.

Further, a circuit including a transistor Q3, a resistor R10, a Zener diode D9, and a capacitor CIO is used to obtain a voltage for supplying a power supply voltage to an operational amplifier 6, which will be described later, to a terminal A.

R8 is a load current detection resistor connected in series with the load 5, and the voltage generated across it is inputted to the non-inverting input terminal of the operational amplifier 6 as a feedback voltage via the resistor R9. C9 is for buffering sudden changes in the feedback voltage.

The operational amplifier 6 is connected to the non-domain anode side with an amplification factor of 1+ (R6/R7).

When the load is reduced and the feedback voltage increases, the operational amplifier 6 increases the output voltage, increases the current flowing through the diode, and lowers the conductivity of the phototransistor. Therefore, the parallel combined resistance value with the resistor R4 decreases, the time constant decreases, and the oscillation frequency by the oscillation section 2, that is, the switching frequency increases.

As a result, the load current supplied to the load 5 decreases as shown in FIG. 7. Conversely, when the load 5 becomes overloaded and the feedback voltage decreases, the current flowing through the au pair is reduced and the conductivity of the phototransistor is increased. Therefore, the parallel combined resistance value with resistor R4 increases, the time constant increases, and the oscillation frequency by the oscillator 2 increases.
In other words, the switching frequency decreases. As a result, the load current supplied to the load 5 increases as shown in FIG. 7.

That is, according to this embodiment, when the load is reduced, the switching frequency increases, so noise is not generated as in the conventional case. Further, even in an overload state, the switching frequency is set in advance so as not to drop below the resonant frequency, so no noise occurs in this case as well.

Next, FIG. 2 is a circuit diagram showing another embodiment, in which the same reference numerals as in FIG. 1 indicate the same components.

This embodiment is a resonant inverter that controls the charging of a storage battery 7, and changes the switching frequency and, in addition, the pulse duty of the switch control pulse, depending on the charging status of the storage battery 7. It controls the current.

In the same figure, a storage battery 7 for charging is connected in place of the load 5 shown in FIG. P2 has been newly added. Note that R13 and R14 are voltage dividing resistors of the transistor Q4.

Now, the power supply current converted to DC is passed through the switching element Q
1, Q2, a series resonant circuit, and a transformer T3, the current is rectified and smoothed, and then supplied to the storage battery 7 as a charging current.

Since the switch SW is in an open state until the storage battery 7 reaches full charge, the transistor 04 and the photocoupler P2 are in an off state. Therefore, in this state, the same operation as the circuit of FIG. 1 is performed. In other words, Mitsuru? l1f
While u is small, the feedback voltage is high and the output voltage of the operational amplifier 6 is high, so the time constant becomes small, the switching frequency increases, and a relatively small current is supplied.

As the amount of charge increases and the voltage between the terminals of the storage battery rises, the voltage of the wave pack decreases and the output voltage of the operational amplifier 6 decreases, so the time constant increases and the switching frequency decreases. , the current gradually increases. Such current control allows sufficient current to be supplied until full charge.

Thereafter, when full charge is detected, the charge control section 8 operates to switch the switch SW to the open state.

By switching this switch SW, trickle charging (by trickle current) after full charging is started, as will be described later.
A fully charged state is maintained by replenishing the self-discharged amount.

When the switch SW is closed, a current flows from the terminal A to the terminal B to the plate I diode constituting the photocoupler P2, which causes the duty determining unit 3 to reduce the pulse duty of the switching elements Ql and Q2. Reduces the energy itself supplied to the series resonant circuit. As a result, compared to controlling only by the switching frequency, the
The current flowing into the secondary battery 7 can be reduced.

Furthermore, when the switch SW is closed, the transistor Q4
is turned on, and the amplification factor of operational amplifier 6 is 1+(R6/ (
R7+R12>) However, (R7+R12> is changed to be greater than the amplification factor until full charge, like the parallel combined resistance value of R7 and R12.

Therefore, even if the feedback voltage is low after full charge, the output voltage of operational amplifier 6 becomes relatively high and the time constant becomes small, so the switching frequency is considered to be higher than the same feedback voltage until full charge. Become. As a result, as shown in FIG. 7, the series resonant circuit output becomes even smaller, and the charging current can be suppressed to a trickle current.

In this way, this embodiment uses the suppression control of the full 15fa flow, which synergistically takes into account changes in the pulse duty and the amplification factor of the operational amplifier to the switching frequency. Eliminates the difficulty of obtaining a suitable trickle current,
The adjustment range of the inverter output can be expanded.

Next, FIG. 3 shows another circuit example in which a part of the series resonant circuit surrounded by the broken line in FIG. 1 or 2 is extracted. In the figure, coils Lll and L12 are connected in series with coil T31 to the primary side of transformer T3. FIG. 4 is a sectional view showing an example of the structure of the transformer T3. In the figure, reference numeral 40 denotes an iron core having a figure-8 cross section, and a central piece 401 has a plurality of coils wound around it. That is, the coil Ll
l and L12 are respectively wound around both ends of the center piece 401,
The coil T31 is wound approximately at the center of the center piece 401,
Furthermore, secondary coils T32 and T33 are coaxially wound around the outer periphery of coil T31.

Coil T31 generates a magnetic flux φ3 that interlinks all other coils and a leakage magnetic flux φ31. Both of these magnetic fluxes interlink secondary coils T32 and T33, and transformer T3
fulfills its original role. In addition, most of the magnetic flux generated by the coil Lll and the coil L12 becomes leakage magnetic flux φ11 and φ12, and the secondary coils T32 and T33
Since the magnetic flux interlinking with the coil L is reduced, the coil L
11 and coil L12. The leakage inductance can be set large. By arranging the coils L11 and L12 in this way, they can be treated equivalently to the coil L1 shown in FIGS. 1 and 2. Therefore, by using the transformer T3 having the configuration shown in this embodiment, the coil L1 can be omitted and the number of parts can be reduced.

Similar to FIG. 3, FIG. 5 shows another circuit example in which a part of the series resonant circuit is extracted.

In this figure, the transformer T3 shown in FIGS. 1 and 2 is removed, and a resonant circuit consisting of the coil L1 and the capacitor C6, as well as diodes D6 and D for full-wave rectification of the AC output of the capacitor C6 are shown.
7, a diode bridge composed of D61 and D71 is used.

Of the AC output from capacitor C6, diodes D6 and D7 act in the forward direction when the coil Ll side is positive.
Conversely, when the coil Ll side is negative, it is output from diodes D61 and D71 which are in the forward direction, and is supplied to the load 5 or storage reservoir 7.

Similar to FIG. 3, FIG. 6 shows another circuit example in which a part of the series resonant circuit is extracted.

In this figure, in place of the transformer T3 shown in FIGS. 1 and 2,
The coil L1 is divided into a coil Lll and a coil L12 to form a transformer, and a capacitor C6 is connected in parallel to this transformer to form a resonant circuit.

The AC output generated across the capacitor C6 is alternately full-wave rectified by the diodes D6 and D7 and then supplied to the load side.

That is, in this embodiment, the load is not connected in series with two diodes, D6 and D7, or D61 and D71, as is the case, for example, in FIG. Since D7L is not connected in series, the influence of the forward voltage drop of the diode is small, the rectification loss is also minimized, and the inverter efficiency is improved.

〔Effect of the invention〕

According to the resonant inverter according to the present invention, the switching frequency of the switching element is always set higher than the resonant frequency of the resonant circuit, so even if the switching frequency changes according to load fluctuations, the switching frequency remains below the resonant frequency. Since it does not descend and does not enter the audible range, it is possible to prevent the generation of noise.

[Brief explanation of the drawing]

FIG. 1 is a circuit diagram showing one embodiment of a resonant inverter according to the present invention, FIG. 2 is a circuit diagram showing another embodiment, FIG. 3 is another circuit example of the resonant circuit portion, and FIG. A sectional view of the transformer T3 shown in FIG. 3, FIGS. 5 and 6 are other circuit examples of the resonant circuit portion, and FIG. 7 is a characteristic diagram showing the relationship between the switching frequency and the amount of energy flowing in the resonant circuit. FIG. 8 is a circuit diagram of a conventional resonant inverter.

Claims (1)

[Claims] 1. Current is supplied during the ON period of one of two switching elements connected in series to a DC power supply, the resonant circuit formed during the ON period of the other resonates, and the output of this resonant circuit is extracted and rectified. In a resonant inverter configured to supply power to a load, the switching frequency of the switching element is controlled to vary according to fluctuations in the load, the switching frequency being set higher than the resonant frequency of the resonant circuit. A resonant inverter characterized by: