JPH02239286A - Musical sound waveform generator - Google Patents

Abstract

(57) [Summary] This bulletin contains application data before electronic filing, so abstract data is not recorded.

Description

[Detailed Description of the Invention] [Industrial Application Field] The present invention relates to a musical sound waveform generator for an electronic musical instrument,
More specifically, the present invention relates to a musical sound waveform generator that performs frequency modulation to generate musical sound waveforms having various overtone characteristics.

[Conventional technology]

With the advancement of digital signal processing technology, as the first conventional example of electronic musical instruments using digital processing, it is possible to not only generate musical sound waveforms with simple characteristics, but also tones such as musical sounds of natural instruments, human or natural voices (hereinafter referred to as , collectively referred to as natural sounds), can be directly sampled and stored, and can be played back at any pitch.

On the other hand, as a second conventional example of an electronic musical instrument that can digitally generate musical sound waveforms with various types of complex characteristics,
There are electronic musical instruments based on the FM system described in Japanese Patent Application Laid-open No. 4-33525 or Japanese Patent Application Laid-open No. 126406/1982. This method is basically e = A-stn (ωct+I(L) sin (
L). L}...(1) The waveform output e obtained by the calculation formula is a musical sound waveform, and the carrier wave frequency ω. and a modulation wave frequency ω1 for modulating it in an appropriate ratio, set a modulation index r(t) that can change over time, and also set an amplitude coefficient A that can also change over time. By doing so, it is possible to obtain a very unique synthesized sound that has complex overtone characteristics and whose overtone characteristics can change over time.

Further, as a third conventional example of an improved FM system, there is an electronic musical instrument described in Japanese Patent Publication No. 12279/1983.

This method uses a triangular wave operation instead of the sine operation in equation (1), and obtains e−A・T (α+T(t) T (θ)) −−・
The waveform output e obtained by the arithmetic expression (2) is used as a musical sound waveform. Here, T(θ) is a triangular wave function generated by the carrier phase angle θ.

Then, by advancing the carrier wave phase angle α and the modulating wave phase angle θ at an appropriate advancing speed ratio, and setting the modulation index 1 (t) and amplitude coefficient A in the same manner as in the first conventional example, the musical sound waveform is can be synthesized. [Problem to be solved by the invention] With the background of the above-mentioned conventional technology, in recent years, electronic musical instruments have been improved in their ability to dynamically produce sounds ranging from the very unique musical tones unique to electronic instruments to natural sounds. It has been demanded.

However, the first conventional example, the PCM electronic musical instrument,
Although it is very good at producing natural sounds themselves, it is not good at processing natural sounds to produce unique tones. In other words, for example, when you want to change the original sound continuously into a sine wave, etc., you use a digital filter or an analog filter to remove the harmonic components of the original sound to obtain a sine wave, but with a digital filter, the circuit size is small. becomes relatively thick, and if you try to change its characteristics using a time function such as an envelope, it is necessary to store filter coefficients corresponding to the filter characteristics in addition to the natural sound data. . On the other hand, analog filters have problems in that desired characteristics are difficult to obtain and also they cannot perform time-sharing operations to generate a plurality of musical tones in parallel.

Furthermore, contrary to the above, if you want to continuously change the original sound to a musical tone with a more complex harmonic structure, it is not possible to generate new harmonic components using methods such as removing the harmonic structure of the original sound with the above filter. The problem is that it is impossible.

On the other hand, for example, the musical tones of an actual musical instrument such as a piano include, in addition to the fundamental wave component based on the pitch frequency, overtone components at a plurality of frequencies that are integral multiples of the fundamental wave component, and even considerably high-order overtone components exist. Furthermore, overtone components of non-integer multiples may be included. Additionally, the proportion of each higher-order overtone varies depending on the type of musical instrument, and each musical instrument has various overtone characteristics. In this way, musical tones with rich tonal qualities are generated by the presence of overtone components unique to each musical instrument. However, electronic musical instruments based on the FM method, which is the second or third conventional method, are very good at manipulating the overtone structure of the musical tones produced, but as an output, they have the above-mentioned characteristics unique to each instrument. When it is desired to obtain a desired musical tone, it is difficult to set the parameters universally.

That is, since the second conventional example is based on modulation using a sine wave, the frequency components of the musical tone generated by the equation (1) are concentrated in low-order (low-frequency) overtone components, Even if the modulation index 1 (t) is set to a large value and the modulation is applied deeply, the high-order (high frequency) harmonic components do not appear well. Therefore, the second conventional example has a problem in that it is not possible to generate musical tones with a rich quality similar to actual musical tones, and the quality of musical tones that can be generated is limited.

On the other hand, in the third conventional example based on the above formula (2), since the modulation is based on a triangular wave that originally contains many harmonics, musical tones that clearly exist even high-order harmonic components as frequency components are used. However, if you want to obtain a desired musical tone as an output, the carrier wave phase angle α and the modulated wave phase angle θ in the above equation (2) can be changed accordingly.
It is difficult to optimally determine the traveling speed ratio, modulation index r (t), amplitude coefficient A, etc. of . In addition to this, the third conventional example is a method of driving a triangular wave with a triangular wave, so for example, in the process of a musical tone gradually attenuating after it begins to sound, the amplitude decreases in order from high-order harmonic components. However, there is a problem in that it is impossible to realize a process in which there is only a single sine wave component corresponding to the pitch frequency.

The object of the present invention is to make it possible to faithfully produce natural sounds with a small circuit scale, to easily and continuously control its overtone components, and to easily synthesize musical sounds such as a single sine wave. It's about doing.

[Means to solve the problem]

The present invention first includes a carrier signal generating means for generating a carrier signal. The means receives as input a carrier wave phase angle signal in which the phase angle repeats an operation in which the phase angle increases sequentially and linearly over time, for example, and converts it according to a certain function and outputs it as a carrier signal. , a ROM, etc., which receives a carrier wave phase angle signal as an address input.
Note that the characteristics of the output carrier signal will be described later.

Next, it has modulation signal generation means for generating a modulation signal.
The means is a means for inputting, for example, the carrier wave phase angle signal, converting it according to a certain function and outputting it as a modulation signal, and is constituted by a ROM or the like having the carrier wave phase angle signal as an address input. Note that the characteristics of the output modulated signal will be described later.

Further, when the modulated signal is mixed with the carrier signal generated from the carrier signal generating means, the mixing ratio of the modulated signal to the carrier signal is controlled between 0 and an arbitrary mixing ratio, It has a mixing control means for outputting a mixed signal in which the modulated signal is mixed at the mixing rate. The means includes, for example, a multiplier that multiplies the modulation signal output from the modulation signal generation means by a modulation index whose value can vary between O and 1, and an output signal of the multiplier and the carrier. Adding the carrier signals generated from the signal generating means,
This is an adder that outputs a mixed signal. Note that the mixing ratio may change over time after the tone waveform starts producing sound. That is, for example, the modulation index multiplied by the above multiplier is
The multiplier operates to perform multiplication at each time that elapses after the start of sounding the musical sound waveform.

Furthermore, the apparatus further includes a waveform output means for outputting a musical sound waveform in which the input and the output have a predetermined functional relationship, and the mixed signal output from the mixing control means is manually modulated. The means is, for example, a decoder that converts the mixed signal according to the predetermined functional relationship and outputs it as a musical sound waveform. Alternatively, it is a ROM etc. that uses the mixed signal as an address input.

In addition to the above configuration, the predetermined functional relationship and the carrier signal are such that when the mixing control means controls the mixing ratio of the modulation signal to 0, the musical sound waveform generated from the waveform output means is The relationship is such that it becomes a sine wave or cosine wave with a single frequency. Further, the modulated signal is generated by the waveform output means when the mixing control means controls the mixing ratio of the modulated signal to a predetermined mixing ratio of 1, for example, with respect to the predetermined functional relationship and the carrier signal. The settings are made so that the desired tone waveform can be obtained. In this case, the carrier signal and the modulation signal are output from the carrier signal generating means and the modulating signal generating means at the predetermined period, respectively, so that the desired musical sound waveform is repeatedly output from the waveform output means at a predetermined period. This is a signal that is repeatedly generated. Alternatively, the carrier signal generating means and the modulating signal generating means are configured such that the desired musical sound waveform of the corresponding waveform section is outputted from the waveform output means for each of a plurality of waveform sections at constant periodic intervals. Generates a carrier signal and a modulating signal.

In addition to the above configurations, the present invention includes an amplitude envelope control means for temporally changing the amplitude envelope characteristic of the musical tone waveform output from the waveform output means. The means multiplies, for example, the musical sound waveform output from the waveform output means by an amplitude coefficient whose value can change over time, for example, from O to 1, according to a predetermined amplitude envelope function after the musical sound waveform starts producing sound. It is a multiplier. Note that the means may be a means for normalizing the amplitude of the output of the waveform output means to match the desired musical sound waveform. [Function] The function of the present invention is as follows.

The musical sound waveform outputted from the waveform output means basically has characteristics obtained by converting the carrier signal outputted from the carrier signal generation means according to a predetermined functional relationship, and furthermore, the musical waveform is converted into the carrier signal by the mixing control means. By mixing the modulation signals, characteristics of the musical sound waveform modulated by the modulation signals are added.

In this case, the relationship between the predetermined functional relationship in the waveform output means and the carrier signal from the carrier signal generating means is
The relationship is set such that when the mixing ratio of the modulation signal is controlled to be O by the mixing control means, the musical sound waveform generated from the waveform outputting means becomes a sine wave or a cosine wave of a single frequency.

Thereby, if the mixing ratio of the modulation signal is set to O in advance by the mixing control means, it is possible to generate a musical sound waveform consisting only of a sine wave or a cosine wave of a single frequency.

Furthermore, when the modulation signal is controlled by the mixing control means so that the mixing ratio of the modulation signal is, for example, 1 with respect to the predetermined functional relationship and the carrier signal, a desired musical tone is output from the waveform output means. The settings are made so that a waveform can be obtained. Thereby, if the mixing ratio of the modulation signal is set to 1 in advance by the mixing control means, it is possible to obtain a desired musical sound waveform, such as a musical tone of a natural musical instrument.

Also, during a performance, by setting the mixing ratio to 1 immediately after the start of sound generation, and then increasing the mixing ratio to 0 as time passes, it is possible to change the state of the desired musical sound waveform to a single sine wave component or a single sine wave component. The frequency characteristics of the musical sound waveform can be gradually controlled so that it contains only one cosine wave component. Alternatively, by continuously changing the mixing ratio to, for example, 1 or more, it is possible to control a desired musical sound waveform so that a unique musical tone having a more complex overtone structure is produced.

Along with the above operation, the amplitude envelope characteristic of the musical sound wave output from the waveform output means is also controlled by the amplitude envelope control means so as to attenuate over time, so that it can be reproduced from the start of sound generation, like the musical tone of an actual musical instrument. , it is possible to realize a process in which the musical sound waveform gradually attenuates.

〔Example〕

Embodiments of the present invention will be described below with reference to the drawings. FIG. 1 of 1 is a diagram showing the principle configuration of a first embodiment of a musical sound waveform generator according to the present invention.

The carrier wave phase angle ω whose value sequentially increases linearly between O and 2π (rad), and after reaching just before 2π, repeats the state of increasing from 0 again. , is taken as the address of the carrier wave ROMI and causes the carrier signal Wc to be read out. Here, the carrier wave phase angle ω, t is the value obtained by multiplying the angular velocity ω6 (rad/sec) by the time t (see), but unless otherwise mentioned, "." will be collectively expressed as a subscript. .

Further, the carrier wave phase angle ω. is also the address of the modulated wave ROM2, and the modulated signal output from here is added to the multiplier (
The modulated signal WM is obtained by multiplying the modulation index r(t) which can change over time by 3 (hereinafter referred to as MUL).

The modulation signal Wκ and the carrier signal Wc are added by an adder (hereinafter referred to as ADD) 4, and the sum waveform Wc+W
κ Crad ) is further decoded by a decoder 5 to obtain a decoded output D.

Then, the decoded output D is multiplied by the amplitude coefficient A in MUL6, and the final waveform output e is obtained as this output.

In the musical sound waveform generator having the above configuration, first, the carrier wave R
OMI stores the function waveform shown in Figure 2. Now, let π be pi, and the carrier wave phase angle ωCL (rad) and the carrier signal W( (r
ad), respectively, We = (K/2
) stn (1) ct... (Nod area I: 0≦ω.≦
π/2) Wc = π− (π/2) sin lJct・・ (Region ■: π/2≦ωct≦3π/2) W( =2π+ (
π/2) stn ωct... (area ■; 3π/2≦ω
．． ≦2π)...(3)

On the other hand, carrier wave phase angle ωd in modulated wave phase angle ROM2
The relationship between and the modulated signal Wκ[rad) after passing through MUL3 is wl4=t(t)Δf(ωcL) (4).

However, the function Δf is a phase difference function described later.

The carrier signal Wc and modulation signal WPl calculated by the above equations (3) and (4) are added and input to the decoder 5, which outputs the decoded output D. The waveform output e after being multiplied by the coefficient A is: e=A 4RI { (g/2) sin (1) at+
I (t) Δf(ωct) ) ... (O≦ω,,≦π/2) e=A 4RI {z-(π/2) sin a)ct
10t (t) Δr(ωc,) ) ... (π/2≦ω.≦3π/2) e=A-TRI (2 x+ (z/2) sin ω
ct+I(t)Δf(ωct)) (3π/2≦ωct≦2π) (5).

Here, first, when the value of the modulation index 1 (t) is O, that is, no modulation, the input waveform to the decoder 5 becomes the carrier signal Wc itself determined by the above equation (3). That is, e = A-TRI (We)...
- (6). Note that the carrier signal W, and the carrier phase angle ω
From equation (3) above or from FIG. 2, ct is determined by the relationship A in FIG. 3.
It is indicated by.

On the other hand, the triangular wave function DT calculated in the decoder 5
RI (X) (where X is the input) is D=TRI
(x) = (2/π) x... (Area I: 0≦X≦
π/2) D -TRI (x) =-1+ (2/π) (3π/
2-x)... (Area ■: π/2≦X≦3π/2)D=
TRI (X) = 1 + (2/x)(x-3
z/2) (region ■; 3π/2≦X≦2π) (7) This is a function shown in relation B in FIG.

As can be seen from the relationship A and relationship B in FIG. Or, it is a monotonically increasing function in each of the nodal ranges r, ■, and ■ defined in equation (7), and therefore, the carrier phase angle ω, which is the input in equation (3) above, is a monotonically increasing function. Since the input X in the equation (7) always takes a value in the same interval, the equations (3), (6), and (7) can be combined in the same interval. That is, by substituting the above equations (3) and (7) into the above equation (6), e=A-TRI ((π
/2) sin ωet)=A ・(2/i)hr/2
) sin ωet" A ' Sin ωct ・(Region I: 0≦ω.≦π/2) e=A-TRI [K - (z/2) sin ωet
)=A {-1+(2/π)(3π/2-π+ (π
/2) sin ωct) )= A −Sin ω
ct... (?J region ■: π/2≦ω.≦3π/2) e=A-
TRI (2π+ (z/2)sin ωct)=A・
(-1+(2/π)(2π+ +(π/2)sfnωct 3π/2)}"" A
-Sffl (+) ct... (Area ■: 3π/2
≦ω． ≦2π)・・・・(8) That is, when there is no modulation, the carrier phase angle ωcL
A single sine wave A-sinωeL containing no high-order overtones is output for any region.

That is, for example, if the amplitude coefficient A=1, the relationship between the carrier wave phase angle ωd and the waveform output e when no modulation is performed becomes a single sine wave as shown in the relationship C in FIG. From the above relationship, in order to realize the process in which a musical tone attenuates into only a single sine wave component, or the generation of a musical tone consisting only of a single sine wave component, the modulation index 1 must be It can be seen that the value of (t) should be brought closer to 0 with time.

Next, the phase difference function Δr in the modulated wave ROM2 in FIG.
(ωct) is the modulation index 1 (t
) is set to 1, and the addition waveform We is output from ADD4.
+w. =wC+Δf(ωcL) (see equation (4) above)
When accessing the decoder 5 using , the waveform output e is, for example, a sawtooth wave for one waveform section in which the maximum absolute value of the amplitude is normalized to 1, as shown in FIG. 4A (C). is the data that is read out, and the addition waveform Wc+
The same figure (b) shows the subtraction of the carrier signal Wc from Δf(ωct)
The waveform shown in the shaded area, ie, the waveform shown in FIG. 3A, is stored.

In addition, as shown in FIG. 4B (C), for example, a rectangular wave for one waveform section whose maximum absolute value of amplitude is normalized to 1 is read out as the waveform output e, and the addition The waveform of the diagonally shaded portion of the waveform we+Δr(ωct) minus the carrier signal WC, that is, the waveform shown in FIG. 3(a), is stored.

As yet another example, as the waveform output e, for example, FIG. 4C (
As shown in C), the data is such that an asymmetric rectangular wave for one waveform section whose maximum absolute value of amplitude is normalized to 1 is read out, and the carrier signal Wc is subtracted from the added waveform WC +Δf(ωct). The waveform shown in the shaded area in FIG. 5(b), that is, the waveform in FIG. 4(a), is stored.

As can be seen from these relationships, when the value of the modulation index r (t) is 1, the desired sawtooth wave,
If you want to obtain a rectangular wave or an asymmetrical rectangular wave, you need to find the corresponding phase difference function Δr(ωcL). The method for finding it is as follows.

■First, any carrier wave phase angle ω of one period of waveform output e whose amplitude is normalized within ±1. Find the amplitude of the normalized waveform output e corresponding to that phase angle. ■A triangular wave function whose maximum absolute value of amplitude is 1 (the above ((
7)), find the phase angle at the same position as the amplitude found in step (2) above.

■From the phase angle determined by the above ■, the carrier wave phase angle ωc of the above ■
The value of the carrier signal We corresponding to t is subtracted to obtain the phase difference function Δf(ωct) corresponding to the carrier wave phase angle ωcl of (2). (2) The above processes (2) to (2) are repeated by changing the carrier wave phase angle ωcl over the entire range of the waveform output e, and each carrier wave phase angle ωcl of the waveform output e is changed. Find the phase difference function Δr(ωcL) corresponding to .

The phase difference function Δr(ωct) obtained through the above processes ① to ② is stored in the modulated wave ROM 2 shown in FIG.

The above phase difference function Δr(ωet) is obtained for a plurality of waveform outputs e illustrated in FIGS. 4A to 4C, etc., and the modulated wave R
By storing it in OM2, selecting it as needed and outputting it, and setting the value of the modulation index (Bt) multiplied by MUL3 to 1, the sawtooth as illustrated in FIGS. 4A to 4C, etc. Various waveform outputs such as a shaped wave, a rectangular wave, and an asymmetrical rectangular wave can be obtained.

Next, in FIG. 1, consider the case where the value of the modulation index I(t) multiplied by MUL3 is changed.

First, if the modulation index I(t) = 0.0 (temporally constant value), the addition waveform wc+W as shown in FIG. 5(a)
．． = Wc, and an unmodulated sine wave waveform output e can be obtained.

Also, the values of modulation index Bt) are set to 0.3, 0.6, 1
．． By changing it to 0 (temporally constant value), the fifth
It is possible to obtain waveform outputs e that are sequentially deeply modulated as shown in FIGS. In this way, by changing the value of the modulation index rB), various modulated waveforms from the sine wave to the original waveform can be obtained.

In addition, by continuously changing the modulation index 1 (t) from the start of sound generation to the end of the sound, a waveform output e can be obtained that changes from a deeply modulated state to a sine wave as the musical tone decays. It also becomes possible. Furthermore, if the value of the modulation index 1 (t) is set to 1 or more, a waveform output e with deeper modulation can be obtained.

In the principle configuration of the above-mentioned musical waveform generator, the waveform output e of the decoder 5 having the characteristics shown in the equation (7) or the relationship shown in FIG. By storing the carrier signal Wc in the carrier wave ROM 1 as shown in the equation (3) above or the relationship A in FIG. 2 or 3, it is possible to generate a single sine wave, and the modulation index I When (t)=O, a sine wave was obtained as the waveform output e from the above equation (6).

In addition, the modulated wave ROM 2 is stored in Fig. 4A (a) and Fig. 4B (a).
) or a phase difference function Δ as exemplified in FIG. 4C(a)
By storing f(ωcL) and setting the modulation index 1 (L) = 1 in the equation (5) above, the result can be obtained as illustrated in Fig. 4A (C), Fig. 4B (C), and Fig. 4C (C). A desired waveform output e was obtained.

Therefore, if a sine wave is obtained when the modulation index I(t)=O, and a desired waveform output e is obtained when the modulation index I(L)-1, then the characteristics of the decoder 5 and the carrier stored in the carrier ROMI The combination of characteristics of the signal WC is expressed by the above formula (7) or the third
The decoder 5 is not limited to the combination of the relationship B shown in the figure and the above equation (3) or the relationship A shown in FIGS. 2 or 3. A calculation is performed, a function such that the decoded output D becomes a sine wave is stored in the carrier wave ROMI, and then a phase difference function Δf is stored in the modulated wave ROM2 so that the waveform output e becomes a desired waveform. A similar effect can be obtained by storing (ωct). In Figures 6(a) to (d),
An example of a combination of a function calculated by the decoder 5 and a function stored in the carrier wave ROMI will be shown. In the figure, carrier phase angle ω. A function that associates the input X with the carrier signal Wc is stored in the carrier wave ROMI, and a function that associates the input X with the decoded output D is calculated by the decoder 5. Also, the 6th
The characteristics corresponding to figures (a) to (d) are shown below. First, the function calculated by the decoder 5 of FIG. 1 corresponding to FIG. 6(a) is as follows.

D= (4/π)X ・ ・ ・ (0≦X≦π/4) D=1 ・ ・ ・ (π/4≦X≦3π/4) D=− (4
/π)X thousand 4 ・ ・ ・ (3π/4 ≦
/π)x-8 (7π/4≦X≦2π) (9) Also, corresponding to FIG.
The functions stored in are as follows.

Wc − (π/4) sin (dct...
(0≦ω.≦π/2) Wc = − (π/4) sin ωct+π...
(π/2≦ω.,≦3π/2)We = (π/4)s
in ω, t+2π... (3π/2≦ω.≦2π) ...Oo) Next, the function calculated by the decoder 5 in FIG. 1 corresponding to FIG. It becomes like this.

D=sin2x D=1 (0≦X≦π/4) (π/4≦X≦3π/4) Furthermore, the function calculated by the decoder 5 in FIG. 1 corresponding to FIG. 6(C) is It will look like this:

D=sin x ・ (O≦X≦π/2) D=stn (2X−π) ・ ・ ・ (3π/4≦X≦5π/4) D=−1 D−− (2/π)x+2 ・・ ・ (π/2≦X≦3π/2)D=sin
x ・ ・ ・ (5π/4≦X≦7π/4)D=sin
(2x-2π) (7π/4≦X≦2π) ・ ・ ・01) Also, corresponding to FIG. 6(b), the carrier wave ROM1 in FIG.
The functions stored in are as follows.

WC = ωct/2 ・ ・ ・ (0≦ω.≦π/2) Wc = ωct/2+z/2 ・ ・ ・ (π/2≦ωct≦3π/2) Wc"ωc
t/2+π (3π/2≦ω.≦2π) ・ ・ ・02) ・ ・ ・ (3π/2≦X≦2π) ・ ・ ・03) Also, in correspondence with Fig. 6 (C), Fig. 1 carrier wave R○M1
The functions stored in are as follows.

Wc=ωcL ・ ・ ・ (0≦ω.≦π/2) WC =− (π/ 2) sin ω1+π・ ・
・ (π/2≦ωc1≦3π/2) Wc”ω. ・ ・ ・ (3π/2≦ω., ≦2π)・ ・ ・0
Then, the function calculated by the decoder 5 of FIG. 1 corresponding to FIG. 6(d) is as follows.

D= (2/π)X ・ ・ ・ (0≦X≦π/2) D=sin x ・ ・ ・ (π/2≦X≦3π/2) D= (2/
π)
The functions stored in are as follows.

Wc = (π/2) sin ωct ... (O≦ωct≦π/2) WC =ωcL ・ ・ ・ (π/2≦ωct≦3π/2) Wc =
(π/2) stn ωct+2π・ ・ ・ (
3π/2≦ωcL≦2π)・06) Above, equation (9) and GO) equation, OD equation and 021 equation, 03)
When the value of modulation index 1 (L) in MUL3 in FIG. By inputting it as input X to , a single sine wave can be output as the waveform output e. In addition, a decoder 5 as shown in FIGS. 6(a) to 6(d)
In order to obtain a desired waveform output such as a sawtooth wave, a rectangular wave, or an asymmetrical rectangular wave as the waveform output e when the value of the modulation depth function I (t) is set to 1 by the function of FIG. - Determine the phase difference function Δr (ωct) in the same manner as the steps ① to ① explained in the case of Fig. 4C, and calculate the modulated wave RO
All you have to do is store it in M2.

Although the amplitude coefficient A multiplied by MUL6 in FIG. 1 has been explained as a constant value in each of the above embodiments, in reality it can change over time. Characteristics can be added.

Next, FIG. 7 is a concrete configuration diagram of the first embodiment of the musical sound waveform generator. This figure is based on the principle configuration of the first embodiment of the musical sound waveform generator shown in FIG.
The following description will be made with reference to FIG. 1 and the like from time to time.

The controller 7 operates according to the setting state in a parameter setting section (not shown) and the operation of a keyboard section, etc.
This is means for generating and outputting carrier frequency CF, modulator frequency MF, and envelope information ED (each rate value, level value, etc. of the envelope).

The adder 8 feeds back its output to the addend terminal B, and inputs the carrier frequency CF to the addition terminal A, thereby generating a carrier wave phase angle ωCtO to ωcLIO of 10 bits that increases sequentially by the step width of the frequency. This is an accumulator for Here, the carrier phase angle ωc
tO~ωcL10 is the carrier wave phase angle ω. in FIG. corresponds to Also, the carrier frequency CF is the carrier wave phase angle ωct
angular velocity ω. Corresponds to The carrier wave phase angles ωctO to ωctlo are input to the carrier wave generation circuit 9 and the modulated wave generation circuit 10 as respective address signals. Note that the carrier wave generation circuit 9 corresponds to the carrier wave ROM1 in FIG. 1, and the modulated wave generation circuit 10 similarly corresponds to the modulated wave phase angle ROM2.

On the other hand, the envelope generator 11 operates the terminals C and M based on the envelope information ED from the controller 7.
Modulation index 10 to 110 and amplitude coefficient AMPO to A M P 10 for two channels of 11 bits and 10 bits
Output. Note that each of these is the modulation index I in FIG.
(t) and the amplitude coefficient A, both of which can vary over time.

Modulation index ■ 0 to 110 takes a value of 1 or less, and the multiplier 12
The 11-bit modulated signal WH is inputted to terminal B of
Output O-WMIO. Here, the multiplier 12 and modulation signals WMO to W910 correspond to MUL3 and modulation signal W4 in FIG. 1, respectively. Carrier signals WcO~WclO and modulation signals WMO~ output from the carrier generation circuit 9 and the multiplier 12
WMIO is inputted to terminals A and B of adder l3, added together, and outputs an 11-bit addition waveform OO to 010. Note that the adder 13 and addition waveforms 00 to oio are ADD4 and addition waveform wc+W. in FIG. 1, respectively. corresponds to

The addition waveform OO~010 becomes the address signal of the triangular wave decoder 14, and the 10-bit decode output MAO~
Output MA9. Here, triangular wave decoder 14 and decode output MA O -MA 9 correspond to decoder 5 and decode output D in FIG. 1, respectively.

The above decoded outputs MAO~MA9 are further inputted to the terminal A of the multiplier 15, and the amplitude coefficients AMPO~ inputted to the terminal B.
It is multiplied by AMP9 and amplitude modulated. Note that the amplitude coefficients AMPO to AMP9 take values of 1 or less. The digital musical tone signal generated in this way is a D/A
The signal is converted into an analog musical tone signal by a converter 16 and a low-pass filter 17, and is emitted from a sound system (not shown).

With the above configuration, by controlling the carrier frequency CF and envelope information ED output from the controller 7 in response to the performance operation by the performer, the pitch, volume, tone color, etc. are controlled based on the performance operation. musical tones,
It is possible to output and emit sound in exactly the same manner as in the case of FIG.

Next, FIG. 8 shows the configuration of a first embodiment of the carrier wave generation circuit 9 of FIG. 7.

#0 to #9 exclusive OR circuit (hereinafter referred to as EOR) 1
9, the carrier wave phase angle ωedO of the most significant bit from the adder 8 in FIG.
O~ωct9 is input.

The output AO-A9 of the EOR 19 of #0 to #9 is input as each address signal of the A-wave carrier wave ROM 1B.

η wave carrier wave ROM18(7) ROM output DO~D9 is
Each of them is manually input to the first input terminal of EOR 20 #0 to #9. Further, the carrier wave phase angle ωcdO of the most significant bit is input to the second input terminal of the EOR 20 of #0 to #9.

Each output of EOR2 0 of #θ to #9 and the carrier wave phase angle ω of the most significant bit, L10 are carrier signals W c O to W
c10 is output to the adder 13 in FIG.

The operation of the first embodiment will be explained based on the operation diagram of FIG. 9.

First, the two-wave carrier wave ROM 1B in FIG. 8 stores a waveform corresponding to η periods (0 to π (rad)) of the carrier signal We explained in FIG. 2 or equation (3) above. There is. That is, from the above equation (3), if Y1 is the value determined by the outputs DO to D9 of the two-wave carrier wave ROM 1B in FIG. 8, then Y1= (π/2) sin (J).・・ (0≦ω.≦π/2) Yl=z − (g/2) sin (J)eL・・
(π/2≦ω.≦π) ...0'r) A waveform is stored. Note that the carrier wave phase angle ω, t here refers to a value determined by ωCLO to ωct9.

On the other hand, the carrier wave phase angle ω output from the adder 8 in FIG.
etO~ωctl. o can specify a phase angle of 0 to π (rad) by the full range of the lower 10 bits ωctO to ωet9 when the most significant bit ωetlo is logic 0, and furthermore, the most significant bit ωct10 is logic 1. Then, a phase angle of π to 2π (rad) can be specified by the full range of the lower 10 bits ωetO to ωct9.

Therefore, now, in the adder 8 of FIG. 7, the carrier phase angle ωctO
Letting T be the time during which the full range of ~ωctlo is designated, first, at time 0-T/2, as shown in FIG. 9(b), the carrier phase angle ω of the most significant bit is changed. 10 is logical O,
In addition, as shown in FIG. 4(a), the full range of carrier phase angles ωCtO to ωct9 of the lower 10 bits is specified. At this time, the carrier wave phase angle ωe of the most significant bit of logic O is input to the first input terminal of EOR 19 of #0 to #9.
Since t10 is input, the bottom 10 at time 0-T/2
The values of carrier wave phase angles ωctO to ωct9 of bits increase sequentially, and address signals AO to A9 are obtained which sequentially increase in exactly the same relationship as shown in FIG. 9(C). As a result, the outputs DO to D9 of the two-wave carrier wave ROM 18 in FIG.
As shown in FIG. 9(d), waveforms in the range O to π (rad) based on the equation θ'7) are sequentially read out.

This waveform is input to the first input terminal of the EOR 20 #0 to #9, but the carrier wave phase angle ωetlo of the most significant bit of logic O is input to the second manual terminal of the EOR 20.
is input, the carrier signal WcO-Wc9 of the lower 10 bits of the output is as shown in FIG. 9(e).
A waveform completely similar to the output Do-D9 in d) is output.

Furthermore, since the carrier signal WclO of the most significant bit is equal to the carrier wave phase angle ωct10 of the most significant bit and is logical O, as shown at time 0-T/2 in FIG. Waveforms similar to the output DO-09 of the field are output as carrier signals WcO to WclO.

Next, from time T/2 to T, as shown in FIG. 9(b), the carrier wave phase angle ωctlo of the most significant bit is logical 1, and as shown in FIG. 9(a), the carrier wave of the lower 10 bits is Phase angle ωCtO~ω. The full range of t9 is specified. At this time, the first input terminal of EOR1.9 of #0 to #9 has the carrier wave phase angle ωctl of the highest logic l and the
Since the carrier wave phase angle ωctO~ω.O of the lower 10 bits is input at the time T/2~T. By sequentially increasing the value of 9, as shown in FIG. 9(C), address signals AO to A9 are obtained which sequentially decrease in a completely opposite relationship.
As a result, the z-wave carrier wave ROM1B (7) output Do-D9 in FIG. 8 has a waveform in the range of 0 to π (rad) based on the surface equation in the opposite direction, as shown in FIG. 9(d). Read out.

This waveform is input to the first input terminal of the EOR 20 #0 to #9, but the carrier wave phase angle ωcilo of the most significant bit of logic 1 is input to the second input terminal of the EOR 20.
is input, the output carrier signal W c O = W c 9 of the lower IO bit is as shown in FIG. 9(e).
A waveform whose increase/decrease relationship is inverted with respect to the output Do-D9 in FIG. In addition, the carrier signal WdO of the most significant bit is equal to the carrier phase angle ω-tlo of the most significant bit (logical 1), so the above output corresponds to the full range of the carrier phase angle ωctO to ωct9 of the lower 10 bits. An offset ν of π (ra.d) is superimposed, and as a result, the waveform shown from time T/2 to T in FIG. 9(f) is output as carrier signals WcO to WdO.

As can be seen from the above operation, the waveform output in the range of time 0 to T is exactly the same as the carrier signal Wc explained in FIG. 2 or equation (3) above. In the case of this first embodiment, the two-wave carrier wave ROM 18 shown in FIG. The memory capacity can be reduced to % compared to the case where waveforms corresponding to cycles are stored.

FIG. 10 shows the configuration of a second embodiment of the carrier signal generating circuit 9 of FIG. 7. In FIG.

Each first input terminal of EOR22 #0 to #8 has a seventh
Figure 10th bit carrier phase angle ωct9 from adder 8
is input, and carrier wave phase angles θ to 8 bits ωctO to ωct8 are input to each second input terminal.

The outputs AO to A8 of the EORs 22 of #0 to #8 become respective address signals of the X-wave carrier wave ROM 21.

The two-wave carrier wave ROM2 1 (7) ROM output DO-DB is input to the first input terminal of the EOR 23 #0 to #8, respectively. Further, the 10th bit carrier wave phase angle ωct9 is input to the second input terminal of the EOR 23 of #0 to #8.

Each output of EOR20 of #0 to #8, 10th bit carrier wave phase angle ω. , 9 and the carrier phase angle of the most significant bit (J
) etlo is the seventh carrier signal WcO-WcLO.
It is output to the adder 13 in the figure.

The operation of the second embodiment will be explained based on the operation diagram of FIG. 11.

First, in Figure 10, X: I1. The vi transmitter ROM 21 includes
Of the carrier signal Wc explained in FIG. 2 or equation (3) above, a waveform corresponding to X periods (0 to π/2 (rad)) is stored. That is, from the above equation (3), the output Do~D from the X-wave carrier wave ROM 21 in FIG.
If the value determined by 8 is Y2, then Y2 - (z/2) sin (1) CL... (0≦
The following waveform is stored: ωct≦π/2)...08). Note that the carrier wave phase angle ω here. , means a value determined by ωCtO to ωct8.

On the other hand, the carrier wave phase angle ωc output from the adder 8 in FIG.
tO~ωct10 is a logical combination of the most significant bit ω,,10 and the 10th bit ωct9 (ωct10, ωc
t9), the lower 9 bits ωCt are (0, 0).
Due to the full range of O~ωet8, 0~π/2 (r
ad ] can be specified, and when it becomes (0, 1), the full range of the lower 9 bits ωeto to ωct8 can specify the phase angle of π/2 to π(rad), and (L
O), similarly π~3π/2 (rad)
Similarly, if the phase angle of (1, 1) can be specified, then 3
You can specify the phase angle between π/2 and 2π (rad). Each of the above four cases will be explained below.

Now, in the adder 8 of FIG. 7, the carrier phase angle ωctO ~ ωct
If the time when the full range of lo is specified is T, then (ωctlo, ωct9) = (0,
0) corresponds to time 0-T/4 as shown in FIGS. 11(B) and (C). In this time range, since the carrier wave phase angle ωCL9 of the 10th bit of logic O is input to the first input terminal of EOR22 of #0 to #8, the lower 9 bits are input at time 0 - T/4. carrier phase angle ωCtO~ω. , 8 are sequentially increased, an address signal AO-A8 is obtained which sequentially increases in exactly the same manner as shown in FIG. 11(d). As a result, the output DO~D of %u transmitter ROM 2 1 in Figure 1O
8, as shown in FIG. 11(e), waveforms in the range of O to π/2 (rad) based on the equation 08) are sequentially read out. Then, this waveform is input to the first input terminal of EOR23 of #0 to #8, but the above EOR2-3
Since the carrier wave phase angle ωcL9 of the 10th bit of logic O is input to the second input terminal of , the carrier signal W c O = W c 8 of the lower 9 bits of the output is
As shown in Figure (f), a waveform completely similar to the output DO-DB in Figure (e) is output. Further, the carrier signals W c 9 and WdO of the 10th bit and the most significant bit have carrier wave phase angles ωct9 and ωc of the 10th bit and the most significant bit.
Since both are equal to L10 and are logic 0, in the end, the first and first
Figure (as shown in the rumored time 0-T/4, Figure (e)
The carrier signal WcO~ has the same waveform as the output DO~D8 of
Output as WclO.

Next, as the second case, (ωctlo, ωCL9)=
(0, 1) corresponds to time T/4 to T/2 as shown in FIGS. 11(b) and (C). In this time range, since the carrier wave phase angle ωct9 of the 10th bit of logic l is manually input to the first input terminal of the EOR 22 of #0 to #8, the lower 9 bits are By sequentially increasing the values of the digital carrier phase angles ωcl to ωct8, address signals AO to A8 which sequentially decrease in a completely opposite relationship are obtained as shown in FIG. 11(d). As a result, the outputs Do to D8 of the A-wave carrier wave ROM 21 in FIG. 10 are as shown in FIG.
) A waveform in the range of 0 to π/2 (rad) is read out in the reverse direction. And the same waveform is #0 to #8 EO
It is input to the first input terminal of R23, but the above EOR23
Since the carrier wave phase angle ωct9 of the 1010th logical bit is input to the second input terminal of
A waveform in which the increase/decrease relationship is inverted with respect to the output DO-DB in FIG. 4(e) is output as shown in FIG. In addition to this, the carrier signals W c 9 and Wcl of the 10th bit and the most significant bit
O is the carrier phase angle ω of the 10th bit and the most significant bit
Since the respective logics are l and O, the carrier wave phase angle ω of the lower 10 bits is input to the above output.
π/2 (ra
The offset of dl is superimposed, and in the end, as shown in Fig. 11 ((to)
A waveform shown by the time T/4 to T/2 is output as the carrier signal W e O to W c 10.

Next, as the third case, (ωc(10, ωct9)
If = (1, 0), then Figure 11 (b) and (C)
As in, it corresponds to time T/2 to 3T/4. In this time range, the 10th bit carrier wave phase angle (d c
t 9 (0 3m Rika O Tealta reserve, EOR2
The operations in the two-wave carrier wave ROM 2 1 and the EOR 23 are exactly the same as in the first case. Therefore, E.O.
The carrier signal WeO of the lower 9 bits which is the output of R23
= Wc 8, as shown in Figure 11), outputs a waveform that is exactly the same as the outputs DO to D8 in Figure 11 (8).

In addition, since the carrier signals W c 9 and W c 10 of the 10th bit and the most significant bit are equal to the carrier wave phase angles ωct9 and ωcLIO of the 10th bit and the most significant bit, and their respective logics are O and l, The lower 9 in the above output
An offset of π (rad), which is twice the full range of the carrier wave phase angle ωcl to ωct8 of the bit, is superimposed, and in the end, the carrier signal has a waveform as shown in FIG. It is output as WCO to WclO.

Finally, as the fourth case, (ωct10, ωct9)
If = (1, 1), then Figure 11 (b) and (C)
As in, time 3T/4. ~ Corresponds to T. In this time range, the logic of the 10th bit carrier wave phase angle ωCL9 is 1, so the operations in the EOR 22, the two-wave carrier wave ROM 21, and the EOR 23 are exactly the same as in the second case. Therefore, the carrier signal W c O = W c 8 of the lower 9 bits of the output of the EOR 23 is as shown in FIG. 11(f).
A waveform in which the increase/decrease relationship is inverted with respect to the outputs DO to D8 in FIG. In addition, the carrier signals Wc9 and WdO of the 10th bit and the most significant bit have a carrier wave phase angle ωct of the 10th bit and the most significant bit.
9 and ωctlo, both of which are formula 1, the carrier wave phase angle ωCtO to ωct8 of the lower 9 bits is added to the above output.
3π/2 (rad
) offsets are superimposed, and eventually the waveform shown in FIG. 11 (powder time 3T/4 to T) is the carrier signal WcO=Wc
It is output as lO. As can be seen from the above operation, the waveform output in the range of time 0 to T is exactly the same as the carrier signal Wc explained in FIG. 2 or equation (3) above. In the case of this second embodiment, the Z-wave carrier wave ROM2 in FIG.
In Embodiment 1, since it is sufficient to store the waveform for X cycles in comparison to the waveform for 1 cycle in FIG. Compared to the case where the minute waveform is recorded, it can be set to X.

Next, FIG. 12 shows an embodiment of the triangular wave decoder 14 shown in FIG. 7. The two input terminals of #9 have addition waveforms 09 and 01 of the 10th bit and most significant bit from the adder 13 in FIG.
0 is input, and this output is input to each first input terminal of the EOR 24 #0 to #8. Also, EOR2 of #0 to #8
Each second input terminal of 4 has an addition waveform of O to 8 bits 00.
~08 is input.

Each output of EOR24 of #0 to #8 above is decoded output M
As AO-MA8, addition waveform 0 of the most significant bit
1.0 is output to the multiplier 15 in FIG. 7 as the decoded output MA9 of the most significant bit representing the sign bit.

The operation of the above embodiment will be explained below.

Now, assuming that the value Z determined by the addition waveforms 00 to 010 increases sequentially in direct proportion to the passage of time, the addition waveforms 00 to 010
One period in the full range of O~2π (rad)
Suppose that we can specify the phase angle of . First, in the first case, the logical combination (010, 09) of the most significant bit 010 of the addition waveform and the 10th bit o9 is (0, 0).
In this case, the value indicated by the addition waveforms 00 to 010 changes from O to one-fourth of the full range, that is, π/2 (rad). And in this range, #9 E
Since the output of OR24 becomes logic 0, EO of #0 to #8
As the addition waveforms OO to 08 input to R24 increase sequentially over time, waveforms exactly similar to them are added to the lower 9.
The decoded output of the bits appears as MA O - MA 8. Furthermore, the decoded output MA9 of the most significant bit, which is the sign bit, is equal to the addition waveform 010 of the most significant bit, which is logic O, and therefore generates a positive decoded output in the above range. Expressing this in a formula, the decode output MAO
~If the value determined by MA9 is W, then W=Z However, (0≦Z≦π/2)・・・・0
9) becomes. In the second case, when (010, 09) = (0, 1), the value indicated by the addition waveforms 00 to 010 is π/2
This is a case where the value changes up to π (rad). In this range, the output of the EOR24 of #9 becomes logic 1, so the addition waveform input to the EOR24 of #0 to #8 is OO to 0.
As 8 increases sequentially over time, a waveform that sequentially decreases in a completely opposite relationship is output as the lower 9 bits of decoded outputs MAO to MA8. Furthermore, the decoded output MA9 of the most significant bit, which is the sign bit, is equal to the addition waveform 010 of the most significant bit, which is logic 0, and therefore generates a positive decoded output in the above range.

Expressing this in a formula, W=-Z+π However, (π/2≦Z≦π) ・ ・ ・12G.

In the third case, when (010, 09) = (1, 0), the values indicated by the addition waveforms 00 to 010 are π to 3
This is a case where it changes up to π/2 (rad).

In this range, the output of the EOR24 of #9 becomes logic 1 as in the second case, so the output of the EOR24 of #0 to #8 is
The state of R24 is also the same as in the second case, and as the input addition waveform oO~08 increases sequentially over time, a waveform that decreases sequentially in a completely opposite relationship is used as the decoded output MAO-MA8 of the lower 9 bits. Output. On the other hand, the decoded output MA of the most significant bit which is the sign bit
9 generates a negative decoded output in the above range because the addition waveform 010 of the most significant bit has changed to logic 1. Expressing this in a formula, W=-Z+π However, (π≦Z≦3π/2) (21).

In the fourth case, when (010, 09) = (1, 1), the value indicated by the addition waveforms 00 to 010 is 3π/
This is a case where it changes from 2 to 2π (rad). In this range, the output of EOR24 of #9 becomes logic 0 as in the first case, so EOR24 of #0 to #8
The state of No. 4 is also the same as the first case, and as the input addition waveforms OO to 08 increase sequentially over time, a waveform exactly the same as the input addition waveform OO to 08 becomes the decoded output MA of the lower 9 bits.
Output as O-MA8. On the other hand, the decoded output MA9 of the most significant bit, which is the sign bit, generates a negative decoded output in the above range because the addition waveform 010 of the most significant bit is logic 1. Expressing this as a formula, W=Z-2π However, (3π/2≦2≦2π)・・・・(22). (19) to (22) corresponding to the first to fourth cases above
To summarize the formula, W-Z-Z However, (0≦Z≦π/2) W=-Z+π However, (π/2≦Z≦3π/2) W-Z-2π However, (3π/2≦2≦ 2π) ・ ・ ・(23)

Here, if we transform the equation (7) already shown as the characteristic of the decoder 5 in FIG. 1, D= (2/π)X ・ ・ (0≦X≦π/2) D− (2/π) (=X1π) ・ (π/2≦X≦3π/2) D− (2/π) (x − 2π) ・ ・ (3π/2≦X≦2π) ・ ・ ・(24) Comparing Equation (24) and Equation (23) above, the relationship between input and output is essentially the same, with only a difference of 2/π in the overall gain, and therefore, as shown in Figure 12. It can be seen that the triangular wave decoder 14 in FIG. 7 operates in exactly the same way as the decoder 5 in FIG. 1, which is shown by the characteristic of equation (7) above.

Subsequently, the modulated signal generation circuit 10 of FIG. 7 corresponds to the modulated wave phase angle ROM 2 of FIG.

First, FIG. 13 shows the configuration of the modulated wave phase angle ROM2.
This ROM has a 13-bit address input of AO-A12, and first, the upper 2-bit address AIlSA1
By inputting the value of θ to 2 (decimal) as the waveform number select signal WNα to 2, the address area where each of the three types of phase difference functions Δf[ωcL) shown in FIGS. 4A to 4C is stored, for example. Any one of them can be specified. Note that this designation can be made arbitrarily by the performer using a selection switch (not shown), and the controller 7 in FIG.
．． may be supplied to the modulation signal generation circuit 10.

After selecting the phase difference function Δf(ωct) in this way, the carrier wave phase angles ωCtO to ωct10 are inputted from the adder 8 in FIG. can be outputted via the modulation signal generation circuit 10 and multiplier 12 shown in FIG.

Above, specific circuits have been shown for the carrier signal generation circuit 9, modulation signal generation circuit 10, and triangular wave decoder 14 of the embodiment shown in FIG. 7, but the adder 8, IO, 14, multipliers l2, 16, etc. The envelope generator 11 can also be realized by using a circuit known in the art for electronic musical instruments.

Further, in the embodiment shown in FIG. 7, the circuit was explained as a circuit for outputting a musical sound waveform of one note, but the adder 8, carrier signal generation circuit 9, modulation signal generation circuit 10, envelope generator 11, Multiplier l2, adder l3, triangular wave decoder 14, and multiplier 15 are configured to operate in time division, and musical tones of each time division channel are accumulated at each sampling period at the input stage of D/A converter 16. By doing this, it becomes possible to generate multiple musical sound waveforms in parallel. 2. Next, FIG. 14 is a diagram showing the principle configuration of a second embodiment of the musical sound waveform generator according to the present invention. In the first embodiment shown in FIGS. 1 and 7, a desired waveform such as a sawtooth wave, a rectangular wave, an asymmetrical rectangular wave, etc. that repeats regularly at each pitch cycle can be obtained as the waveform output e, and The waveform characteristics can be continuously changed between the waveform and the sine wave. On the other hand, in the second embodiment described below, it is possible to further obtain an arbitrary waveform corresponding to the musical sound waveform of each pitch period from the start of sound generation to the end of sound of a natural instrument as a waveform output, and , is characterized in that the waveform characteristics can be continuously changed between the waveform and the sine wave. Hereinafter, the explanation will be given sequentially from FIG. 14 onwards.

First, periodic data F is output from the waveform data ROM 25 in accordance with address data Add(t) that increases over time.
(t), phase difference data M(t), and a normalization coefficient K (section) having a constant value for each waveform section (described later) are read out in synchronization with each other. First, phase difference data M(t.)(r
ad 3 is multiplied by the modulation index ■ by the multiplier 2, and then
The adder 27 adds the periodic data F(t) (rad) to obtain phase angle data F(t)+IM(t) for modulating a sine wave. The data is stored in a ROM memory that stores sine wave data.
The sfn wave is modulated from the in-ROM 26 and inputted to the same ROM as an address signal for reading. sin
The modulated output D(t) read out from the ROM 26 is multiplied by the normalization coefficient K (interval) read out for each waveform section from the waveform data ROM 25 in the multiplier 29, and then is output as the waveform output OUT(t). Output.

Here, the sin stored in sin RUM 2 6
The maximum absolute value of the wave amplitude is normalized to be 1.

The operation of the musical sound waveform generator shown in FIG. 14 based on the above-mentioned principle structure will be explained below.

First, the original waveform ORG(t) of the musical sound waveform of a natural instrument, etc. is
As shown in FIG. 15(a), for example, each periodic section of the fundamental wave (
The waveform is divided into waveform sections A-D for each pitch period). and,
Within each waveform section, O (rad) or more 2π (rad
), the periodic data F(t) read from the waveform data ROM 25 in FIG.
(rad). Now, by setting the modulation index I in FIG. 14 to 0 and using the periodic data F(t) itself, sin R
When the phase angle of the sine wave stored in the OM 2 6 is specified linearly to obtain the readout output sin (F (t)), each waveform section A to D is A sine wave of one period corresponding to a phase angle of 0 to 2π (rad) is read out without modulation. Note that each waveform section A-D does not need to correspond exactly to the pitch period; in particular, for musical sounds with weak periodicity such as percussion sounds, for example, from an appropriate zero-crossing point (when the amplitude is 0) to the zero-crossing point. may be taken as one waveform section.

Next, the phase difference data M(t) read from the waveform data ROM 25 in FIG. When the phase angle data F(t)+s(t) in (b) is used to modulate and read out one period of the sine wave stored in the sine ROM 26, the result is as shown in FIG.
As shown in a), the data is such that the original waveform ORG (t) for one waveform section in which the maximum absolute value of the amplitude is normalized to 1 is read out as the waveform output OUT (t),
This is shown in FIG. 16(C).

Further, the normalization coefficient K (interval) for each waveform section read from the waveform data ROM 25 in FIG. 14 is determined by the amplitude of the sine wave stored in the sine ROM 26 in FIG. Since the maximum absolute value is normalized to 1, the final original waveform ORG(t
) is a coefficient for returning to the amplitude of

From the above relationship, OUT(t) −K(interval) ・sin(F(t)+I
− M(t))・・・・(25) ORG (mu)1K(interval) ・sin(F(t)+M
(t)) ・ ・ ・(26) It can be seen that the following relationship exists. As can be seen from these relationships, for each waveform section, when the value of the modulation index ■ is set to 1, the desired original waveform ORG (
t), the original waveform ORG(
period data F(t) and phase difference data M(t) corresponding to
t) and the normalization coefficient K (interval). The way to obtain it is shown in waveform section A of the original waveform ORG(t) in Figure 15.
The case will be explained using an example.

First, regarding the method of deriving the periodic data F(t), see the 15th
This has already been explained in Figure (b). Next, in the waveform section A of FIG. 15(a), the original waveform ORG(t
) is set as the normalization coefficient K (interval A). Then, by dividing each amplitude value of the original waveform ORG(t) in the same interval by the normalization coefficient K (interval), the amplitude is normalized to within ±1 as shown in Fig. 16(a). do. In addition, in FIG. 16(a), the maximum value of both the positive and negative absolute values is 1, but the maximum value of either one may be 1 and the other may be 1 or less.

Next, the normalized original waveform OR obtained in this way
Using G (t), phase difference data M(t) is obtained by the following processes (1) to (2) (see FIG. 17).

■First, waveform section A of the normalized original waveform ORG(t)
Any time within 1. Therefore, the time L. The amplitude A. of the normalized original waveform ORG(t) corresponding to (Figure 17(a)). ■On a sine wave whose maximum absolute value is 1, the amplitude A. Find the phase angle ρ at a position equal to (Fig. 17)) In this case, the original waveform ORG
The position of time L8 within the sine wave and the position of phase angle p8 within the sine wave are determined to have approximately the same relationship. That is, for example, if time L8 is within about 1/4 from the beginning of waveform section 8, the phase angle is also within about 174 from the beginning, from 0 to π.
Try to decide around /2. ■Using the phase angle p1 found in the above (■) and the period data F(t) of the waveform section A determined in advance, the time 1. The phase difference data M(t,) corresponding to is obtained (FIG. 17(C)).

■Time 1. By changing the above throughout the waveform section A,
Repeat the process (2) to obtain phase difference data M (L) corresponding to each time t of waveform section 8.

The above processing of ■ to ■ is performed on the original waveform ORG (
t) is repeated for each waveform section A-D, and the period data p(L), phase difference data M(t), and normalization coefficient K (section) found for each waveform section are stored in the waveform data ROM shown in FIG.
25.

By performing the modulation operation with the value of the modulation index ■ multiplied by the multiplier 28 in FIG. 14 as 1 together with each of the above data, the waveform output OUT(t) in FIG. 14 is converted to the waveform output OUT(t) in FIG. An original waveform ORG(t) can be obtained.

Next, in FIG. 14, by changing the value of the modulation index 1 multiplied by the multiplier 28, various modulated waveform outputs OUT (L) can be obtained.

First, if the modulation index is set to [-0, an unmodulated sine wave can be obtained within each waveform section, as already shown in FIG. 15(C).

Also, the value of the modulation index ■ is 1.0, I. By changing 5 and 2.0, sin RO) 4 in Fig. 14 is obtained.
As the modulated output D(t) from 2 6, Fig. 18(a)
, (b), and (C), the waveforms are sequentially deeply modulated. As described above, by changing the value of the modulation index {circle around (2)}, various modulated waveforms can be obtained, from a sine wave to a deeply modulated waveform, centering on the original waveform.

In addition, by continuously changing the modulation index ■ from the start of sound generation to the end of the sound, a waveform output OUT (t) that changes from a deeply modulated state to a sine wave as the musical tone decays, for example, can be obtained. It also becomes possible.

Next, FIG. 19 is a specific configuration diagram of a second embodiment of the musical sound waveform generator. In the figure, parts given the same numbers as those in the basic configuration of FIG. 14 have the same functions as in FIG. 14. Note that the waveform data ROM 25 in FIG.
In Figure 9, FM ROM1 and K ROM1.2
It consists of two ROMs.

In FIG. 19, periodic data F(
t. ) and the address data ADD#1 for reading out the phase difference data M(t) are latched into the latch 30 in an accumulation section consisting of an adder 31, a selector ε2, and a latch 33, with the start address a as an initial value. The pitch data d is sequentially accumulated at address intervals in synchronization with the rising edge of the clock CLK#1 input to the latch 33. In this case, the start address a is output from a control section (not shown), selected by the selector 33 while the start pulse STRT is at a high level, and given to the latch 33 as the initial value of the accumulated value. If the start pulse STRT is at a low level, the output of the adder 31 is selected to perform the accumulation operation. Further, the pitch data d is output from a control section (not shown) and latched into the latch 30 in synchronization with the rise of the start pulse STRT. The phase difference data M(L) outputted from the FM ROML is multiplied by the modulation index I latched in the latch 40 in the multiplier 28, and the phase difference data M(L) output from the FM ROMII is multiplied by the modulation index I latched in the latch 40 in the adder 27.
It is added to the periodic data F(t) output from . The phase angle data F(L)+I − M(t
) is manually entered into the sin ROM 26 as a read address. Note that the modulation index {circle around (2)} is output from a control section (not shown), and is latched into the latch 40 in synchronization with the rising edge of the start pulse STRT.

As a result, the modulated output D(t) is output from the sin ROM 26 to the multiplier 29.

On the other hand, the section identification data IB output from the FM ROM1+ is manually transferred to a D flip-flop (F/F, hereinafter the same) 34 that operates in synchronization with the rising edge of the clock obtained by inverting the clock CLK#1 with the inverter 35. At the same time, it is input to the first input of an exclusive OR circuit (EOR, hereinafter the same) 36. Further, the positive logic output Q of the F/F 34 is input to the second input of the EOR 36. With the above circuit configuration, when there is a change in the value of the section identification data IB sequentially output from the FM ROM 11, the output of the EOR 36 becomes logic "1".

Address data ADD#2, which is the address input to the K ROM1z, is sent to the adder 37, selector 38, and latch 3.
In the accumulating unit consisting of 9, the first address b is used as an initial value, and each air address is accumulated sequentially each time the output of the EOR 36 becomes logic "1". Note that the start address b is output from a control unit (not shown), and is output from a start pulse S.
While TRT is at high level, it is selected by the selector 38 and given to the latch 39 as the initial value of the accumulated value. If the start pulse STRT is at a low level, the output of the adder 37 is selected to perform the accumulation operation. As a result, address data ADD# is stored in K ROM12.
2 is given, and a normalization coefficient K (interval) is output from the K ROM 12 to the multiplier 29.

In the multiplier 29, the modulation output D(t) output from the sin ROM 2 6 is multiplied by the normalization coefficient K (interval), and this multiplication result is further generated from the envelope generator 41 in the multiplier 42. Multiplied by envelope value.

The result of this multiplication is latched by the latch 43 in synchronization with the rising edge of the clock CLK#1 inverted by the inverter 44, and is output as a waveform output out(t).

In the above configuration, clocks CLK#1, CLK#
2 and the start pulse STRT are output from a control section, not particularly shown.

Next, the FM ROMI. Figure 20 shows the structure of the data stored in . In the figure, one set of waveform data from the start of sound generation to the end of sound is stored in order from the first address a, and one address has one cycle data F(t) and one phase difference data M(t). ) and 1-bit interval identification? Data IB are stored in pairs. In this case, as the addresses advance, the first
Data at each sampling point of each waveform section A, B, C, D, . . . in Fig. 5(a) is stored. Also, as the waveform section changes from A, B, C, D,...
The value of each address in the section identification data IB is "
"O" in section B, "l" in section C, "0" in section D, "1" in section D, and so on. This section identification data IB is data for identifying the boundary of the section and updating the address specified in KROMI■, as will be described later.

Next, FIG. 21 shows the structure of data stored in the K ROM 12 of FIG. 19.

In the figure, each waveform section A from the start of sound generation to the end of sound,
Corresponding to B, C, D, . . . , normalization coefficients K (sections) are stored one by one in order from the top address b.

The operation of the embodiment with the above configuration will be explained according to the operation timing chart of FIG. 22. In the following, unless otherwise specified, Figure 19 will be referred to. At the start of sound generation, a start pulse STRT output from a control section (hereinafter simply referred to as the control section), which is not particularly shown, is at t1 in FIG.
It rises to logic "1" (hereinafter simply referred to as "1". The same applies to logic "o") at the timing of , and immediately after that, the sound generation operation starts from the timing of L2 when clock CLK#1 becomes "1". do. In this case, the clock CLK
The period of #1 corresponds to the sampling period of musical tone generation, and the clock CLK#2 generated from the control section has the same period as the clock CLK#1, and is 1/4 from the same clock.
This is a clock delayed by a period. The following operations are controlled according to the two clocks CLK#1 and CLK#2. In addition, the start pulse STRT maintains "1" for one cycle from when the clock CLK#1 falls to "O" until the next fall to "O" at the time of starting sound generation, and thereafter Maintains "0" until the start of pronunciation.

First, the start pulse STRT from the control section is
Timing L when it rises to logic “1” as shown in figure (a)
1, the pitch data d from the control section is latched into the latch 30 as shown in FIG. Subsequently, during timings L1 to L4 when the start pulse STRT is "1", the selector 32 selects the start address a from the control section, and this start address a is selected by the clock CL.
K#1 latches 3 at timing t2 when it rises to IJ.
3, and the initial value of address data ADD#I is determined as shown in FIG. 22(d).

As a result, after a short delay time from t2, FI4
Period data F(t), phase difference data M(t.), and section identification data IB (20th
(see figure) are read out as shown in Fig. 22(e), (2) and (1).

Thereafter, the address data ADD#1 output from the latch 33 is fed back to the adder 31, and the pitch data d set in the latch 30 is successively accumulated. In this case, the above cumulative value is calculated at each timing 5, t8, and t11 shown in FIG. 22(d) when the clock CLK#1 rises to "1".
, tl4, tl7, etc., the new address data ADD is sequentially latched by the latch 33 via the selector 32.
Designated as #1, FM? The period data F(t), phase difference data M(t), and section identification data 1B (see FIG. 20) of the corresponding address on OM1+ are shown in FIG.
), (g) and (1.). In addition,
Since the start pulse STRT falls to r2 at t4, the selector 32 selects the output of the adder 31 after t4.

In this case, the instruction to start producing a musical tone is given by the performer pressing any key on the keyboard (not shown), for example, and if the pressed key at that time is a high-pitched key, , pitch data d having a large value is latched into the latch 30 from the control section. As a result, the address width skipped on the FM ROMII increases, and a waveform output OUT(t) with a high pitch period can be obtained. Conversely, for example, when the lowest pitch key is pressed, the value 1 is latched as the pitch data d, and each data is read out one address at a time on the FM ROM1+, and the waveform output of the lowest pitch cycle is output.
UT(L) is obtained.

Among the data read out from the FM ROMi as described above, the phase difference data M(t) is input to the multiplier 28. Now, at L1 when the start pulse STRT rises to "L", the control section sends the modulation index I to the latch 40.
is set. Therefore, in the multiplier 28, the phase difference data M(t) is multiplied by the modulation index 1. This output is input to the adder 27, where it is added to the periodic data F(t) output from FMROMI, to obtain phase angle data F(t)+I - M(t).

From the above phase angle data F(t)+I − M(t), s
in ROM 26 is accessed and the periodic data F(
t) and output of phase difference data M(t) (Figure 22 (b)
, (1) each timing L2, t5, t8, tll, t
l4, tl7, etc.) after a short delay, sin
A modulated output 0 (1) is output from the ROM 26 as shown in FIG. 22(n).

On the other hand, the timing t when the start pulse STRT is "1"
1 to t4, the selector 38 selects the start address b from the control section, and this start address b is the clock CLK.
Latch 39 at timing t3 when #2 rises to “1”
is latched, and the initial value of address data ADD#2 is determined as shown in FIG. 22(i).

This causes the κROM to open after a short delay from L3.
The normalization coefficient K (section A) of the first address b of 12 is the second
It is read out as shown in Figure 2 (j). Thereafter, the address data ADD#2 output from the latch 39 is fed back to the adder 37, and the logical output value "0" or r1 of the EOR 36 is
, are accumulated sequentially. In this case, the above cumulative value is calculated at each timing t6, t9, t10, tl2, t15, t when the clock CLK#2 rises to "1" in FIG. 22(i).
lB, etc., the latch 39 is sequentially selected via the selector 38.
is latched and designated as new address data ADD#2, and the normalization coefficient K (section) of the corresponding address on KROMlz is read out as shown in FIG. 22(j). In addition, the start pulse STRT is "
0'', the selector 38 selects the output of the adder 37 after t4.

In parallel with the above operation, the clock CLK#1 rises to "1" at each timing t2, t5, t8 of FIG. 22 5),
After a slight delay from tll, tl4, tl7, etc., FM ROMI. The section identification data TB are sequentially outputted from the section identification data TB. This data is clock C
At each timing t4, t7, t10, tl3, L16, tl9, etc. of FIG. 22~) when LK#1 falls to "0", it is set in the F/F 34 and its positive logic output Q is set as shown in FIG. ) are determined sequentially. Then, the EOR 36 calculates the exclusive OR of this positive logic output Q and the section identification data IB from the FM ROMI.

Here, the section identification data IB output from the FM ROMII is "
0" or "1" are stored alternately, so each time the waveform section changes from A to B, B to C, C to D, etc., "0" or "1" is stored alternately.
” to “1,” “1” to “0,” “0” to “1,” and so on. Therefore, the output of the EOR 36 becomes "1" only when the waveform section changes, and becomes "0" at other times. In the example of FIG. 22, the clock CL of ti6 starts immediately after tl4 when the waveform section changes from A to B.
The output of the EOR 36 becomes "l" only while K#l is "1". From this, only within the above timing, the adder 37 adds 1 to the value b of the address data ADD#2 from the latch 39.
is accumulated, and this accumulated value b+1 is obtained at timing L15 in FIG. 22 (c) when clock CLK#2 rises to "1".
, new address data ADD# is stored in the latch 39.
It is latched as 2. Therefore, after the timing tl5 when the clock CLK#2 rises to "1", based on the value b+1 of this new address data ADD#2, < K
The normalization coefficient K (section B) on the ROM 12 is shown in FIG.
) is read out as follows.

On the other hand, at other timings when the output of the EOR 36 is "OJ", the adder 37 does not perform the accumulation operation, so the latch 39 contains the same address data AD as one timing before.
D#2 is latched. Therefore, at timings L6, L9, L12, tlB, etc. in FIG.
1z as shown in FIG. 22(j).

In this way, FM ROMP. At the timing when the modulation output D(t) based on the period data F(t) and phase difference data M(t) of waveform section A from K ROMh is read out from stn ROM26, corresponding to waveform section A from K ROMh. The normalization coefficient K (section A) corresponding to waveform section B is read out, and similarly, the normalization coefficient K (section B) corresponding to waveform section B is read out from the K ROM 1z in waveform section B. Corresponding to the output of 0(1), the normalization coefficient K (interval) of that waveform interval becomes KRO
Read from M1z.

As described above, the clock CLK#1 of FIG.
Each timing t2, t5, t8 when rises to "1",
After some delay from Lll, tl4, tl7 etc., si
The modulated output D(t) is outputted from the ROM 2 6 as shown in FIG. 22(n), . In parallel with this, Figure 22 (5
22 (j
), the normalization coefficient K (interval) is output. Then, the modulation output D(t) at each timing above and the normalization coefficient K
(interval) is multiplied by the multiplier 29 and then further multiplied by the multiplier 4.
2 by the envelope data from the envelope generator 41 to generate the clock CLK# of FIG.
Each timing t4, t7, t1 when 1 falls to "0"
In 0Stl3, tl6, tl9, etc., Fig. 22 (0
), the waveform output 01JT (t) is determined for each timing. Here, although the envelope generator 41 and the multiplier 42 are not shown in FIG. 14, they may be operated when it is desired to add an envelope other than the original waveform to the waveform output OUT(t), and are not necessarily necessary. .

By the operation shown above, in the embodiment of FIG. 19,
The same operation as shown in FIG. 14 shown in FIGS. 15 to 18 can be realized.

In the electronic musical instrument shown in FIG. 19 above, those corresponding to the waveform data ROM25 shown in FIG.
Although the configuration is made up of two ROMs (ROM12), it may be configured with one ROM if it is acceptable that the address specification becomes somewhat complicated. Also, in FM ROM1+, periodic data F(t)
The values for each sampling point are stored as shown in Figure 20, but within one waveform section, the periodic data F(t) is stored at each waveform interval width as shown in Figure 15(b). Since the phase angle has a linear characteristic that changes from 0 to 2π (rad), if there is enough calculation time, by calculating and outputting the periodic data F(t) for each sampling point,
Saves storage space.

In particular, if the cycles of each waveform section A, B, C, D, etc. in FIG. 15 are selected to be constant over the entire section, for example, the upper bits of address data ADD#1 in FIG.
, and select the phase difference data M(t) in each interval using the lower bits, and directly set the lower bits as each value of the periodic data F(t). There is no need to memorize F(t).

In addition, in Fig. 19, the waveform output OUT for one musical tone is
(t), but it is also possible to output a plurality of musical sound waveforms in parallel by operating each section in a time-division manner.

In the principle configuration of the second embodiment of the musical sound waveform generator shown in FIG. 14, from the equation (25) above, when the modulation index 1=0, (Interval) ・sin(F(t))
...(27), the periodic data F(t) is data that linearly increases within each waveform interval as shown in Fig. 15, etc., and the normalization coefficient K (interval) is data that increases linearly within each waveform interval. Since the waveform output OUT (t) is constant, s
A kn wave was obtained. In addition, when the modulation index 1=1, as in equation (26) above, oRc(t) −K(interval) ・sin(F(t)+M
(t))...(26) The original waveform ORG(t.) was obtained. Then, by changing the value of modulation index 1 between O and 1, s
The waveform can be continuously changed from the in wave to the original waveform. Further, if the value of the modulation index 1 is set to 1 or more, it is possible to continuously change the waveform from the original waveform to a further modulated waveform.

In this way, in this embodiment, at least s
It is characterized by being able to output in-waves and original waveforms. Therefore, if the skn wave can be output when I-0, and the original waveform can be output when I=1, there is no need to adhere to equations (25) and (26), etc., and OUT(t) = K( section) -
f ( g (F(t)) + I − M(t))...
・(28) Any embodiment may be used as long as it realizes the calculations satisfying the following relationship.

For example, in the above equation (2nd), let the input be α, and let f be f(α
)=(2/π)α... (0≦α≦π/2) f(α)=-1+(2/π)(3π/2-α)... (
π/2≦α≦3π/2) f(α)=-1+(2/π)(α-3π/2)... (
3π/2≦α≦2π) ・ ・(29) Defined as a triangular wave function that satisfies the following, and with the input as β and g as g(β)=( π/2)simouth β ・・(0≦β≦ π/2) g(β)=π−(π/2) sinβ... (π/2≦β≦3π/2) g(β)=2π+(π/,7) sinβ... (3π/
2≦β≦2π) ・If defined as a function that satisfies (30), when the value of modulation index I is 0, that is, no modulation, the above equations (29) and (30) can be substituted into the above equation (28). By this, f(g(β))=
K (interval) ・r ((π, / 2) sin β)
=K(interval)・(2/π)(π/2)si loβ=K(
interval)・sin β・(O≦β≦π/2) f(g(β))=K(interval)・f((π−(π/2)・
sin β)) = K(section door (-1+(2/π)(3π/2-π+ (
π/2) sin β)) = K (interval)・Sfn β・(π/2≦β≦3π/2) f(g(β))−K(interval)・f(2π+(π/2)・
sin β) = K (interval)・(-1-}-(2/π)(2π++(π
/2) sin β-3π/2)) = K (interval)・si
nβ (3π/2≦β≦2π) (31). That is, a single sine wave is output when no modulation is performed.

Also, by substituting the above equations (29) and (30) into the above equation (2nd) and setting the modulation index 1=1, the original waveform ORG(t)
In order to obtain , the sine wave shown in FIG. 3
0), the phase difference data M(t
．． ).

In this case, as means for generating the triangular wave corresponding to stn R[lM 2 6 in FIG. 14, it is also possible to generate the triangular wave by using a decoder circuit or the like in addition to the ROM.

In addition to the above aspects, functions f and g that satisfy the above (2nd) formula
You can define various combinations of functions as .

[Effects of the Invention] According to the present invention, if the mixing rate of the modulation signal is set to 0 in advance by the mixing control means, a musical sound waveform consisting only of a sine wave or a cosine wave of a single frequency can be generated, and the mixing control If the mixing ratio of the modulation signal is set in advance to a predetermined mixing ratio (for example, 1) by the means, it is possible to obtain a desired musical sound waveform such as the musical tone of a natural musical instrument. Therefore, during a performance, by setting the mixing ratio to 1, for example, immediately after the start of a musical tone, and then increasing the mixing ratio to 0 as time passes, the desired musical sound waveform can be changed to a single sine wave component or a single sine wave component. The frequency characteristics of the musical sound waveform can be gradually controlled so that it contains only one cosine wave component. Alternatively, by continuously changing the mixing ratio to, for example, 1 or more, it is possible to control the state of the desired musical sound waveform so that a unique musical tone having a more complex overtone structure is produced. Along with the above operation, the amplitude envelope characteristic of the musical sound wave output from the waveform output means is also controlled by the amplitude envelope control means so as to be attenuated over time, so that the sound waveform can be reproduced after the start of sound generation, like the musical tone of an actual musical instrument. It is possible to realize a process in which a musical sound waveform gradually attenuates.

As described above, the present invention can easily generate both a state in which natural musical tones are produced and a state in which only a single sine wave component or a single cosine wave component is generated, and also in a variety of ways. It is possible to generate overtone characteristics.

Moreover, as a configuration to realize this, a normal RO
Since it can be realized only by combining M, decoders, adders, multipliers, etc., it is possible to realize complex musical sound waveforms with a simple circuit configuration, and as a result, high-quality electronic musical instruments can be provided at low cost. becomes possible.

[Brief explanation of drawings]

FIG. 1 is a diagram showing the basic configuration of the first embodiment of the musical waveform generator according to the present invention, FIG. 2 is a diagram showing the storage contents of the carrier wave ROM, and FIG. FIG. 4A, an explanatory diagram of the operation of the embodiment when no modulation is performed, shows Δf(ωc
t)・Wc, Figure 4B shows Δf(ωct
), Wc, and Figure 4C shows the relationship between Δt(
Figure 5 shows the relationship between We and W when I (t) is changed.
FIG. 6 is a diagram showing another embodiment of the stored waveforms of the carrier wave ROM and the triangular wave decoder. FIG. 7 is a diagram showing the specific example of the first embodiment of the musical waveform generator. FIG. 8 is a configuration diagram of the first embodiment of the carrier signal generation circuit. FIG. 9 is an explanatory diagram of the operation of the first embodiment of the carrier signal generation circuit. FIG. 10 is a diagram of the carrier signal generation circuit. A block diagram of the second embodiment of the circuit, FIG. 11 is an explanatory diagram of the operation of the second embodiment of the carrier signal generation circuit, FIG. 12 is a block diagram of the triangular wave decoder, and FIG. 13 is a modulated wave ROM 14 is a diagram showing the principle of the second embodiment of the musical waveform generator according to the present invention. FIG. 15 is a diagram showing the relationship between ORG (t) and F(t). The figure shows the relationship between ORG(t), F(t), and M(t), Figure 17 shows how to obtain M(t), and Figure 18 shows the relationship between ORG(t), F(t), and M(t). D(t), F(1;) and M(t
), FIG. 19 is a specific configuration diagram of the second embodiment of the musical waveform generator, FIG. 20 is a data configuration diagram of the FM ROM, and FIG. 21 is a diagram showing the data configuration of the K ROM. The configuration diagram, FIG. 22, is an operation timing chart of a specific configuration of the second embodiment of the musical sound waveform generator. E... Carrier wave ROM, 2... Modulation wave ROM, 3, 6, 28, 29... Multiplier, 4, 27... Adder, 5... Decoder, 25... Waveform data ROM , 26...sinROM, ωd...carrier phase angle, ω. t...Modulation wave phase angle, ω...Modulation wave correction phase angle, Wc...Carrier signal, WM...Modulation signal, r(t), ■...Modulation index, Wc +wM ・... Addition waveform, D ... Decode output, A ... Amplitude coefficient, e, our (t) ... Waveform output, AtJd (t) ... Address data, F (L) ...
・Period data, M(t)...Phase difference data, F(t)+T - N(t)...Phase angle data, κ(
section)...normalization coefficient, D(t)...modulation output. Patent distributor Casio Computer Co., Ltd.? S] ζ, Oga and - ROM word Otsu f Onnai, ≧ copy = is shown, Otsu R figure Fig. Mishiki Hose J is the starting point, 1, 1 broom. Fl's unchangeable 1st week Saki's power-made customer illustration Figure 3 Zuhen vAron ROM's Bokuba diagram Figure 3

Claims (1)

[Claims] 1) A carrier signal generating means for generating a carrier signal, a modulating signal generating means for generating a modulated signal, and a method for mixing the modulated signal with the carrier signal generated from the carrier signal generating means. Mixing control means that controls a mixing ratio of the modulated signal to the carrier signal between 0 and an arbitrary mixing ratio, and outputs a mixed signal in which the carrier signal and the modulated signal are mixed at the mixing ratio; waveform output means for outputting a modulated musical waveform by inputting the mixed signal output from the mixing control means, the input and output of which have a predetermined functional relationship, the predetermined functional relationship and the carrier signal; is such that when the mixing rate of the modulation signal is controlled to be 0 by the mixing control means, the musical sound waveform generated from the waveform outputting means becomes a sine wave or a cosine wave of a single frequency. the modulation signal has a relationship with the predetermined functional relationship and the carrier signal, and when the mixing control means controls the mixing ratio of the modulation signal to a predetermined mixing ratio, the waveform output A musical sound waveform generator, characterized in that the device is configured to obtain a desired musical sound waveform from the means. 2) When the carrier signal and the modulation signal are controlled by the mixing control means so that the mixing ratio of the modulation signal becomes the predetermined mixing ratio, the desired musical waveform is output from the waveform outputting means to a predetermined value. so that it is output repeatedly with a period of
2. The musical sound waveform generating device according to claim 1, wherein the signals are repeatedly generated by the carrier signal generating means and the modulating signal generating means at the predetermined period. 3) The carrier signal generating means and the modulating signal generating means are each controlled by the mixing control means so that the mixing ratio of the modulating signal becomes the predetermined mixing ratio for each of a plurality of waveform sections having a constant period. 2. The musical sound waveform generating device according to claim 1, further comprising: generating a carrier signal and a modulation signal such that the desired musical sound waveform of the corresponding waveform section is outputted from the waveform output means when the desired musical sound waveform is generated in the corresponding waveform section. 4) The musical tone waveform generating device according to claim 1, further comprising amplitude envelope control means for temporally changing the amplitude envelope characteristic of the musical tone waveform outputted from the waveform output means.