JPH0354399B2 - - Google Patents

Description

[Detailed Description of the Invention] <Industrial Application Field> The present invention converts the output signal of a detector that detects physical quantities such as pressure and displacement into a corresponding current, and transmits the signal remotely via a two-wire transmission line. The present invention relates to a two-wire transmission circuit for transmitting data to a receiving unit.

<Prior art> As is well known, a two-wire transmission circuit converts the output signal of a detector into a corresponding current and then supplies it to a two-wire transmission line, which is advantageous because it only requires two transmission lines. Widely used in the field of industrial measurement.

The first embodiment of this type of conventional two-wire transmission circuit is described below.
As shown in the figure. Terminals L ₁ and L ₂ are connected to a power source E _B via a receiver A that receives current. The power source E _B is, for example, a 24 V DC power source. An output transistor Q ₁ and a feedback resistor R ₁ are connected in series to terminals L ₁ and L ₂ . A constant current circuit CC1 and a Zener diode _ZD are connected to both ends of the output transistor _Q1 .
A constant voltage E _Z is obtained across the Zener diode Z _D. SC1 is a _sensor circuit that detects physical quantities and converts them into electrical _{signals .}
is energized through. The output of the sensor circuit SC1 is connected from the terminal T ₃ to the operational amplifier Q ₂ via the resistor R ₂
is connected to the non-inverting input terminal of The non-inverting input terminal is also connected to the voltage dividing point of resistor R ₁ via resistor R ₃ , and the voltage obtained by dividing the constant voltage E _Z by resistor R ₄ is applied to the inverting input terminal of operational amplifier Q ₂ . There is. resistance
R ₁ , R ₂ , R ₃ and R ₄ constitute one resistor network. The output transistor _Q1 is connected to be controlled by the output of the operational amplifier _Q2 .

The difference between the voltage signal from sensor circuit SC1 and the zero point setting voltage signal from resistor _R4 controls the current in output transistor _Q1 via operational amplifier _Q2 .
The current flowing through the output transistor Q ₁ and the constant current circuit CC1 flows through the resistor R ₁ , and a voltage signal proportional to this current is fed back to the operational amplifier Q ₂ via the resistor R ₃ , and is applied between both input terminals of the operational amplifier Q ₂ . It operates so that the potential difference between the two becomes zero. As a result, the sensor circuit
The output of SC1 is the transmission current I _L flowing between terminals L ₁ and L ₂
This is displayed on the receiving section A, and functions as a so-called two-wire transmission circuit.

In the conventional embodiment shown in FIG. 1 configured as described above, the voltage signal output from the sensor circuit SC1 is accurately divided by resistors R ₂ and R ₃ and applied to the non-inverting input terminal of the operational amplifier Q ₂ . There is a need to.
For this reason, expensive precision resistors are required as the resistors R ₂ and R ₃ , which has the drawback of increasing the overall cost.

<Object of the Invention> In view of the above-mentioned prior art, an object of the present invention is to provide a two-wire transmission circuit that can obtain necessary accuracy at low cost without using particularly expensive precision resistors.

<Configuration of the present invention> The configuration of the present invention that achieves this object relates to a two-wire transmission circuit that supplies a transmission current to a receiving section via a two-wire transmission path, and a duty cycle according to a physical quantity to be measured. A sensor circuit section that outputs a first pulse signal whose ratio changes, a duty ratio conversion circuit that outputs a second pulse signal whose duty ratio changes according to the voltage corresponding to the transmission current, and a pulse width of the first pulse signal. a pulse calculation circuit that outputs a third pulse signal having a pulse width that is the difference between the pulse width of the second pulse signal and the pulse width of the second pulse signal; The device is characterized in that it has a control section that controls the transmission current based on the output.

<Example> Hereinafter, an example of the present invention will be described based on the drawings.

FIG. 2 is a block diagram showing one embodiment of the sensor circuit section in the present invention. The sensor circuit section SC2 in FIG. 2 includes a detection section PCC that converts differential pressure etc. into capacitance (however, a temperature compensation capacitor is not shown) and a capacitor that converts changes in capacitance in the detection section PCC into a duty ratio. /Duty ratio conversion section
It is composed of CDC.

The detection part PCC has a movable electrode between fixed electrodes SP ₁ and SP ₂ .
An MP is provided so that a mechanical displacement ΔP is generated in response to changes in the physical quantity to be detected.
Fixed electrode SP ₁ is connected to terminal T ₄ , fixed electrode SP ₂ is connected to terminal T ₅
, the movable electrodes MP are respectively connected to the terminals _T6 .

A capacitor C ₁ is formed between the fixed electrode SP ₁ and the movable electrode MP, and a capacitor C ₂ is formed between the fixed electrode SP ₂ and the movable electrode MP. It is designed to vary differentially depending on the displacement ΔP. If the capacitances of capacitors C ₁ and C ₂ when ΔP=0 are C ₀ , each is expressed by the following equations.

C ₁ =C ₀ 1/1−k ₁ ·ΔP (1) C ₂ =C ₀ 1/1+k ₁ ·ΔP (2) However, k ₁ is a constant.

The capacitance/duty ratio converter CDC is connected to the power supply terminal T ₇ ,
It has an output terminal _T8 . A constant voltage _VZ is applied to the power supply terminal _T7 , which is connected to each terminal a of the switches _SW1 and _SW2 via a constant current circuit CC2. Terminals b of switches _SW1 and _SW2 are both connected to a common potential point COM. Switch SW ₁
Terminal C of switch SW 2 is connected to fixed electrode SP ₁ through terminal T ₄ , and terminal C of switch SW ₂ is connected to fixed electrode SP ₂ through terminal T ₅ .
are connected to each. Also, switch
Terminals C of SW ₁ and SW ₂ are connected to one end of resistors R ₅ and R ₆ , respectively, and the other end is connected to the non-inverting input terminal of comparator Q ₃ . A comparison voltage V _C1 controlled by the output of the comparator Q ₄ is applied between the inverting input terminal of the comparator Q ₃ and the common potential point COM.
The output terminal of the comparator Q ₃ is connected to the input terminal CL of the flip-flop FF, and its output terminal Q is connected to the output terminal T ₈ and simultaneously switches the terminals C of the switches SW ₁ and SW ₂ to the terminal a or b. The output terminal Q of the flip-flop FF is also the input terminal of the monostable circuit MM.
CL, and its output terminal Q is connected to a common potential point COM via a series circuit of a resistor _R7 and a capacitor _C3 . A smoothing circuit is made up of resistor _R7 and capacitor _C3 . The connection point between resistor R ₇ and capacitor C ₃ is connected to the inverting input terminal of comparator Q ₄ , and the comparison voltage V _c2 is applied to its non-inverting input terminal.

Next, the sensor circuit section SC2 configured as above
The operation will be explained with reference to FIG. Third
The figure is a waveform diagram showing waveforms of various parts of the embodiment of FIG. 2. First, when the waveform C at the output end of the flip-flop FF becomes "L" level according to the waveform diagram of FIG. 3, terminals a and c of the switch _SW1 are connected. As a result, the capacitor _C1 is charged with a constant current _i1 by the constant current circuit CC2, and the voltage across it increases linearly as shown in FIG. 3A. switch at the same time
Terminals b and c of SW ₂ are also connected, and the charge on capacitor C ₂ is discharged to a common potential point, so that the voltage across them becomes zero potential as shown in FIG. 3B. Therefore, at the non-inverting input of comparator Q ₃ , resistors R ₅ and R ₆
A value obtained by dividing the voltage across capacitor _C1 is applied. Capacitor _C1 is charged until this value matches comparison voltage V _C1 . Assuming that the values of resistors R ₅ and R ₆ are equal, this point is the point where the voltage across capacitor C ₁ becomes 2V _C1 . The time T _A required to reach this point corresponds to the capacitance C ₁ of the capacitor C ₁ and is expressed by the following equation.

T _A = C ₁ /i ₁ x2V _c (3) The voltage at the non-inverting input terminal of comparator Q ₃ is the comparison voltage V _c1
When , the output of comparator Q ₃ is inverted and the switch is
SW ₁ and SW ₂ are switched to terminals b and a, respectively. As a result, the voltage across capacitor C ₁ becomes zero potential (Figure 3A), and now capacitor C ₂
is charged with a constant current i ₁ from the constant current circuit CC2, and the voltage across this terminal increases linearly (Fig. 3B).
When the voltage across capacitor C ₂ becomes 2V _c1 , the output of comparator Q ₃ is inverted. Time until reversal T _B
is proportional to the capacitance of capacitor _C2 , and is expressed as follows.

T _B = C ₂ /i ₁ x2V _c (4) On the other hand, when the waveform at the output terminal Q of the flip-flop FF (C in Figure 3) becomes an “L” level, this is simultaneously applied to the input terminal CL of the monostable circuit MM. , at its output terminal Q, there is a constant pulse width with a peak value of V _Z determined by a monostable circuit MM as shown in Figure 3D.
A pulsed voltage with T _c is obtained. This voltage is smoothed by resistor _R7 and capacitor _C3 , applied to the inverting input terminal of comparator _Q4 , and compared with comparison voltage V _C2 . As a result of the comparison, the voltage at the inverting input terminal is the comparison voltage.
Balance is achieved by adjusting the comparison voltage V _C1 so that when it is larger than V _C2 , the comparison voltage V _C1 is lowered, and when it is smaller than the comparison voltage V _C2 , _it is raised. Therefore, in an equilibrium state, the following equation holds.

T _c /T _A +T _B V _Z = V _C2 (5) Here, since V _Z , V _C2 and T _c are constant,
Assuming that K is a constant value, T _A +T _B =K (6).

As can be seen from the above explanation, a pulse signal S ₀ having a constant period ( _TA + T _B ) is obtained at the terminal T ₈ .

Next, a description will be given of the embodiment shown in FIG. 4 in which a two-wire transmission circuit is constructed using the sensor circuit SC2 described above.

An output transistor Q ₅ , _a diode D ₁ and a feedback resistor R ₈ are connected in series to the terminals L 1 and L ₂ . A constant current circuit CC1 and a Zener diode ZD are connected in series to both ends of the series circuit of the output transistor _Q5 and the diode _D1 , and a constant voltage _VZ is obtained between both ends of the Zener diode. Operational amplifier _Q6 is energized by this constant voltage _VZ , and its output voltage controls output transistor _Q5 . operational amplifier Q ₆
and the output transistor _Q5 constitute a control section that controls the transmission current I _L.

The voltage obtained by subtracting the bias voltage V ₁ from the feedback voltage R ₈ I _L generated across the feedback resistor _R 8 is divided by the voltage division ratio β using the resistor R ₉ to obtain the feedback voltage V _f . power terminal
A constant current circuit CC3 that supplies a constant current _i2 and a temperature compensation capacitor _C4 provided for temperature compensation in the detection section PCC are connected in series between _T7 and the voltage dividing point of the resistor _R9 . There is.

The input terminal C of the D-type flip-flop DFF is connected to the output terminal _T8 to receive the pulse signal _S0 .
This data terminal D is connected to the power supply terminal _T7 and is supplied with a constant voltage _Vz . The output terminal Q of the D-type flip-flop DFF is connected to one end of the input of the NOR gate _G1 , and the other end of the input is connected to the output terminal _T8 via the inverter _G2 . The voltage at the output terminal Q of the D-type flip-flop DFF also controls switch _SW3 . Terminal a of switch _SW3 is connected to the voltage dividing point, and terminal b is connected to constant current circuit CC3,
Connection point and comparator with temperature compensation capacitor C ₄
Connected to the non-inverting input of _Q7 . A comparison voltage V _c3 is applied between this inverting input terminal and the common potential point COM.
is given, and its output is a D-type flip-flop
It is applied to the reset terminal R of the DFF to control it.

On the other hand, the output of NOR gate G ₁ passes through a smoothing circuit consisting of resistor R ₁₀ and capacitor C ₅ to voltage V ₂
is applied to the non-inverting input terminal of operational amplifier _Q6 as
A zero point adjustment voltage V ₃ obtained by dividing the constant voltage V _Z by a resistor R ₁₁ is applied to its inverting input terminal.

Next, the operation of the embodiment of FIG. 4 constructed as above will be explained using the waveform diagram of FIG. 5.

In Fig. 5, A is the waveform of the pulse signal _S0 output from the capacitance/duty ratio converter CDC, B is the waveform of the pulse signal _S1 at the non-inverting input terminal of the comparator _Q7 , and C is the output of the D-type flip-flop DFF. Pulse signal S ₂ at terminal Q
D shows the waveform of the pulse signal _S3 at the output end of the NOR gate _G1 .

First, it is assumed that the pulse signal _S2 is in the "L" state, terminals a and b of the switch _SW3 are short-circuited, and the potential of the non-inverting input terminal of the comparator _Q7 is at the same potential as the feedback voltage _Vf . . When the pulse signal _S0 rises in this state, the output of the D-type flip-flop DFF is inverted to the "H" state, and terminals a and b of the switch _SW3 are opened. Therefore, a constant current i flows from the constant current circuit CC3 to the temperature compensation capacitor _C4 , so the comparator _Q7
The potential at the non-inverting input terminal of _Q7 increases linearly from the feedback voltage V _f and reaches the comparison voltage V _c3 .
The output of is inverted and resets the D-type flip-flop DFF. Therefore, the pulse signals S ₁ and S ₂ are switched
Period from open to short circuit of terminals a and b of SW ₃ T _D
The pulse width is equal to . Therefore, the following equation holds true.

C ₄ (V _c3 − V _f )=i ₂ T _D (7) Pulse inverted from pulse signal S ₀ and pulse signal S ₂
By passing through the NOR gate _G1 , the output terminal receives a pulse signal S3 whose pulse width is the difference between the pulse widths of the pulse signals _S0 and _S2 (T _A − T _B ) and whose peak value is _{V Z.} is obtained. Smoothing this with resistor R ₁₀ and capacitor C ₅ results in voltage V ₂ expressed by the following equation.

V ₂ = T _A − T _D / T _A + T _B xV _Z (8) Operational amplifier Q ₆ connects the output transistor so that voltage V ₂ and voltage V ₃ for zero point adjustment are equal.
The transmission current I _L is controlled via Q ₅ to vary and balance the feedback voltage V _f . In the equilibrium state, V ₂ = V ₃ and the following equation is obtained.

V ₃ =T _A −T _D /T _A +T _B xV _Z (9) Transforming, V ₃ =T _A /T _A +T _B V _Z −T _D /T _A +T _B V _Z (9)′ Here , the first term is C ₁ V _Z / (C ₁ +
C ₂ ), and by substituting equations (1) and (2) into this, it becomes (1+k ₁ ΔP)V _Z /2. In the second paragraph (6),
Using equation (7), we get the following equation.

V ₃ =1+k ₁ ΔP/2・V _Z −V _c3 −V _f ／Ki ₂・C ₄ V _Z (10) The feedback voltage V _f is V _f =−β(R ₈ I _L −V ₁ ) (11) Therefore, by substituting this into equation (10) and rearranging it, I _L =V ₁ /R ₈ +1/β(V _c3 /R ₈ +Ki ₂ /2R ₈ C ₄ −V ₃ Ki ₂ /R ₈ C ₄ V _Z ) + k ₁ Ki ₂ /2R ₈ C ₄・Δp/β (12). The first term in equation (12) sets V ₁ /R ₈ to 4 mA when I _L = 4 to 20 mA, and the second term adjusts the magnitude of V ₃ so that this term becomes zero. do. The third term is a term that provides a transmission current I _L proportional to the mechanical displacement ΔP of the movable electrode MP. The span can be adjusted by changing the partial pressure ratio β.

Note that when the temperature of the detection section PCC changes, the capacitance of the capacitors C ₁ and C ₂ changes, and the times T _A and T _B also change, causing an error _factor. was inserted, and this value also changed in the same way, so as shown in equation (7), the time
T _D changes as well. Therefore, the denominator and numerator of equation (9)' change in the same way, so no span error occurs due to temperature.

According to the embodiment of FIG. 4, the inverter G ₂ ,
Since the elements such as the NOR gate G ₁ , the D-type flip-flop DFF, the switch SW ₃ , and the comparator Q ₇ are digital elements, they can be easily integrated into an IC, and all these elements can be configured with one chip. In particular, when using a gate array (300 gates, 800 gates, etc.), which is a semi-custom IC, a one-chip IC can reduce the resistance by one order of magnitude or less compared to the precision resistors R ₂ and R ₃ of the conventional technology shown in Figure 1. This can be achieved at low cost. Note that an inexpensive resistor _R10 in FIG. 4 can be used.

<Effects of the Invention> As described above in detail with the embodiments, according to the present invention, the output from the sensor circuit section is not processed only by the analog circuit as in the past, but is processed by the duty ratio conversion circuit and the pulse calculation circuit. Digital IC such as
Since it is decided to use elements that are easy to manufacture and low in cost, it is possible to realize a two-wire transmission circuit that is lower in cost than the conventional one as a whole.

[Brief explanation of drawings]

Figure 1 is a block diagram showing a conventional embodiment, Figure 2 is a block diagram showing a conventional embodiment.
The figure is a block diagram showing one embodiment of the sensor circuit section according to the present invention, FIG. 3 is a waveform diagram showing waveforms of each part of the embodiment of FIG. 2, and FIG. The block diagram shown in Fig. 5 is the 4th block diagram.
FIG. 3 is a waveform diagram showing waveforms at various parts of the illustrated embodiment. SC1, SC2...sensor circuit section, PCC...detection section, CDC...capacitance duty ratio conversion section, CC1 to CC
3... Constant current circuit, Q ₃ , Q ₄ , Q ₇ ... Comparator, S ₀
~S ₃ ...Pulse signal, I _L ...Transmission current, V _Z ...Constant voltage, V _f ...Feedback voltage.

Claims (1)

[Claims] 1. In a two-wire transmission circuit that supplies a transmission current to a receiving section via a two-wire transmission path, a sensor circuit section that outputs a first pulse signal whose duty ratio changes depending on a physical quantity to be measured; a duty ratio conversion circuit that outputs a second pulse signal whose duty ratio changes according to a voltage corresponding to a transmission current; and a pulse width that is a difference between the pulse width of the first pulse signal and the pulse width of the second pulse signal. a pulse arithmetic circuit that outputs a third pulse signal having a pulse arithmetic circuit; a smoothing circuit that averages the third pulse signal from the pulse arithmetic circuit; and a control section that controls the transmission current based on the output of the smoothing circuit. A two-wire transmission circuit featuring: