JPH0360376A - Electric power source device - Google Patents

Abstract

PURPOSE:To increase the degree of freedom in control while maintaining the balance between input and output power by intermittently stopping a switching element capable of input control, corresponding to the power source polarity detected by a power source polarity detecting means. CONSTITUTION:First oscillation output out 1 is input in an AND gate G1, and the output of an AND gate G1 becomes the trigger input of a monostable multivibrator MV1. The output of the monostable multivibrator MV1 becomes the trigger input of a monostable multivibrator MV2. The monostable multivibrator MV2 outputs a pulse signal of pulse width capable of containing one output pulse of an oscillation circuit 10. This pulse signal is input together with a power source polarity detection signal into an AND gate G2, and the output of the AND gate G2 is inverted for logic, and is input together with the first oscillation out 1 into an AND gate G4.

Description

DETAILED DESCRIPTION OF THE INVENTION [Field of Industrial Application] The present invention relates to a power supply device that converts alternating current power into direct current power and switches this direct current power to supply power to a load.

[Prior Art] Figure 16 shows a conventional power supply device (Patent Application No. 6446 of 1999).
5) is a circuit diagram. The circuit configuration will be explained below. This circuit includes transistors Q 1 and Q 2 made of power MOSFETs as switching elements. Transistor Q.

The source of is connected to the train of transistor Q2. Transistors Q and Q2 have diodes D, D
, are connected in antiparallel. This diode D...
D2 can be replaced by a reverse diode parasitic between the drain and source of the power MO3FET. The train of transistor Q is connected to the cathode of diode D, - the anode of diode D3 is connected to the cathode of diode D4, and the anode of diode D4 is connected to the source of transistor Q2. Transistor Q1
One end of the capacitor C3 is connected to the drain of the capacitor C3, the other end of the capacitor C8 is connected to one end of the capacitor C1, and the other end of the capacitor C4 is connected to the source of the transistor Q2. Transistor Q 1. A load circuit 2 is connected between the connection point of Q 2 and the connection point of capacitors Cs and C<. Transistor Q 1. Q
The connection point 2 is connected to one end of the AC power supply Vs. The other end of the AC power supply Vs is connected to a diode D2. through inductors L+ and L2. It is connected to the connection point of D. Connection point of inductors L, L, and AC power supply Vs
A capacitor C1 is connected between one end of the capacitor C1 and one end of the capacitor C1.

Inductor L1 and capacitor C7 constitute an AC filter.

A square wave drive signal S is input between the gate and source of the transistor Q, and when the drive signal SI is at a high level, the drive signal S becomes low level, and the drive signal S is input between the gate and source of the transistor Q2. A rectangular wave drive signal S that becomes high level when S is low level is input. As a result, transistors Q, , Q, are turned on and off alternately.

First, when the transistor Q1 is turned on during a positive half cycle of the AC power supply Vs, a current flows from the AC power supply Vs to the inductor L2 through a path passing through the inductor L2, the diode D1, and the transistor Q, and the current value is The voltage increases at a slope proportional to the instantaneous value of the voltage Vin. At this time, the transistor Q also functions as a switching element for the inverter, and the voltage is increased from the capacitor C3 to the transistor Q.
, to cause current to flow through the load circuit Z.

Next, when transistor Q is turned off, inductor L2
, diode D3. The energy of the inductor L2 is released through the path passing through the capacitor C3, the load circuit Z, and the AC power supply Vs, as well as through the inductor L2, diode D1, capacitors C3, C4, diode D2, and the AC power supply Vs. to charge. At this time, transistor Q2 is on and capacitor C
A current flows through the load circuit 2 in the opposite direction to the above through a path that passes through the load circuit Z and the transistor Q.

In this way, during the positive half cycle of the AC power supply Vs, the transistor Q serves as both the switching element for the chopper and the switching element for the inverter, and the transistor Q2
functions only as a switching element for the inverter.

Next, when the transistor Q2 is turned on during the negative half cycle of the AC power supply Vs, the path passes through the AC power supply Vs, the transistor Q2, the diode D1, and the inductor L2.
A current flows through the inductor L2, and the current value increases at a slope proportional to the instantaneous value of the input AC voltage Vin.At this time, the transistor. 2 also functions as a switching element for the inverter, and the capacitor C4 also connects the load circuit 2,
Current flows through the load circuit Z through a path passing through the transistor Q2.

Next, when the transistor Q2 turns off, the AC power supply Vs,
A path passing through load circuit Z, capacitor c4, diode D4, and inductor L2, as well as AC power supply Vs, diode D, . In the path through C, , C, diode D4, and inductor L2, the energy in inductor L2 is released and charges capacitors C1 and C1. At this time, transistor Q is on, and capacitor C
5 to the load circuit Z through the transistor Q in the opposite direction to the above.

In this manner, during the negative half cycle of the AC power supply Vs, the transistor Q2 functions both as a chopper switching element and an inverter switching element, and the transistor Q functions only as an inverter switching element.

Therefore, in the above circuit, the switching element for the inverter also serves as the switching element for the chopper, and is configured with a small number of elements, which has the advantage of reducing power loss and simplifying the circuit configuration. Further, in the above circuit, each transistor Q is activated every half cycle of the AC power supply Vs.

Since Q2 alternately works as a switching element for the chopper and an inverter, there is an advantage that the stress per switching element side is reduced, and the power loss of the switching element (transistor QQ,) is balanced. Therefore, for example, the heat dissipation structure may be the same, and the switching element (transistor Q7.
2) operates as a switching element for the chopper and the inverter, so there is no need to provide a separate chopper drive circuit, and the configuration of the drive circuit is also simplified.

Note that the input current distortion factor can be reduced by inserting an AC filter consisting of an inductor L and a capacitor C1 between the AC power supply Vs and the inductor L2 to make the input current Iin continuous. Since the input current Iin can be made into a sine wave having the same phase as the input voltage Vin, the input power factor becomes approximately 1.

In the above circuit, as the load circuit 2, for example, the first
A circuit as shown in FIG. 7 is used. This circuit is a discharge lamp lighting circuit. For example, an inductor for current limiting and resonance is connected in series to the power supply side terminal of a hot cathode discharge lamp 1a such as a fluorescent lamp, and resonance and preheating are performed between the non-power supply side terminals. A capacitor C2 for current supply is connected in parallel. When the discharge lamp starts, a preheating current flows to the filament via capacitor C2, and the inductor and capacitor C
A high voltage is generated across the discharge lamp 1a due to the series resonance effect of the discharge lamp 1a, and the discharge lamp 1a is lit.

FIG. 18 is an operational waveform diagram of the above circuit. Load current IZ
is an inductor, and a sinusoidal current is generated due to the resonance effect of the capacitor C2. In addition, the chopper current TCH becomes a triangular wave-like high-frequency current, but the input current Iin has its high-frequency components removed by an AC filter consisting of an inductor L and a capacitor C1, and becomes a sine wave similar to the input voltage Vin, and the input current Iin becomes a sinusoidal waveform similar to the input voltage Vin. The input current harmonic distortion is improved and input current harmonic distortion is reduced.

FIGS. 19 and 20 are circuit diagrams respectively showing the configurations of an oscillation circuit and a drive circuit used in the above circuit. First, the first
The oscillation circuit shown in Figure 9 consists of a control IC for a switching regulator (μPC494C manufactured by NEC Corporation).As is well known, this control IC is connected to a power supply terminal (
A control power supply Vcc is applied between the capacitor terminal (bin No. 12) and the ground terminal (bin no. 7), and a capacitor C6 is connected between the capacitor terminal (bin no. 5) and the ground terminal, ) and the ground terminal
It has a built-in oscillator that oscillates at a frequency according to the time constant of 3 and R. The first oscillation output is obtained by alternately shorting and opening the first open collector terminal (pin 11) and the first open emitter terminal (bin 10). , the second oscillation output is
This is obtained by alternately shorting and opening the second open collector terminal (bin 8) and the second open emitter terminal (bin 9).Here, the output control terminal ( When pin 13 is dropped to the ground level, single-ended operation for one stone is performed, and the first oscillation output matches the second oscillation output, but the output control terminal is connected to the reference voltage output terminal. When set to the level of the reference voltage Vref obtained in (bin 14),
A push-pull operation for two stones is performed, and the first oscillation output and the second oscillation output take opposite states after a predetermined dead-off time.

This dead-off time is determined by the dead-off time control terminal (
It is determined by the voltage of the 4th bin), but in this circuit, this terminal is connected to the ground terminal to make the dead-off time zero. Note that the non-inverting input terminals (pin 1, pin 16) and the inverting input terminals (pin 2, pin 15) are the input terminals of the comparator for pulse width control, and are used when pulse width control is not performed. The former is pulled down (or opened) to the ground level GND, and the latter is pulled up to the level of the control power supply Vce.Furthermore, the feedback terminal (bin 3) is a feedback input terminal for pulse width control. Since it is not used in the above circuit, it is connected to the ground terminal.

The first open emitter terminal (bin No. 10) of this oscillation circuit is grounded via a resistor R1, and the second open emitter terminal (bin No. 9) is grounded via a resistor R2. Therefore, when the first open emitter terminal (bin No. 10) and the first open collector terminal (pin No. 11) are short-circuited, the first oscillation output "outl" becomes "High" as compared to the control power supply voltage Vcc. level.

At this time, the second open emitter terminal (pin 9) and the second open collector terminal (pin 8) are open, and the second oscillation output "out2" is pulled down by the resistor R2 to "L". os” level. Furthermore, when the first open emitter terminal (pin 10) and the first open collector terminal (bin 11) are opened, the first oscillation output "outl" is pulled down by the resistor R1 and becomes "L os “It becomes a level. At this time, the second open emitter terminal (pin 9) and the second open collector terminal (pin 8) are short-circuited, and the second open emitter terminal (pin 9) is short-circuited.
The oscillation output "out2" is determined by the control power supply voltage Vcc.
It becomes "High" level.

Next, the drive circuit shown in FIG. 20 will be explained. 1st
The oscillation output "outl" is applied to the series circuit of the primary winding of the pulse transformer PT and the capacitor C6 through a totem pole circuit made up of bipolar transistors Q, , Q. The secondary winding of the pulse transformer PT is connected between the gate and source of the transistor Q via a forward bias resistor R6 and a reverse bias diode D.

Between the gate and source of transistor Q, there is an inverse series circuit of Zener diodes ZD, ZD2 and a resistor R.

are connected in parallel. In addition, the second oscillation output out2
" is supplied between the gate and source of the transistor Q2 via a totem pole circuit consisting of bipolar transistors Q, , Q@ and a bias resistor R2.

A resistor R3 is connected in parallel between the gate and source of the transistor Q2.

When the first oscillation output "out 1" reaches the "High" level, the transistor Qs is turned on and the transistor Q6 is turned off, and the control power supply voltage Vcc turns the transistor Q
, current flows through the primary winding of the pulse transformer PT and the capacitor C6, and the capacitor C6 is charged. At this time, the voltage generated in the secondary winding of the pulse transformer PT charges the gate-source capacitance of the transistor Q via the resistor R1, and when the gate potential becomes higher than the threshold voltage, the transistor Q1 is turned on. Then, when the first oscillation output "outl" goes to the "Low" level, the transistor Q is turned off, and the transistor Q6 is turned on by the charging voltage of the capacitor C6.

As a result, a current flows from the capacitor C6 through the primary winding of the pulse transformer PT and the transistor Q6, and a voltage having a polarity opposite to that described above is generated in the secondary winding of the pulse transformer PT. As a result, the accumulated charge in the gate-source capacitance of transistor Q is extracted through diode D7, and when the gate potential becomes lower than the threshold voltage, transistor Q is turned off. Note that a Zener diode ZD is connected between the gate and source of the transistor Q.
, ZD2 and the resistor R6 are connected in parallel, it is possible to prevent the gate-source voltage of the transistor Q from becoming extremely high due to the induced voltage of the pulse transformer PT.

Next, when the second oscillation output "out2" becomes "High" level, the transistor Q is turned on and the transistor Q is turned on.
I is turned off, and the gate of transistor Q2 is turned off by control power supply voltage Vcc through transistor Q and resistor R2.
When the source-to-source capacitance is charged and its gate potential becomes higher than the threshold voltage, transistor Q2 is turned on. Next, the second oscillation output “out 2” becomes “Low”.
” level, the transistor Q7 is turned off, and the transistor Q is turned on by the charging voltage of the gate-source capacitance of the transistor Q2.As a result, the accumulated charge in the gate-source capacitance of the transistor Q2 is transferred to the resistor R.
,,,R.

When the gate potential becomes lower than the threshold voltage, transistor Q2 is turned off.

[Problems to be Solved by the Invention] In the above-mentioned conventional example, since the inverter circuit and the chopper circuit share the same switching element, when the control state of the inverter circuit is changed to control the output power,
This also affects the chopper circuit, and conversely, if you change the control state of the chopper circuit for input power control, it will also affect the inverter circuit, causing the problem that the degree of freedom in control is small. Ta. Therefore, the present inventor would like to propose that the inverter circuit be controlled mainly by frequency control, and the chopper circuit be controlled mainly by on-duty control.
See No. 34397.

However, for example, if the switching frequency of the transistors Q, Q2 is increased to dim the discharge lamp 1a,
The power consumption in the load circuit Z due to the operation of the inverter circuit decreases, but the AC power consumption due to the operation of the chopper circuit decreases.
The power supply from s does not change much, so the power supply from the AC power supply Vs becomes excessive compared to the power consumption in the load circuit Z, and all this excess power is stored in the capacitor C5°C1. Ru. Therefore, capacitors C,,C
The charging voltage VDC of C4 increases, and capacitors C and C4 are destroyed due to overvoltage resistance. Also, transistor QQ2
Circuit elements such as diodes D1 and D2 may also be destroyed due to overvoltage breakdown. Furthermore, a high voltage will be generated before the element is destroyed, which is not desirable from a safety standpoint. Therefore, the switching frequency cannot be made too high during dimming, and the degree of freedom in control is reduced. Further, even when compensation for fluctuations in input power and output power is performed by controlling switching elements, there is a problem that the degree of freedom of control is small, so the fluctuation compensation range becomes small and the stability of the power supply device deteriorates.

The present invention has been made in view of these points, and its object is to provide a power supply device in which a chopper circuit and an inverter circuit also serve as switching elements.
The goal is to increase the degree of freedom in control while maintaining a balance between input power and output power.

[Means for Solving the Problems] In the present invention, in order to solve the above problems, the first
As shown in the figure, the alternating current voltage of the alternating current power source Vs is chopped at high frequency by a switching element (transistors Q, ,Q,) and applied to the inductor L2, and the energy stored in the inductor L2 is transferred to the capacitor Cs,
A chopper circuit that charges C4 to obtain a DC power supply, and an inverter circuit that receives the DC voltage of the DC power supply as input and supplies power to the load circuit 2 through on/off operations of first and second switching elements connected in series. The first switching element (transistor Q) is connected to an AC power source Vs.
It is also used as a switching element of the chopper circuit when one power supply polarity (V in > O) of the AC power supply Vs, and the second switching element (transistor Q2) is activated when the other power supply polarity (V in < O) of the AC power supply Vs. In a power supply device that is also used as a switching element of a chopper circuit, a power supply polarity detection means that detects the power supply polarity of an AC power supply Vs, and a detection output of the power supply polarity detection means that detects the power supply polarity of the AC power supply Vs, The present invention is characterized in that it includes a control means for intermittently stopping the operation of the switching element.

[Operation] In the circuit shown in FIG. 1, there is a one-to-one correspondence between the power supply polarity of the AC power supply Vs and the switching element that controls the input chopper. That is, the number of switching elements whose input can be controlled is one for each power supply polarity of the AC power supply Vs. The power supply polarity of AC power supply Vs is positive (V in>
0), transistor Q1 can control the input, but transistor Q cannot control the input.
On the contrary, when the power supply polarity of the AC power supply Vs is negative (Vin<O), the input of the transistor Q2 can be controlled, but the input of the transistor Q cannot be controlled. Therefore, the input-controllable switching element is intermittently stopped in accordance with the power supply polarity detected by the power supply polarity detection means. Specifically, as shown in the operating waveform diagram in Figure 2,
When the power supply polarity of the AC power supply Vs is positive (V in > O), the operation of the transistor Q1 is stopped intermittently.
On the other hand, when the power supply polarity of the AC power supply Vs is negative (Vin<O), the operation of the transistor Q2 is intermittently stopped. In this way, by intermittently stopping the operation of the input-controllable switching elements, it is possible to freely reduce the power supply from the AC power supply Vs, and due to excessive power supply, the voltage of the capacitors C,,C, It is possible to prevent an increase in VOC.

[Example 1] FIG. 1 is a circuit diagram of a first embodiment of the present invention.

In this circuit, the discharge lamp lighting circuit shown in FIG. 17 is connected as the load circuit 2 of the half-bridge circuit shown in FIG. 16. In this embodiment, as shown in the operating waveform diagram of FIG. When the power supply polarity of the AC power supply Vs is positive (Vin>O), the drive signal Sl of the transistor Q1 is intermittently stopped, and the power supply polarity of the AC power supply Vs is negative (Vin<O).
At this time, the drive signal S2 of the transistor Q2 is intermittently stopped.

FIG. 3 shows the i-precepts of the control circuit for realizing the above operation. In FIG. 1, the configuration of the oscillation circuit 10 is the same as the circuit shown in FIG. This embodiment, which is the same as the circuit shown in the figure, is characterized in that an intermittent drive circuit 11 is inserted between an oscillation circuit 10 and a drive circuit 12.

The configuration of this intermittent drive circuit 11 is shown in FIG.
In the figure, VIN indicates that the power supply polarity of the AC power supply Vs is positive (V in
> O), “High” level, negative (Vin<O)
), this signal becomes the "L am" level, and is the output of the power supply polarity detection means. The power supply polarity detection means may be any circuit that can generate the above-mentioned signal VIN, and its configuration is not particularly limited. The output CK1 of the gate G becomes the trigger for the monostable multivibrator MVI. Therefore, the monostable multivibrator MVl has a positive power supply polarity (Vin>O
), it is triggered at the falling edge of the first oscillation output "outl".

When the monostable multivibrator MVI is triggered,
A pulse signal P1 that remains at a "High" level for a predetermined period is output. The pulse width of this pulse signal P1 is set to several times the switching period.

The output of the monostable multivibrator MVI is used as a trigger force for the monostable multivibrator MV2. The monostable multivibrator MV2 outputs a pulse signal P2 having a pulse width that can include one output pulse of the oscillation circuit 10. This pulse signal P2 is input to the AND gate G2 together with the power supply polarity detection signal vIN, and the output of the AND gate G2 has its logic inverted by the NOT circuit and is input to the AND gate G together with the first oscillation output "outl". Ru.

The signal output from AND gate G is transmitted to transistor Q.
, the oscillation output for driving becomes “out” 1”.

FIG. 6 shows operating waveforms of each part of the circuit. As shown in the figure, when the power supply polarity is positive (Vin>O), the monostable multivibrator MV2 is triggered at a period determined by the monostable multivibrator MVI, and outputs the first oscillation output "outl". A signal P2 for cutting one pulse is generated, and the first oscillation output “ou
t1'' pulses are thinned out.The power supply polarity is negative (V i
When n < O), the power supply polarity detection signal VIN is set to NO.
A monostable multivibrator M can be created by applying the logic-inverted signal by the T circuit circuits s and G to the circuit with the same configuration as above, along with the second oscillation output "out2".
At a period determined by V3, the monostable multivibrator MV4 is triggered to generate the second oscillation output “out
A signal P4 is generated to remove one pulse from 2'' at a constant cycle.

The pulses of the second oscillation output "out2" are thinned out.

FIG. 5 shows the configuration of monostable multivibrators MVI to MV4. This circuit includes a general-purpose timer IC (Signetics, 11NE555). When the input falls, the resistors R,,,R,.

timer I via a differentiating circuit consisting of capacitor C0 and
The voltage at the trigger terminal 1 (bin 2) of C falls. In this timer IC, the trigger terminal (bin 2) is (1/
3) When the voltage drops below Vcc, it is triggered and the output terminal (3
pin No. 7) becomes “HiF!h” level, and the discharge terminal (pin No. 7) becomes “HiF!h” level.
pin) becomes high impedance. Also, when the threshold terminal (bin 6) becomes (2/3) Vcc, the output terminal (bin 3) goes to "Low" level, and the discharge terminal (bin 7) also goes to "Low" level.The power supply terminal (
A control power supply voltage Vcc is applied between the ground terminal (pin No. 8) and the ground terminal (bin No. 1). In addition, the reset terminal (
Bin No. 4) is connected to the power terminal (Bin No. 8),
The frequency control terminal (bin No. 5) is connected to the ground terminal (bin No. 1) via a decoupling capacitor C1.

Control power supply voltage Vcc is applied to a series circuit of resistor R+3 and capacitor CI□, which constitutes a time constant circuit of the timer IC. The connection point between resistor R1 and capacitor CI2 is connected to the threshold terminal (bin 6) and discharge terminal (bin 7) of the timer IC, so that the timer IC operates as a monostable multivibrator. In other words, when the trigger terminal (bin 2) goes low and is triggered, the time constant of the capacitor Cat and resistor RI3 causes the threshold terminal (bin 6
No. bin> has a predetermined threshold voltage V th= (2/
3) The output terminal (bin 3) is “
During this time, even if the voltage at the input terminal (bin 2) fluctuates, the output terminal (pin 3) remains at the "high" level until the voltage of capacitor C1□ reaches the threshold voltage vth. do.

In addition, in the monostable multivibrators MVI and MV3, the time constant of resistor R13 and capacitor chi is set large, and in the monostable multivibrator MV, MV, the above time constant is set small. Output signals P1 to P4 having different pulse widths as shown in FIG. 1 can be obtained.

With the above control circuit, in this embodiment, when the power supply polarity of the AC power supply Vs is positive (Vin>O), the drive signal S1 of the transistor Q1 is intermittently stopped, and the AC power supply Vs
When the power supply polarity is negative (Vin<O), the drive signal S2 of the transistor Q2 can be stopped intermittently. If the drive signal SI of the transistor Q1 is stopped when the power supply polarity is positive (V in > O), the operation of the chopper circuit is stopped, and therefore the power supply from the AC power supply Vs is stopped. On the other hand, regarding the inverter circuit, the power consumption in the load circuit Z is reduced by stopping the transistor Q, but since the transistor Q2 is operating, the power consumption is only halved. Vi
The same applies when n<O). Therefore, the load circuit Z
If the power supply from the AC power supply Vs is excessive compared to the power consumption at the
, it is possible to prevent the voltage VDC from increasing abnormally. In addition, the cycle for intermittently stopping the drive signals S, , S, can be freely controlled by the monostable multivibrators MVI and MV3, and the period for stopping the drive signals S, S, can be controlled freely by the monostable multivibrator M
V2. Since it can be freely controlled by MV4, by controlling these, it is possible to control the power supply from the AC power supply Vs over a fairly wide range, and in this case, the change in power consumption in the load circuit Z is small. For example, when combined with switching frequency control, it becomes possible to substantially independently control the power supply (input power) from the AC power supply Vs and the power consumption (output power) in the load circuit Z. In other words, when the switching frequency is changed, the change in output power is large, but the change in input power is small, so by combining it with the control method of the present invention, input power and output power can be controlled separately.

Further, the drive signal S1. If the periods in which S2 is intermittently stopped are evenly spaced over a half cycle of the AC power source Vs, the input current Iin will be similar to the input voltage Vin as shown in FIG. 2, and the input current harmonics will be does not increase much, and the input power factor can also be increased.

Note that in FIG. 2, for convenience of illustration, transistors Q1.
Although the period of the drive signal S, , S, of Q2 is expressed as long, it goes without saying that the period is actually much shorter than shown.

[Example 2] FIG. 7 is a circuit diagram of a second embodiment of the present invention.

In the present invention, a full bridge circuit is used as the inverter circuit. That is, in the half-bridge circuit shown in FIG. 1, transistor Q is used instead of capacitor C1, and transistor Q4 is used instead of capacitor C1.
Connect the transistor Q and the transistor Q
2 is turned on and off at the same time, and transistor Q is turned on and off at the same time as transistor Q1. Transistor Q.

Diodes D, , D, are connected in antiparallel to Q4. These diodes Ds and Da are power MO
The drains of transistors Q, Q, consisting of 3FETs
It can be replaced by a reverse diode parasitic between the sources. Also, transistor Q3. Q.

A smoothing capacitor C0 is connected in parallel to both ends of the series circuit. The drive signal S3 of the transistor Q is the same as the drive signal S2 of the transistor Q2, and the drive signal S4 of the transistor Q4 is the same as the drive signal S of the transistor Q.
, is the same as .

The operation of this embodiment is the same as that of the circuit shown in FIG. 1, except that the voltage applied to the load circuit is approximately twice that of the circuit shown in FIG.

[Example 3] FIG. 8 is a circuit diagram of a third embodiment of the present invention.

In this embodiment, in the half-bridge circuit shown in FIG. 1, power MO
A bipolar transistor is used instead of the 8FET. In this case, anti-parallel diodes D I, D
2 needs to be connected as a separate part.

FIG. 9 is an operational waveform diagram of this embodiment. First, the operating waveform in the fully lit state is the same as the operating waveform shown in FIG. 18. In this fully lit state, the switching frequency is f, and the period of the drive signals S+ and St is T I =1
/r+, then the switching frequency is f, to f
2 and the period of the drive signals S + and S 2 is changed to 72=1/f2, the discharge lamp 1a enters the dimming state. That is, as the switching frequency increases, the current limiting impedance due to the inductor increases, so the amplitude of the lamp current Ila decreases, and the light output of the discharge lamp 1a decreases. This reduces the power consumption by the load circuit, but as mentioned above, the power supply by the chopper circuit is not affected much by changes in the switching frequency, so the power supply becomes excessive compared to the power consumption, and the capacitors C, , C , the voltage Voc will rise. Therefore, in this embodiment, when the power supply polarity is Vin>O, the drive signal S of the transistor Q that controls the chopper circuit is intermittently stopped, and when the power supply polarity is Vin<O, the chopper circuit is controlled. The drive signal S2 of the transistor Q2 is intermittently stopped. Thereby, the power supply by the chopper circuit can be reduced, and the voltage VDC of the capacitors C, , C can be prevented from increasing abnormally.

Note that, as shown in the operating waveform diagram of FIG. 10, the drive signal S
1. If the stop period of S2 is further increased, the power supply by the chopper circuit can be further reduced, and if the power consumption of the load circuit is correspondingly reduced, the discharge lamp 1a can be dimmed to a deeper level. be able to. However, if the stop period of the drive signal S82 is too long, the stop period of the chopper current ICH will become longer, the input current harmonic components will increase, and the input power factor will also decrease.
It is necessary to set the dimming range in consideration of the filter design.

[Embodiment 4] FIG. 11 is a circuit diagram of the fourth embodiment of the present invention.
The figure shows its operating waveform diagram. In this embodiment, an incandescent lamp is used as the load circuit and does not include a resonant circuit, so the waveform of the lamp current ea is a rectangular wave.

Also, even if the switching frequency is changed, the lamp current 11a
does not change, so dimming cannot be done by frequency control.
Therefore, in this embodiment, when the power supply polarity is Vin>O, the drive signal S is intermittently stopped, and when Vin<O, the drive signal S2 is intermittently stopped, thereby reducing the power supply by the chopper circuit. At the same time, the power consumption by the load circuit is also reduced to dim the incandescent light bulb 1a. In this case, the decrease in power supply from AC power supply Vs is
This is more noticeable than the reduction in power consumption in the load circuit, so the voltage .Dc across capacitors C3 and C4 decreases. This reduces power consumption in the load circuit and reduces the light output of the incandescent bulb 1a.

[Embodiment 5] FIG. 13 is a circuit diagram of the fifth embodiment of the present invention.
The figure is an operating waveform diagram. In this embodiment, when the input voltage from the AC power supply Vs is the rated value, only the drive signal Sl is generated when the power supply polarity is Vin>O.
When Vin<O, only the drive signal S2 is generated,
The polarity of the voltage applied to the incandescent light bulb 42a is alternately reversed every half cycle of the AC power source Vs. In this case, the effective value of the lamp current 11a is half of the peak value.

Next, when the input voltage from the AC power supply Vs increases due to power supply fluctuations, the drive signal SI is intermittently stopped when the power supply polarity is Vin>O, and the drive signal S2 is intermittently stopped when the power supply polarity is Vin<0. It has been stopped. Therefore, even if the input voltage increases, the power supply by the chopper circuit does not change, and when the power supply polarity is Vin>O, the drive signal S
, the drive signal S2 is generated immediately after the intermittently stops the drive signal S2, and the drive signal S1 is generated immediately after the drive signal S2 is intermittently stopped when the power supply polarity is V in < O. There is. Therefore, the effective value of the lamp current Ifa is half of the peak value, and the power consumption by the load circuit does not change. Therefore, in this embodiment, even if the input voltage from the AC power supply Vs rises above the rated value due to power supply fluctuations, , capacitor Cs, voltage V of C4. 0 or the light output of the incandescent light bulb 1a can be controlled so as not to fluctuate.

In the above embodiment, as shown in FIG. 15(a), the inductor L2 is connected to the connection point of the diodes D3 and D4, but as shown in FIG. 15(b). As shown, transistor Q1. An inductor L2 may be connected to the connection point of Qz. Also, FIG. 15 (C).

As shown in (d), the inductor L may be connected to one end of the AC power source Vs, and the inductor L2 may be connected to the other end of the AC power source Vs.

In each of the above-mentioned embodiments, the inverter circuit is a half-bridge circuit or a full-bridge circuit, but a modified half-bridge circuit or other circuit may also be used. , a circuit in which the negative electrode of capacitor C1 in the circuit of FIG. 1 is connected to the anode of diode D instead of the positive electrode of capacitor C1, or the positive electrode of capacitor C4 is connected to the cathode of diode D instead of the negative electrode of capacitor C3. , and can perform substantially the same operation as a half-bridge circuit. In this modified half-bridge circuit, the diode D
3. The capacitor connected in parallel to the DC circuit D serves as a smoothing capacitor for the DC power supply, and the capacitor connected in series with the load circuit Z serves as a coupling capacitor for cutting DC components and for the power supply.

[Effects of the Invention] According to the present invention, as described above, the DC power obtained from the AC power through the chopper circuit is input, and the on/off operation of the first and second switching elements connected in series is performed. In a power supply device that supplies power to a load circuit according to the power supply polarity of the AC power supply, the first or second switching element is alternately used as the switching element of the chopper circuit according to the power supply polarity of the AC power supply. hand,
Since the operation of the switching element that is also used as the switching element of the chopper circuit is stopped intermittently, the power supplied from the AC power source by the chopper circuit can be freely reduced, and therefore, the power supply from the AC power source can be reduced. The power supply and the power consumption by the load circuit can be controlled separately, which has the effect of increasing the degree of freedom in control.

[Brief explanation of drawings]

FIG. 1 is a circuit diagram of the first embodiment of the present invention, FIG. 2 is an operation waveform diagram of the same, and FIG. 3 is a circuit diagram of a control circuit used in the same.
Figure 4 is a circuit diagram of the intermittent drive circuit used in the above, Figure 5 is a circuit diagram of a monostable multivibrator used in the same.
The figure is an operation waveform diagram of the intermittent drive circuit same as above, FIG. 7 is a circuit diagram of the second embodiment of the present invention, FIG. 8 is a circuit diagram of the third embodiment of the present invention, and FIGS. 9 and 10 are Same as above operating waveform diagram, 1st
1 is a circuit diagram of a fourth embodiment of the present invention, FIG. 12 is an operation waveform diagram of the same as above, and FIG. 13 is a circuit diagram of a fifth embodiment of the present invention.
FIG. 14 is an operation waveform diagram of the same as above, and FIGS. 15(a) to (c)
l) is a circuit diagram illustrating a power supply device to which the present invention can be applied;
Fig. 16 is a circuit diagram of the conventional example, Fig. 17 is a circuit diagram of a load circuit used in the above, Fig. 18 is an operation waveform diagram of the same as the above, and Fig. 19 is a circuit diagram of a load circuit used in the same.
The figure is a circuit diagram of an oscillation circuit used in the above, and FIG. 20 is a circuit diagram of a drive circuit used in the same. Vs is an AC power supply, L2 is an inductor, C, , C, is a capacitor, Q, , Q2 is a transistor, 2 is a load circuit, 1
1 is an intermittent drive circuit.

Claims (1)

[Claims] (1) A chopper circuit that chops the AC voltage of an AC power supply at high frequency using a switching element and applies it to an inductor, and charges a capacitor with the energy stored in the inductor to obtain DC power; and a DC voltage of the DC power supply. and an inverter circuit that supplies power to a load circuit through the on/off operations of first and second switching elements connected in series, the first switching element having a In a power supply device that is also used as a switching element of a chopper circuit, and a second switching element is also used as a switching element of the chopper circuit when the other switching element of the AC power supply has the other power supply polarity, a power supply polarity detection means for detecting the power supply polarity of the AC power supply. and a control means for intermittently stopping the operation of a switching element that also serves as a switching element of a chopper circuit, based on the detection output of the power supply polarity detection means.