JPH05191973A - Dc-dc converter - Google Patents

Abstract

PURPOSE:To make it possible to employ a low voltage low ON resistance switching element as a main switch by storing current energy, induced in the secondary winding when secondary current supply to a load is blocked, in a snubber circuit. CONSTITUTION:Upon turn OFF of a main switch 3, a voltage to be applied on the main switch 3 due to a resonant current flowing through a primary winding 6 rises with sinusoidal waveform. At the same time, electromotive force having the same period is induced in a secondary winding 7 in reverse direction. Upon turn OFF of a diode(DD) 11, all electric paths other than those passing through a coil 10 are interrupted from a smoothing circuit 12. When the electromotive force induced reversely in the secondary winding 7 exceeds a reverse bias being applied on the DD9 under that state, the DD9 is turned ON to conduct secondary resonant current forwardly. When the voltage of a capacitor 8 reaches a peak value, the reverse resonant current is blocked by the DD9 to stop secondary resonance. Consequently, the capacitor 8 is discharged through the coil 10 to the smoothing circuit 12.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a DC-DC converter, and more particularly, to improvement of power loss characteristics of a main switch of a voltage resonance type DC-DC forward converter, and multi-output DC
-Improving the device for adjusting the output voltage of a DC converter.

[0002]

2. Description of the Related Art As is well known, the turn-on loss of a DC-DC converter is zero volt switching (ZVS).
The method has been significantly improved by applying it to a voltage resonance type or current resonance type DC-DC converter. Especially, the voltage resonance type converter is used as a main switch.
Since the output capacitance of the SFET can be used as a part of the resonance capacitor, it is suitable for increasing the switching frequency. Therefore, much research is being conducted to obtain high efficiency at high frequencies by using a voltage resonance type DC-DC converter.

FIG. 12 is a circuit diagram of a conventional example of a one-stone voltage resonance DC-DC forward converter. The circuit includes a DC power source 1, a main transformer 5, a main switch 3, a capacitor 4 connected in parallel with the main switch 3, a voltage clamper 70, a rectifying diode 11, a current snubber circuit 71, an input capacitor 2, an output capacitor 12, and a main switch. It comprises a loop control circuit 14 for controlling the switching of No. 3 of FIG. In the figure, the coil 15 represents the leakage inductance of the main transformer and, if necessary, the inductance of an external coil.

The main switch 3 and the capacitor 4 are connected in series to the main transformer 5, and the main switch 3 is connected to the loop control circuit 1.
It is driven by control No. 4 and regulates the output of the converter. The capacitor 4 is the inductance L of the main transformer 5.
_A series resonance circuit is formed together with _P , and the resonance circuit outputs a voltage of zero volt when the main switch 3 is turned on.
Apply to. Thus the power loss at turn-on,
That is, the turn-on loss can be reduced.

The voltage clamper 70 resets the magnetic energy stored in the core of the main transformer 5 during the ON period of the main switch 3 to prevent the core from being magnetically saturated. The voltage clamper 70 includes a diode, a resistor and a capacitor connected in parallel with each other, and is connected in parallel with the primary winding 6 of the main transformer 5. The magnetic energy stored in the core is dissipated by the resistor in the voltage clamper 70 during the OFF period of the main switch 3, and the surge voltage at the turn-off is absorbed by the capacitor. The current snubber circuit 71 protects the diode 11 by shunting the surge current induced by the coil 15 when the secondary current of the main transformer 5 is turned off. This circuit consists of a resistor and a capacitor connected in series, the resistor dissipates the energy of the current and the capacitor smooths the current.

This voltage resonance type DC-DC forward converter operates as follows. First, when the main switch 3 is turned off, a resonance current starts to flow in the primary winding 6, and the voltage V _DS of the main switch 3 rises from 0 volt. When the voltage V _DS returns to 0 after half a cycle, the main switch 3 is turned on by the control of the loop control circuit 14. In this way turn-on losses are prevented. The length of the ON period is proportional to the set value of the output voltage V _O of the converter, and if there is a deviation from the set value, the length of the ON period is adjusted so as to compensate for the deviation.

FIG. 13 shows a multi-output single-stone voltage resonance type DC-D.
It is a block diagram of the conventional example of a C forward converter. For simplicity, only the main output circuit 72 and one auxiliary output circuit 73 are shown. Normally, the highest output is selected as the main output, and the selected main output is controlled by the voltage resonance switching of the main switch 3. The auxiliary output circuit 73 has its own output adjusting circuit so that the control of the main output does not affect the auxiliary output. The converter composed of the voltage resonance switching circuit and the main output circuit shown in FIG. 13 is the most ordinary converter (see, for example, the Institute of Electronics, Information and Communication Engineers, Vol. 90, No. 439, PE 90-69), and the auxiliary output circuit 73 is the main. The output current of the auxiliary secondary winding 77 of the transformer is rectified by the rectifier diode 11 and output. The output voltage V _AO is adjusted by the series dropper 76 independently of the main output circuit 72.

[0008]

In the conventional voltage resonance DC-DC converter shown in FIG. 12, in order to apply a sinusoidal resonance voltage to the main switch, the peak voltage is necessarily applied to the switch element. As a result of applying the voltage, the high peak voltage causes a voltage stress in the switch element. Although it is necessary to use a high voltage switch to prevent this voltage stress, the high voltage switch usually has a high on-resistance, which causes an on-resistance loss, which causes the efficiency of the converter to be reduced. Further, if the voltage clamper 70 and the current snubber circuit 71 are used to prevent the harmful effects due to the application of a high voltage, not only power loss occurs in these circuits, but also resonance occurs if these circuits are not carefully used. The voltage waveform itself is deformed, which also causes turn-on loss of the main switch.

A conventional multi-output type DC shown in FIG.
In the -DC converter, there is a problem that power loss is inevitably generated when the fluctuation of the input voltage of the auxiliary output circuit is compensated by using the series dropper. The change in the input voltage of the auxiliary output circuit is usually caused by the change of the load current of the main output circuit or the change of the input voltage to the main output circuit. It occurs when the frequency changes.

A first object of the present invention is to use a low-voltage, low-on-resistance main switch, and to reduce the power loss of a rectifier diode. A voltage resonance type DC-DC forward with a snubber circuit. To provide a converter.

A second object of the present invention is to provide a multi-output DC having an output circuit having an output voltage adjusting circuit with low power loss.
-To provide a DC converter.

[0012]

In order to achieve the above-mentioned first object, the voltage resonance type DC-DC forward converter of the present invention comprises a capacitor and a first diode connected in series. A series circuit connected to both ends of the secondary winding to form a resonance circuit that conducts a resonance current in the forward direction of the first diode together with the secondary winding; and a charge accumulated in the capacitor, A discharge means for discharging the first diode during the off period, and a forward current of the first diode is such that a secondary current is supplied from the secondary winding to the smoothing circuit via the second diode for rectification. The snubber means is set so as to prevent conduction of the resonance current during the period.

In order to achieve the above-mentioned second object, the multi-output DC-DC converter of the present invention has at least one of its output circuits a main transformer 2 of the DC-DC converter.
The switch means for switching the secondary current supplied from the secondary winding to the load and the switch means so as to take the first switch state in a predetermined phase relationship with the switching operation of the main switch of the DC-DC converter. A first switching control means for controlling the means, a current / voltage converting means for generating a voltage signal that changes the same as the secondary current, and a deviation signal corresponding to the deviation of the output voltage of the output circuit from the set value. The absolute value of the difference between the current voltage signal and the first voltage signal when the voltage signal when the comparing means for outputting and the switch means makes a transition to the first switching state is the first voltage signal. When the absolute value signal exceeds a threshold determined as a decreasing function of the deviation signal, the switch means is controlled so as to assume the second switching state. And an output voltage regulating circuit and a second switching control unit.

[0014]

The operation of the voltage resonance type DC-DC forward converter of the present invention can be explained by dividing it into three periods. First, after the main switch is turned off, the second
During the period until the forward voltage of the diode becomes 0 and the supply of the secondary current to the load is stopped (period I), the voltage applied to the first diode becomes the forward direction and the resonance circuit becomes conductive. , The capacitor is charged by the forward resonance current of the first diode, the forward voltage of the first diode decreases as the charging voltage of the capacitor increases, and finally the period until the first diode turns off ( Period II),
During the period after the turning off of the first diode until the end of the off period of the main switch, the charge accumulated in the capacitor is discharged through the discharging means during this period (period III).

In the following description, the resonance circuits formed on the primary side and the secondary side of the main transformer are referred to as a primary resonance circuit and a secondary resonance circuit, respectively, and the resonance currents flowing in the respective resonance circuits are primary. It is described as a resonance current and a secondary resonance current. Since the resonance in the secondary resonance circuit is blocked in the period I, the voltage applied to the main switch rises with a time constant corresponding to the resonance cycle T ₁ of the primary resonance circuit. In period II, the secondary resonance circuit becomes conductive and causes resonance, which acts on the resonance of the primary resonance circuit via the main transformer. Therefore, the resonance of the primary resonance circuit is converted to the primary side. The capacitance and the inductance of the secondary resonant circuit are performed in the resonant period T ₂ of the equivalent resonant circuit connected in parallel with the self-induction of the primary winding. Therefore, in the period II, the voltage applied to the main switch changes slowly with the time constant T ₂ larger than T ₁ . In the period III, the secondary resonance circuit is blocked from conducting again, so that the voltage applied to the main switch is again in the period T ₁
Draw a sine wave of and drop down to 0 volts. Thus, the voltage applied to the main switch is
Then, it rises and falls with a sinusoidal waveform, but changes slowly in period II. Therefore, the applied voltage of the main switch is
The peak portion becomes a truncated sine wave, that is, a truncated sine wave, and high peak voltage is prevented from being applied to the main switch.

As described above, in the primary circuit of the main transformer, the primary resonance current flows in the period II so that the voltage applied to the main switch becomes a truncated sine wave. As a result, a slowly changing secondary voltage is generated in the secondary winding during the period II. As is well known, after the second diode (rectifying diode) is turned off, the secondary voltage of the main transformer is applied to the second diode as a reverse voltage. Therefore, the period II in which the reverse recovery current flows through the second diode II
When the change in the secondary voltage (rise in the reverse direction of the secondary voltage) is small in the initial stage of, the power loss due to the secondary voltage of the reverse recovery current is small. Since the snubber circuit of the present invention slowly changes the secondary voltage in the period II, the power loss is reduced.

The output voltage adjusting circuit provided in the output circuit of the DC-DC converter of the present invention is turned on by utilizing that the secondary current supplied to the load of the converter, that is, the rectified current has a simple waveform with respect to time.・ The output voltage is adjusted by the off-phase control. As is well known, DC-DC
In the case of a forward converter, the rectified current increases as a linear function of time during the on-time of the main switch, DC-
In the case of a DC flyback converter, it usually decreases as a linear function of time during the off period of the main switch, and in the case of a voltage resonant DC-DC flyback converter, it decreases with a triangular function waveform close to a cosine function. Thus, D
The rectified current of the C-DC converter is a monotonically increasing function of time,
Alternatively, since it has a waveform of a monotonically decreasing function, each time or each phase point where the rectified current is output in one switching cycle can correspond to the current value of the rectified current, that is, the voltage signal. As a result, the absolute value of the difference between the two voltage signals or the absolute value signal corresponds to the time between the two times corresponding to those voltage signals. Now, assuming that the first switching state of the switch element is on and the switch element is turned on at time t _{1 under} the control of the first switching control circuit, supply of current to the load is started. Therefore, the current supply time t until time t
-T ₁ corresponds to the absolute value signal absΔv. Where ab
s represents an absolute value, Δv represents v−v ₁ , and v, v
₁ indicates the voltage signal at the times t and t ₁ , respectively.
Since the voltage signal v changes the same as the rectified current, v is also a monotonic function of time. Therefore, absΔv increases with time and reaches the threshold Δv _TH at a certain time t ₂ .
At that time, the second switching control circuit turns off the switch element. Thus the time t ₂ -t ₁ is a total time of supplying the rectified current to the load in the switching period, which corresponds to the threshold Delta] v _TH of the absolute value signal.

When adjusting the output voltage of the output circuit to the set value, if the output voltage is larger than the set value, the current supply time per switching cycle must be shortened. In other words, when the deviation signal is large, the threshold value Δv _TH
Must be small. Therefore, the threshold Δv _TH
Must be defined as a decreasing function of the deviation signal.

[0019]

Embodiments of the present invention will now be described with reference to the drawings. FIG. 1 shows a voltage resonance DC-provided with the snubber circuit of the present invention.
It is a circuit diagram of a DC forward converter. Since the main part of the converter has already been described with reference to FIG. 12 above, the description thereof will be omitted. The snubber circuit 18
Series connection of capacitor 8 and diode 9 and coil 10
And consists of. Both ends of the series connection are connected to both ends of the secondary winding 7 of the main transformer 5, and the secondary winding 7, the diode 9 and the capacitor 8 constitute a series resonance circuit which conducts only in the forward direction of the diode 9. ing. Hereinafter, this resonance circuit will be referred to as a secondary resonance circuit, and the resonance current generated in the secondary resonance circuit will be referred to as a secondary resonance current. In the forward direction of the diode 9, the secondary winding 7 passes through the rectifying diode 11 and the smoothing circuit 1
While the secondary current is being supplied to 2, the diode 9 is oriented in the direction of blocking the generation of the secondary resonance current. The coil 10 is connected between the connection between the capacitor 8 and the diode 9 and the cathode of the rectifying diode 11, and forms a discharge path for the electric charge accumulated in the capacitor 8. The discharge is performed when the diode 9 is off. The coil 15 represents the leakage inductance of the main transformer 5 and the inductance of an externally attached coil when necessary. The capacitance of the capacitor 8 makes the rise of the resonance voltage applied to the main switch 3 gentle when the main switch 3 is turned off and lowers the peak, and does not make the off period of the main switch 3 longer than a predetermined value. Is determined. Coil 1
The zero inductance is set large enough to prevent ripple at the converter output, but small enough to reset the charge on capacitor 8 before the next cycle of voltage resonant switching begins. Therefore, the inductance of the coil 10 is determined so that the time constant due to the capacitance of the capacitor 8 and the inductance of the coil 10 is approximately the same as the average period of voltage resonance switching.

Next, the operation of this snubber circuit will be described. FIG. 2 is a time chart of signals of respective parts for explaining the operation of the snubber circuit of the present invention, and FIG. 3 is from the secondary side of the voltage resonance DC-DC forward converter during the period when the secondary resonance circuit is conducting. The equivalent circuit seen is shown. When the main switch 3 is turned on at time t ₀ , the primary current flows to the primary winding 6
Rises with a constant gradient, and thereby a constant induced electromotive force E ₂ is generated in the secondary winding (see FIG. 2B). The induced electromotive force E ₂ generates a secondary current that rises with a constant gradient (see FIG. 2 (c)). Then the gradient is (E ₂ −V _O
−V _F ) / L _e . Here, E ₂ is the equivalent electromotive force viewed from the secondary side, and E is the electromotive force of the DC power source 1, and n ₂ / n ₁
E ₂ = (n ₂ / n ₁ ) E, where is the winding ratio of the main transformer. V _O and V _F are the output voltage of the converter and the forward voltage applied to the diode 11, respectively,
L _e is the inductance of the coil 15.

Time t₁ And the main switch 3 is turned off.
And the resonance current generated in the primary winding 6 (hereinafter, primary resonance
Voltage applied to the main switch 3 by
V_DSIs the period T₁= 2π (L_PC_P)^1/2 Rise with the waveform of the sine function of
It Where L_P , C_P Is the primary winding inductor
And the capacitance of the resonance capacitor 4. At the same time 2
In the next winding 7, the same period T₁ The induced electromotive force of
Induction electromotive force generated in the secondary winding 7 when the switch 3 is in the ON state
The direction of force is positive. ), And as a result,
Voltage V across the secondary winding 7₇ Is shown in Figure 2 (b)
So as to fall. Voltage V₇ Diode by the drop of
11 forward current I_D Decreases at time t ₂ Current I_D Is
It becomes 0.

Time t₂ Turns off the diode 11
Then, the secondary resonance circuit becomes a circuit other than the electric path passing through the coil 10.
Is cut off from the smoothing circuit 12. Secondary under this condition
The electromotive force generated in the winding 7 in the opposite direction is applied to the diode 9.
Diode 9 turns on when the applied reverse bias is exceeded.
Then, the secondary resonance current is conducted in the forward direction. Figure
3 is the voltage resonance DC-DC flux when the secondary resonance occurs.
An equivalent circuit of a forward converter is shown. Resonance is induc
Closet L_P , Capacity C_P And C_S (Inductance L_e 
(Ignore), and the resonance period T₂ Is 2π [L_P(C_P
＋ (n₂/ n₁)²C_S}] ^1/2 become. In this way, V in FIG._DS
And V₇ As shown in the time chart of
Kana rising and low peak values are achieved. Capacitor
When the voltage of 8 reaches the maximum value, the reverse resonance current
The secondary resonance stops because it is blocked by the ode 9.
The charge stored in the capacitor 8 is the coil 10
It is discharged through and is input to the smoothing circuit 12.

As is clear from the above description, the secondary resonance
Is blocked at time t₁ From time t₂ Period (term
In the interval I), the voltage V_DSAnd V₇ The curve is short
Period T ₁ Rises with the waveform of the sine wave of and secondary resonance occurs.
Time t₂ From time t₃ Up to period (Period II)
Is the voltage V_DSAnd V₇ Curve is long except for transition period
Cycle T₂ Changes with the waveform of the sine wave, and secondary resonance is blocked again
Time t₃ From time t_Four (Main switch 3 off period
Voltage (V) during the period (term III)_DSAnd
And V₇ The curve of is a sine wave with a short cycle and falls. Term
Voltage V at intervals I and III_DSThe steep rise of
Shortening the OFF period of the main switch 3 by lowering
(If the off period is too long, the converter output is reset.
Can be plucked). In addition, the voltage V in the period II_DSNoyu
Voltage V due to surprised behavior _DSLower the peak value of
As a result, low voltage, low on-resistance switch
The switching element can be used as a main switch.

The snubber circuit of the present invention has another advantage.
I have it. Time t at the beginning of period II ₂ At Daio
When the switch 11 is switched from the on state to the off state, the die
A reverse recovery current flows through the ode 11. This current is
Shown as reverse current in (c). Reverse recovery current is time
t₂ From time t_Five Until period (period IV) flows. As above
As described above, the diode 9 is in the ON state in the period IV.
Voltage V₇ Via diode 9 and coil 10
The reverse voltage is applied to the diode 11. As a result, the voltage
V₇ And the reverse recovery current create a positive power factor, which
Force loss occurs. This power loss is
Pressure V₇ The less steep the gradient, the less. The present invention
Voltage V₇ Stands slowly during period IV.
Do not use the snubber circuit of the present invention because it goes up (reverse direction)
If (in this case the cycle T₁ The sine wave of
Power loss can be reduced as compared with

The operation of the snubber circuit of the present invention is summarized as follows from the viewpoint of energy conservation. When the main switch is turned off, the residual magnetic energy of the main transformer causes a primary resonance current to flow. If the snubber circuit of the present invention is not provided, the energy of this primary resonance current is the charging voltage of the resonance capacitor 4 as shown in FIG.
It is large enough to make the peak represented by the dotted line. However, when the snubber circuit is provided, a part of the residual magnetic energy is stored in the capacitor 8 as electrostatic energy via the secondary resonance current. Therefore, the energy of the primary resonance current is reduced by the amount of the electrostatic energy of the capacitor 8, and as a result, the voltage of the resonance capacitor 4 (= V _DS ).
Becomes a waveform in which the peak part is cut off because it cannot rise. On the other hand, the magnetic energy stored as electrostatic energy in the capacitor 8 is supplied to the load via the coil 10 and is effectively used there.

The snubber circuit of the present invention can be used in half-wave and full-wave voltage resonant DC-DC forward converters, as shown in FIGS. 4 and 5, respectively.

Next, the adjusting circuit for adjusting the output voltage of the multi-output type DC-DC converter of the present invention will be described. FIG. 6 is a circuit diagram showing a first embodiment of the adjusting circuit provided in one of the output circuits. The output circuit includes a smoothing circuit 12 and a rectifying diode 11 connected to the secondary current circuit of the main transformer 5 as in the case of a normal output circuit. In this embodiment, the diode 11 is connected to the positive potential circuit of the secondary current. The adjustment circuit 20 switches N for switching ON / OFF of the secondary current of the main transformer 5.
Channel MOSFET 21, resistor 22 (first switching control means) for controlling MOSFET 21, series dropper resistor 23 (current / voltage conversion means), comparison circuit 24 (comparison means), switching control circuit 25 (second switching control) Means).

The drain current conducting path of the MOSFET 21 is connected in series to the load to the negative potential path of the secondary current,
The gate is connected to the positive potential side of the secondary current circuit via the resistor 22. Thereby, the MOSFET 21 is turned on in synchronization with the switching on of the main switch of the converter. Hereinafter, the time when the MOSFET 21 is turned on in synchronization with the turning on of the main switch is referred to as t ₀ . Series dropper resistance 2 with resistance value R _D
3 is connected in series to the load and is used to detect the secondary current value supplied to the load. Hereinafter, the voltage generated across the series dropper resistor 23 by the secondary current is referred to as a voltage signal. Since the voltage signal is ₀ at time t ₀ and increases as a linear function of time in the case of the forward converter, the voltage signal at the current time t corresponds to the absolute value signal described above. The comparison circuit 24 is a shunt regulator element 26.
And a resistor 27 connected in series, the resistor 27 is connected to a point a on the load side of the series dropper resistor 23, and the shunt regulator element 26 is connected to a point b on the negative potential circuit of the secondary current. The comparison circuit 24 further includes voltage dividing resistors 28 and 29, and the resistors 28 and 29 have two points a and b.
The output signal S corresponding to the voltage between them, that is, the output voltage V _O of the output circuit is output to the shunt regulator element 26. The shunt regulator element 26 has an output voltage V
A constant voltage source for generating a reference output signal S ₀ corresponding to the set value of _O is provided, and the output signal S and the reference output signal S ₀ are compared,
The shunt current I _S corresponding to the deviation of the output signal S from the reference output signal S ₀ is conducted. When the output signal S matches the reference output signal, the shunt current I _S has the zero deviation value I _S0 . The shunt current I _S is the resistance 27 of the resistance value R _S.
Generates a voltage V _S across Therefore, the larger the deviation S-S ₀ of the output signal, the higher the voltage V _S, and the lower the potential E at the connection point d between the resistor 27 and the shunt regulator element 26. Hereinafter, the voltage V _S will be referred to as a deviation signal.

The switching control circuit 25 includes a pnp transistor 30 and an N-channel MOSFET 31. The emitter of the pnp transistor is the diode 11
Is connected to the connection point of the series dropper resistor 23, the base is connected to the connection point d, and the collector is MOSFET3.
1 is connected to the gate. The source of the MOSFET 31 is connected to the negative potential circuit of the secondary current, and the drain is M
It is connected to the gate of the OSFET 21. Furthermore, it is desirable that the switching control circuit 25 includes a discharge circuit including a resistor 32 and a diode 33. This discharge circuit operates when the transistor 30 is turned on.
It is provided to discharge the electric charge charged in the input capacitance of the SFET 31.

Next, the operation of this embodiment will be described. When the main switch turns on, the primary current rises as a linear function of time, which induces a constant induced electromotive force in the secondary winding during the on period of the main switch. In this way, a constant voltage V ₇ is generated as shown in FIG. Since the gate of the MOSFET 21 is biased by the voltage V ₇ through the resistor 22, the MOSFET 21 is turned on with the generation of the voltage V ₇ , and the secondary current I _D (hereinafter, referred to as current I _D ) passes through the rectifying diode 11. It begins to flow. The base-emitter voltage is

[0031]

## EQU1 ## Since V _BE = I _D R _D + I _S R _S (1) is given, the right side of the equation (1) is the transistor 30.
Transistor 30 turns on when its base-emitter threshold voltage V _TH is exceeded. Therefore, the threshold value of the absolute value signal I _D R _D is given by the following equation.

[0032]

(2) (I_D R_D)_TH= V_TH-I_S R_S (2) As is clear from the equation (2), the threshold value (I_D 
R_D)_THIs the deviation signal I_S R _S Is a decreasing function of. Formula (2)
Is represented as follows.

[0033]

## _EQU3 ## I _DTH R _D = V _TH −I _S R _S (3) Here, I _DTH represents the value of the current I _{D at} the time t _TH when the transistor 30 is turned on. As is apparent from FIG. 2C, the current I _D increases as a linear function of the time t during the ON period of the main switch. Therefore, the current I _D can be expressed in the following form.

[0034]

## EQU4 ## I _D = I _O + kt (4) Therefore,

[0035]

## _EQU5 ## I _DTH = I _O + kt _TH (5) From equations (3) and (5), the value of the time t _TH is the deviation signal I _S.
The larger R _S is, the smaller it is. Therefore, when the deviation signal is large, the transistor 30 turns on early.
When the transistor 30 turns on, the MOSFET 31
Turns on, which shorts the gate and source of MOSFET 21 and turns MOSFET 21 off. In this way, when the deviation of the output voltage V _O is large, the MOSFET 21 turns off early, and as a result, the deviation is compensated.

The coil 15 is preferably provided at the position shown in FIG. 7 rather than the position shown in FIG.
The reason is that, in the case of FIG. 7, the gate of the MOSFET is connected to the secondary winding 7 of the main transformer 5 without passing through the coil 15, so a high voltage is applied to the gate,
As a result, the on-resistance of the MOSFET 21 is reduced, and the discharge circuits 32 and 33 are directly connected to the secondary winding 7, so that the discharge time constant is shortened.

FIG. 8 shows a second embodiment of the adjusting circuit of the present invention. Also in the adjusting circuit 40 of the present embodiment, as in the adjusting circuit of FIG. 6, the MOSFET 21 (switch element), the switching control circuit 42 (first switching control means),
A current / voltage conversion circuit 43 (current / voltage conversion means), a comparison circuit 24 (comparison means), and a switching control circuit 44 (second switching control means) are provided. The switching control circuit 44 further comprises a switching control unit 25 and a differential amplifier 45. Of these, MO
The SFET 21, the comparison circuit 24, and the switching control unit 25 are the same as the corresponding circuits in FIG. The discharge circuit consisting of the resistor 32 and the diode 33 of FIG. 6 is not shown in FIG.

The switching control circuit 42 is composed of a pulse transformer 41, a reset circuit including a resistor 49 and a diode 50 connected in series. The reset circuit is provided for the purpose of dissipating the surge energy in order to reduce the flyback voltage generated in the drive transistor provided in the primary circuit of the main transformer 5. By the switching control circuit 42,
The MOSFET 21 can turn on the MOSFET 21 in a predetermined phase relationship with the switching operation of the main switch. The current / voltage conversion circuit 43 is
It is composed of a current transformer 46, a diode 47 and a capacitor 48. The capacitor 48 and the inductance of the secondary winding of the current transformer form a one-way resonance circuit in the forward direction of the diode 47. Therefore, the capacitor 4
The voltage at 8 changes as a sine function at time t. However, since the resonance cycle is sufficiently longer than the switching cycle of the main switch, the voltage of the capacitor 48, that is, the voltage signal I _V , corresponds to the primary current of the current transformer 46 as a linear function of time. To increase. Similar to the embodiment of FIG. 6, the secondary current of the current transformer 46 is ₀ at time t ₀ . Therefore, the voltage signal I _{V at} the current time t represents an absolute value signal.

The inverting and non-inverting input of the differential amplifier 45 are respectively I _V and V _O -I _S R _S. Where V
_O , I _S and R _S are defined quantities as in the circuit of FIG. The output of the differential amplifier 45 is V _O − (I _S R _S + I _V ). Therefore, the base-emitter voltage of the transistor 30 becomes − (I _S R _S + I _V ), and − (I _S R _S + R in FIG.
_{Same as D} I _D ). Therefore, the transistor 30 operates exactly as in the case of FIG.

FIG. 9 is a time chart of the current I _D output from the adjusting circuits of FIGS. 6 and 8. FIG. 9A shows 1
When resonance in the next circuit occurs due to the inductance of the primary winding 6 and the resonance capacitor 4 (hereinafter referred to as case
9B shows a current I _D in FIG. 9A, and FIG. 9B shows a current in the case where resonance in the primary circuit is caused by the external coil 2 and the resonance capacitor 4 (hereinafter referred to as case (b)). Represents I _D. The solid and dotted lines in FIG. 9 represent the current output from the regulation circuits 20 and 40, respectively.

When the adjusting circuit 20 is used, the MOSFET 2 is synchronized with the rise of the induced electromotive force of the secondary winding.
1 turns on, and at the same time, the current I _D starts to flow. If all the coils provided in the primary circuit of the main transformer 5 are reset during the off period of the main switch 3, that is, in all the coils in the primary circuit during the previous on period. If the stored magnetic energy is dissipated during the current off period, then during the next on period, FIG.
As shown in (a), the current _ID rises from zero. However, on the contrary, if the magnetic energy stored in the immediately preceding ON period remains undissipated by the end of the current OFF period, the primary current rises sharply in the next ON period, and therefore 2 The next current also sharply rises as shown in FIG. 9 (b). Further, due to the resonance generated in the primary circuit by the coil 2 and the capacitor 4, the rise of the secondary current, and thus the rectified current I _D , also has a sinusoidal waveform as shown in FIG. 9B. Stand up.

When the regulation circuit 40 is used, the current I _D begins to flow by the switching control circuit 42 in synchronism with any synchronization signal, as shown by the dotted line in FIG. Current exceeds the base-emitter threshold voltage of transistor 30, the preferential control of switching control unit 25 blocks current I _D. Here, the priority control means that in FIG.
When the transistor 30 is turned on, the MOSFET 2
The gate and source of 1 are short-circuited. Therefore, MO
This means that the SFET 21 is cut off from the control of the switching control circuit 42 and turned off by the control of the switching control unit 25.

As is well known, in the DC-DC forward converter, during the ON period of the main switch 3, the current I
_D changes as a linear function of time t. Now, let I ₀ be the initial current, that is, the value of the current I _D when the main switch 3 is turned on, and let k be the time t of the current I _D.
If it is a gradient with respect to, the following equation holds.

[0044]

## EQU6 ## I _D = I _O + kt (6) As described above with reference to FIG. 2C, k is represented by the following equation.

[0045]

Equation 7] _{k = (E 2 -V O -V} F) / L e (7) power supplied from the conditioning circuit to the load is given by the following equation.

[0046]

[Formula 8] P = f∫I _D V _O dt (8)
Here, f is the switching frequency, and the integration is performed from time t ₁ to time t ₂ shown in FIG. Output voltage V
_{Since O} is almost constant, the power P is given by the following equation.

[0047]

[Equation 9] P = (T _ON + T _OFF ) ⁻¹ _VO A (9) = [2 (T _ON + T _OFF )] ⁻¹ (I _D1 + I _D2 ) (t ₂ −t ₁ ) (10) where A represents the area of the I _D -t plane between times t ₁ and t ₂ . In the equation (10), I _D2 and t ₂ are automatically selected by the switching control circuits 25 and 44 depending on the deviation signal, and I _D1 is determined by the equation (6) from time t ₁ . Therefore, the power P is adjusted to minimize the deviation of the output V _O.

FIG. 10 shows an application example of the present invention. This embodiment is a circuit of a multi-output voltage resonant DC-DC forward converter to which the snubber circuit of the present invention and the adjusting circuit 20 or 40 are applied. The converter is a monolithic voltage resonant DC-DC forward converter with two outputs. The main output circuit 60 comprises the snubber circuit of FIG. 1, and the output of the main output circuit is fed back via the loop control circuit 14 to control the switching of the main switch 3. The auxiliary output circuit 61 is the regulating circuit 20 or 40 shown in FIGS.
And the auxiliary output is adjusted independently of the main output by this adjusting circuit.

FIG. 11 shows another application example of the present invention. In this embodiment, the drive circuit 63 for driving the main switch 3 is independent of the multi-output circuit and is the main transformer 1
It is provided in the next circuit. Each of the multiple output circuits has its own regulation circuit 20 or 40 to provide the regulated output to the load, respectively, independently of the other circuits.

[0050]

As described above, the DC-DC of the present invention
The converter has the following effects. 1. A voltage resonance DC-DC forward converter with a snubber circuit, which is a first DC-DC converter of the present invention,
By storing the energy of the induced current flowing through the secondary winding of the main transformer as electrostatic energy in the capacitor of the snubber circuit while the supply of the secondary current to the load is blocked, (a) it is applied to the main switch. The waveform of the applied voltage can be a truncated sine wave, so that the low voltage,
A switch element having a low on-resistance can be used as a main switch, and (b) when the rectifying diode is turned off, the rising slope of the reverse voltage applied to the diode can be made gentle, and The power loss due to the reverse recovery current can be reduced by (c)
By discharging the electric charge accumulated in the capacitor of the snubber circuit to the load through the coil, the electrostatic energy of the capacitor can be effectively used. 2. A multi-output DC-DC converter having an output voltage adjusting circuit which is a second DC-DC converter of the present invention,
Of the on / off switching of the secondary current supplied to the load, the first switching is performed with a predetermined phase relationship with the operation of the main switch, and the second switching is performed by using the absolute value signal as a decreasing function of the deviation signal. By performing the operation when the threshold value is exceeded, it is possible to control the current supply time to the load for each switching cycle without loss of electric power, thereby compensating the deviation of the output voltage.

[Brief description of drawings]

FIG. 1 is a voltage resonant DC-D equipped with the snubber circuit of the present invention.
It is a circuit diagram of a C forward converter.

FIG. 2 is a time chart of signals of various parts for explaining the operation of the snubber circuit of the present invention.

FIG. 3 is an equivalent circuit seen from the secondary side of the voltage resonant DC-DC forward converter during a period when the secondary resonant circuit is conducting.

FIG. 4 is a half waveform voltage resonance D using the snubber circuit of the present invention.
It is a C-DC forward converter.

FIG. 5 is a full waveform voltage resonance D using the snubber circuit of the present invention.
It is a C-DC forward converter.

FIG. 6 is a circuit diagram of a first embodiment of the adjusting circuit of the present invention.

FIG. 7 is a circuit diagram of a modified example of the adjustment circuit of FIG.

FIG. 8 is a circuit diagram of a second embodiment of the adjusting circuit of the present invention.

9 is a current I _D output from the adjusting circuit of FIGS. 6 and 8;
Is a time chart of.

FIG. 10 is a configuration diagram of a multi-output voltage resonant DC-DC forward converter to which the snubber circuit and the adjusting circuit of the present invention are applied.

FIG. 11 is a multi-output type voltage resonant DC-DC forward converter in which each output circuit is provided with the adjusting circuit of the present invention, and the main switch is driven independently from the output circuit.

FIG. 12 is a circuit diagram of a conventional example of a one-stone voltage resonance DC-DC forward converter.

FIG. 13 is a configuration diagram of a conventional example of a multi-output single-stone voltage resonance DC-DC forward converter.

[Explanation of symbols]

1 DC Power Supply 2 Input Capacitor 3 Main Switch 4 Resonance Capacitor 5 Main Transformer 6 Primary Winding 7 Secondary Winding 8 Capacitor 9 Diode 10 Coil 11 Rectifying Diode 12 Output Capacitor 13 Load 14 Loop Control Circuit 15 Leakage Inductance 20, 40 Regulator circuit 21, 31 MOSFET 23 series dropper resistor 24, 44 comparison circuit 25 switching control circuit (switching control unit) 26 shunt regulator element 27, 28, 29, 32 resistor 30 transistor 41 pulse transformer 42 switching control circuit 43 current / voltage conversion Circuit 45 Differential amplifier 46 Current transformer

Claims (7)

[Claims] 1. A capacitor and a first diode are connected in series and connected to both ends of a secondary winding of a main transformer to conduct a resonance current in the forward direction of the first diode together with the secondary winding. A series circuit that forms a resonant circuit that operates and a discharging unit that discharges the charge accumulated in the capacitor during the off period of the first diode, and the forward direction of the first diode is the secondary winding. Second for rectification from wire
Voltage resonance type DC-DC having snubber means that is determined so that conduction of the resonance current is blocked during the period in which the secondary current is supplied to the smoothing circuit via the diode.
Forward converter. 2. The converter according to claim 1, wherein the discharging means has an inductive element connected between the connection point of the capacitor and the first diode and the smoothing circuit. 3. A main transformer 2 of a DC-DC converter.
Switch means for switching the secondary current supplied from the secondary winding to the load, and the switch means so as to take the first switch state with a predetermined phase relationship with the switching operation of the main switch of the DC-DC converter. A first switching control means for controlling the means, a current / voltage conversion means for generating a voltage signal that changes the same as the secondary current, and a deviation signal corresponding to the deviation of the output voltage of the output circuit from the set value. Comparing means for outputting and the absolute value of the difference between the current voltage signal and the first voltage signal when the voltage signal when the switching means transits to the first switching state is the first voltage signal. When the absolute value signal exceeds a threshold determined as a decreasing function of the deviation signal, the switch means is controlled so as to assume the second switching state. A multi-output DC-DC converter having at least one output circuit, the output voltage adjusting means having a second switching control means. 4. The multi-output DC-DC converter according to claim 3, wherein the threshold value of the absolute value signal is determined such that the sum of the threshold value and the deviation signal is a predetermined constant. 5. The second switching control means has a transistor, the sum of the absolute value signal and the deviation signal is applied between the base and emitter of the transistor, and the predetermined constant is the base-emitter threshold voltage. 4. The multi-output DC-DC converter described in 4. 6. The multi-output DC-DC converter according to claim 3, wherein the main switch is driven by driving means that operates independently for any output circuit. 7. A multi-output DC-DC converter according to claim 3, wherein the converter is a voltage resonance type forward converter, and one of the output circuits is a main output circuit provided with the snubber means according to claim 1.