JPH0567927A - Amplifier circuit - Google Patents

Abstract

PURPOSE:To improve the efficiency by reducing a loss of an output stage element due to an idle current in the amplifier circuit of class A operation. CONSTITUTION:An input signal is converted into a current by a voltage-current conversion circuit 8. The current is inputted to a current/current conversion circuit 3. The circuit 3 consists of four transistors(TRs) 1-4. The TRs 1,2 act like class A amplifiers (non-cut-off) and output currents I1, I2 in response to the input. The currents I1, I2 are converted into voltages respectively by resistors r1, r2 via current mirror circuits 6,7, amplified by output stage TRs Qa, Qa' to drive a load 10. Power is supplied mainly from a main power supply line 42 in switching drive. Since the main power supply path 42 keeps a voltage VQa of the TRs Qa, Qa' to a minimum value, even when the idle current is large, the loss in the TRs Qa, Qa' is reduced. The leading delay by a smoothing coil L1 is compensated by an auxiliary power supply line 48.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION This invention is a non-cutoff (A
Class) operation, a linear output synthesis characteristic is obtained, and the efficiency is improved with respect to the improvement of an amplifier circuit that can be used for large output amplification.

[0002]

BACKGROUND OF THE INVENTION In a conventional class A amplifier, the base bias is selected so that the operating current (collector current when there is no signal) is 1/2 of the peak current (that is, the center point of the operating curve), and the positive / negative of the input signal is selected. The amplification element is always in the active state during both cycles, and thus linear amplification is performed. FIG. 2 shows a class A push-pull amplifier as an example thereof, in which deep base biases E1 and E2 are added to transistors 1 and 2 having a push-pull structure. The input / output characteristics of the class A push-pull amplifier of FIG. 2 are shown in FIG. According to this, since the base bias is applied to the transistors 1 and 2, the transistors 1 and 2 do not cut off unlike the class B amplifier, and the distortion is low.

However, in such a structure, a large idle current is required, and a range for class A operation is determined, and for an input of a higher level,
One of the transistors has the drawback of being cut off and causing distortion. Therefore, an amplifier circuit described in Japanese Patent Application Laid-Open No. 62-214707 has been proposed as a solution to such a drawback so that the idle current can be reduced and a wide range of class A operation can be obtained. There is.

This uses the current-current conversion circuit shown in FIG. This current-current conversion circuit uses an NPN
The transistor Tr1 and the PNP transistor Tr3 are commonly connected to each other, and a signal is inputted from this common emitter. The base of the transistor Tr1 is NPN
It is connected to the base of the transistor Tr2. Also,
The base of the transistor Tr3 is a PNP transistor Tr
4 is connected to the base. Transistors Tr2, T
Each of r4 has its base and collector commonly connected, and its emitters commonly connected. This common emitter is connected to a reference potential (here, ground potential). A constant current is supplied to each of the transistors Tr2 → Tr4. In this circuit, the collector currents of the transistors Tr1 to Tr4 are set to Ic1 to Ic.
Set to 4. The collector currents Ic2 and Ic4 of the transistors Tr2 and Tr4 are constant currents, and the values thereof are equal to I _D here.

In the circuit of FIG. 4, when I _D = 1 and the values of Ic1 and Ic3 with respect to the input current Ix are obtained and plotted, it becomes as shown in FIG. 5 (for detailed analysis, see Japanese Patent Laid-Open No. 62-214707). See the bulletin). According to this, even if the input current Ix is increased and one of Ic1 and Ic3 is increased, the other only approaches 0 and does not cross zero. The value obtained by adding Ic1 and Ic3 becomes a straight line passing through the origin.

From the above, it can be seen that the circuit of FIG. 4 has the following characteristics. The input current Ix is converted into output currents Ic1 and Ic3. Regardless of whether the input current Ix is positive or negative, the positive and negative of the output currents Ic1 and Ic3 are constant and are not inverted. Ix = Ic1 + Ic3 Ic1 × Ic3 = constant value Therefore, if the output currents Ic1 and Ic3 are respectively amplified by the output stage elements and the output currents of these output stage elements are combined and supplied to the load, The element is always in the active state and is in class A operation.

[0007]

2. Description of the Related Art FIG. 6 shows a conventional amplifier circuit using the current-current conversion circuit shown in FIG. Input signal Vin
Is input to the voltage-current conversion circuit 8. In the voltage-current conversion circuit 8, the transistors Tr5 and Tr6 form a differential amplifier. The transistor Tr9 and the resistor r6 form a constant current circuit of this differential amplifier, and flow a constant current regulated by the voltage across the diode d3 and the resistor r3. A current mirror circuit composed of transistors Tr7 and Tr8 is connected to the differential amplifier. The input signal Vin is input to the base of the transistor Tr5,
The collector output of the transistor Tr5 is input to the base of the transistor Tr10. Therefore, when the input Vin is high, the base potential of the transistor Tr10 drops, and when the input Vin is low, the base potential of the transistor Tr10 is low.
The base potential of rises.

A transistor Tr10, a resistor r4, and transistors Tr11 and Tr1 are provided between the power sources of + B1 and -B1.
2, the resistor r5 is connected in series. Also, voltage + B
The voltage between 1 and -B1 is divided by the resistance r1, the diodes d1, d2, the resistance r2, the diode d3, and the resistance r3. The base of the transistor Tr11 is a diode d
2 and the resistor r2. It is also grounded via a capacitor C ₀ . Therefore, the voltage of the resistor r1 and the voltages of the diodes d1 and d2 are applied between the collector of the transistor Tr10 and the base of the transistor Tr11. Therefore, when the base potential of the transistor Tr10 rises, the transistor Tr10 → the resistor r4 →
The current flowing through the transistor Tr11 increases, and when the base potential of the transistor Tr10 decreases, this current decreases. The collector of the transistor Tr11 is connected to the current-current conversion circuit 3 in the next stage via the line 13.

The transistor Tr12 and the resistor r5 form a constant current circuit, and a constant current defined by the voltage across the diode d3 and the resistor r3 flows. Therefore, the line 13 has the collector current Ic1 of the transistor Tr11.
The current Ix that is the difference between 1 and the collector current Ic12 of the transistor Tr12 flows.

That is, the voltage-current conversion circuit 8 outputs the following current Ix according to the input Vin. When Vin = 0, Ic11 = Ic12 and Ix = Ic11−Ic12
= 0. When Vin <0: The collector potential of the transistor Tr5 (that is, the base potential of the transistor Tr10) rises, and Ic11> Ic
12, and the difference currents Ic11-Ic12 flow out to the current-current conversion circuit 3 as the current Ix. When Vin> 0, the collector potential of the transistor Tr5 drops, and Ic11
<Ic12, and the difference current Ic12-Ic11
Flows from the current-current conversion circuit 3 as the current Ix.

In this way, the input Vin
A current Ix proportional to the current flows and voltage-current conversion is performed. The output current Ix of the line 13 is the current-current conversion circuit 3
Is input to Ic1 + Ic3 and Ic1 × Ic3 = constant as described above, and controls the values of the currents Ic1 and Ic3.

The current mirror circuit 6 includes a differential amplifier 15
The output-stage transistor Tr13 is controlled so that the voltages of both inputs are equal. Since the current Ic1 flows through the resistor r7 connected to the + input of the differential amplifier 15, r7
・ Ic1 voltage drop occurs. Therefore, the resistance r8 connected to the-input of the differential amplifier 15 has a voltage drop equal to that of the resistance r7 such that Ic1.
A current of (r8 / r7) flows. Therefore, the current amplification factor A of the current mirror circuit 6 is A = r8 / r7.

Similarly, the current mirror circuit 7 controls the output stage transistor Tr14 so that the voltages of both inputs of the differential amplifier 17 become equal. Since the current Ic3 flows through the resistor r9 (= r7) connected to the + input of the differential amplifier 17, a voltage drop of r9 · Ic3 occurs. Therefore, the resistance r connected to the-input of the differential amplifier 17
In 10 (= r8), a current of Ic3 · (r10 / r9) flows so that the voltage drop becomes equal to the voltage drop of the resistor r9. Therefore, the current amplification factor A of the current mirror circuit 7 is A = r10 / r9 = r8 / r7.

As described above, the transistor Tr1
Collector currents A.Ic1 and A.Ic3 flow in Tr3 and Tr14. The voltage corresponding to the difference is the output V
As out, it is taken out from the connection point of the transistors Tr13 and Tr14 and supplied to the load 10. Also, the output Vou
t is a transistor Tr6 via resistors r11 and r12.
Negative feedback to the base of.

According to the circuit of FIG. 6, the currents Ic1 and Ic2 are
Does not become 0, the currents A.Ic1 and A.Ic3 also do not become 0, and therefore the output stage transistor Tr1
3, Tr14 does not cut off and becomes Class A operation.

[0016]

In order to effectively utilize the characteristics shown in FIG. 5 in a practical circuit, the output stage transistors Tr13 and Tr14 in the circuit shown in FIG. It is necessary to pass an electric current. However, in the circuit of FIG. 6, the output stage transistor Tr13,
Tr14 is always up may take the output magnitude of the power supply voltage + B _3, because it is driven by -B _3, input a large voltage is applied between the collector and emitter of the output stage transistor Tr 13, Tr14 even at zero ing. Therefore, the power loss (the amount of heat generation) in the output stage transistors Tr13, Tr14 becomes large, and there is a drawback that the efficiency is poor although the performance is high. The present invention intends to provide an amplifier circuit which improves the efficiency by improving the drawbacks of the conventional technique and reducing the loss in the output stage element due to the idle current.

[0017]

According to the present invention, the current-
In an amplifier circuit for extracting an output current having the characteristics shown in FIG. 5 by using a current conversion circuit and a current mirror circuit, a voltage signal corresponding to the output current of the current mirror circuit and the input signal to be amplified are combined and output. Voltage synthesizer circuit, an output stage element for amplifying the output voltage of the voltage synthesizer circuit, an output circuit for synthesizing output currents of these output stage elements and supplying them to a load, and a driving power supply for the output stage elements. A main power supply path, a switching element that is inserted in the main power supply path to turn on / off the main power supply path, and an output of the switching element that is inserted in the main power supply path and is smoothed and supplied to the load. The smoothing circuit and the switching element are variably switched according to the level of the input signal so that the ratio of the ON period and the OFF period is changed, so that the power required by the load is changed. And a switching control means for mainly supplying the power from the main power supply path, and a power supply path for supplying drive power to the output stage element separately from the main power supply path, the input signal having a faster response than the main power supply path. An auxiliary power supply path capable of supplying power following a level change, and an amount of power supply from the auxiliary power supply path to the load inserted in the auxiliary power supply path in a state of being cascade-connected to the output stage element. And an auxiliary electric power supply amount adjusting element for adjusting.

[0018]

According to the present invention, the drive voltage of the output stage element is lowered to the voltage corresponding to the input signal level by the main power supply path which is switching-driven, so that the applied voltage of the output stage element when the input is zero. Even if the idle current of the output stage element is set to a large value, the loss (heat generation) here is reduced, and the efficiency can be improved. Moreover, since the main power supply path itself is switching-driven, the loss in the main power supply path is small. Also, in case of abrupt rising of the input that cannot be followed by the main power supply line due to the presence of the smoothing circuit, power is supplied from the auxiliary power supply line which has better response than the main power supply line. It can follow the sudden rise of the amplitude input and does not clip the output. Even if the auxiliary power supply line is driven with such a power supply voltage, the power is usually supplied mainly from the main power supply line, so the average current value flowing through the auxiliary power supply line can be small. Losses on the road are small. As a result, an amplifier circuit having high performance, high efficiency, and good followability even for a sharp rise of a large amplitude input is realized.

[0019]

FIG. 1 shows an embodiment of the present invention. The input signal is input to the voltage-current conversion circuit 8 and the voltage amplifier 10
The voltage is amplified by. The output of the voltage amplifier 19 has a resistor r3.
Is connected, and a current Iin corresponding to the output voltage Vin of the voltage amplifier 19 is passed through the resistor r3 to perform voltage-current conversion.

The output current Iin of the voltage-current conversion circuit 8 is
It is input to the current-current conversion circuit 3. This circuit 3 has the same structure as that shown in FIG. That is, the NPN transistor Tr1 (first transistor) and the PNP transistor Tr3 (third transistor) are commonly connected to each other, and a signal is input from this common emitter. The base of the transistor Tr1 is connected to the base of the NPN transistor Tr2 (second transistor). The base of the transistor Tr3 is connected to the base of the PNP transistor Tr4 (fourth transistor). The transistors Tr2 and Tr4 have their bases and collectors connected in common, and their emitters connected in common. This common emitter is connected to a reference potential (here, the midpoint potential of the output circuit 21). A constant current I _D is supplied to the transistor Tr2 → transistor Tr4 by the constant current circuits 23 and 25, respectively. As a result, the currents I ₁ (I _c1 ), I ₂ (I
_c2 ) flows, and the difference current I ₁ -I ₂ becomes the output current I _{in of the} voltage-current conversion circuit 8.

The current mirror circuit 6, 7 passing the current I _1, the same current as I ₂ I _1, I ₂ to the resistor r1, r2. The resistors r1 and r2 convert the currents I ₁ and I ₂ into voltages. That is, the other ends of the resistors r1 and r2 are connected to the output terminal of the voltage amplifier 19. In this case, if r1 = r2 = r3 = R, then V _in = −V _r3 = −I _in · R = (I ₂ −I ₁ ) R. Therefore, the output voltages V1, V2 _{are, V1 = I 1 · R-} V in = I 1 · R + (I 2 -I 1) · R = I 2 · R V2 = I 2 · R-V in = I 2 · the _{R- (I 2 -I 1) ·} R = I 1 · R. Therefore, the characteristics from V _in to V1 and V2 are: V1-V2 = I ₂ · R-I ₁ · R = (I ₂ −I ₁ ) R = V _in V ₁ · V ₂ = I ₁ · I ₂ · R ² − I _D ² · R ² = constant, and the output currents I ₁ , I of the current mirror circuits 6, 7
_This means that the voltages have been converted into the voltages V1 and V2 corresponding to ₂ .
These configurations of the resistors r1 and r2 and the voltage amplifier 19 are
The voltage synthesis circuit 9 is formed.

The voltages V1 and V2 pass through the drive stage transistors Qd and Qd ', and the output stage transistors Qa and Qa'.
Of the output stage transistors Qa, Qa '
From the output current I ₁ of the current mirror circuits 6 and 7,
Current I _{ou t1,} I _out2 that current amplification of I ₂ are output.
The currents I _out1 and I _out2 are combined by the output circuit 21 composed of the resistors Ra and Ra ′ and supplied to the load 10 as the load current I _RL . The characteristics of the currents I _out1 , I _out2 and I _RL are shown in FIG. According to this, since the currents I _out1 and I _out2 are always in a constant direction, the output stage transistors Qa and Qa ′ are not cut off, and the class A operation can be performed in a wide range.

Next, the power supply path to the output stage transistors Qa and Qa 'will be described. Here, only the + side will be described, and the “−” side will be given a “′” in common with the + side and description thereof will be omitted. The main power supply path 42 that mainly supplies power to the load 10 includes a power supply 44 (power supply voltage + B), a switching transistor Qs (switching element), a smoothing coil L1 (smoothing circuit), an output stage transistor Qa, and a resistor Ra (= Ra '). The auxiliary power supply path 48 that supplies the shortage of power supply by the main power supply path 42 is
4, resistor Rs, cascade-connected transistor Qb
(Element for adjusting auxiliary power supply amount), output stage transistor Qa, and resistor Ra.

The resistor Rs is a current I flowing through the auxiliary power line 48.
It is a small resistance for detecting _Qb . R by current I _Qb
The voltage V _Rs generated across s is input to the hysteresis comparator 50. Hysteresis comparator 50
Constitutes a switching control means together with the resistor Rs, and is "0" when the voltage V _Rs becomes higher than a predetermined upper limit V _HCH.
Turning the transistor Qs and output, once "0" after reaching the until the voltage V _Rs drops to a predetermined lower limit value V _HCL holds "0", the voltage V _Rs than the lower limit value V _HCL When it becomes lower, "1" is output to turn off the transistor Qs, and once it becomes "1", "1" is held until the voltage V _Rs becomes higher than a predetermined upper limit value V _HCH . This causes the transistor Qs to perform a switching operation. Smoothing coil L1
Is for smoothing the output waveform of the transistor Qs by this switching operation and supplying it to the load. The diode D3 is a flywheel diode that forms a current path of a current flowing out from the smoothing coil L1 when the transistor Qs is turned off.

A DC power supply 52 is connected between the emitter of the output transistor Qa and the base of the transistor Qb. The power supply voltage V1 is set to V1 = V _Qamin + V _QbBE . Here, V _Qamin is a minimum voltage that can ensure the linearity even when the output transistor Qa is at the maximum output, and is usually about 2 to 3V. V _QbBE is the base-emitter voltage of the transistor Qb and is about 0.6 when on.
It is V. From the emitter of this transistor Qa to the power supply 5
The path 51 that goes through 2 to the base of the transistor Qb corresponds to the auxiliary power supply amount control means.

The operation of this power supply path will be described. First,
It is assumed that the output of the hysteresis comparator 50 is "1" and the switching transistor Qs is in the off state. The audio input signal is supplied to the output stage transistor Qa via the transistor Qd. As a result, the current I _Qb flows in the transistor Qa through the auxiliary power supply path 48 and is supplied to the load 10. This current _IQb causes a voltage _VRs across the resistor Rs. This voltage _VRs is a predetermined upper limit value V
_When reaching _HCH , the output of the hysteresis comparator 50 is inverted to "0" and the transistor Qs is turned on. As a result, current is supplied from the main power supply path 42.

Since the resistance value of the main power supply path 42 is smaller than the resistance value of the auxiliary power supply path 48 when the transistor Qs is on, the current I _L1 from the main power supply path 42 increases once the transistor Qs is turned on. The current I _Qb from the auxiliary power supply path 48 decreases accordingly. The currents I _L1 and I _Qb are the addition points 49.
Is added to the load 10 and is supplied to the load 10 as the load current I _RL via the transistor Qa to supply the necessary power to the load 10.

The voltage V _Rs across the resistor Rs reaches a predetermined lower limit value V _{HCL just} before the current I _Qb of the auxiliary power line 48 becomes 0, the output of the comparator 50 is inverted to "1", and the transistor Qs is turned on. Turn off. When the transistor Qs is turned off, the current I _L1 of the smoothing coil L1 flows through the diode D3. As the current I _L1 gradually decreases, the current I _Qb of the auxiliary power supply path 48 increases, and when the voltage V _Rs reaches a predetermined upper limit value V _HCH , the output of the comparator 50 is again inverted to “0” and the transistor is turned on. Qs is turned on, and the same operation is repeated thereafter. In this way, the transistor Qs is switched by self-excited oscillation so that the ratio of the ON period and the OFF period is variable according to the input signal level.

According to such operation, the transistor Q
b is always in the active state, and the load from the transistor Qa is 1
The current I _RL flowing through 0 is supplied from both the main power supply path 42 and the auxiliary power supply path 48. Transistor Qs of main power supply path 42
Is a switching drive, so the loss is small. Further, although the auxiliary power supply path 48 causes a loss in the transistor Qb, the current I _Qb flowing in the transistor _Qb depends on the setting of the value of the resistor Rs and the reference voltages V _HCH and V _HCL of the hysteresis comparator 50. Since the ripple current of the output current I _L1 can be reduced to almost the same value, I
_{Since L1} >> I _Qb , the loss in the transistor Qb can be reduced.

Since the transistor Qb is always on and V _QbBE is always about 0.6 V, the collector-emitter voltage V _Qa of the output-stage transistor _Qa is always V _Qa = V1-V _QbBE = V _Qamin Kept in. Therefore, the loss of the output stage transistor Qa can be minimized. According to this circuit, the effect of keeping the loss of the output stage transistor Qa, which is the maximum loss source, to a minimum is great. This is particularly effective when a large amount of idle current is passed.

FIG. 8 shows the extracted parts of the transistors Qb and Qs of FIG. The operation of this circuit will be described with reference to FIG. Signal source 54 for transistor Qb
Step-like input V _in falls from (t1), initially the transistor Qs is because off, the output current I _RL is supplied from the transistor Qb for all auxiliary power circuit 48. When the voltage V _Rs across the current I _Qb is once increased to the resistance Rs of the transistor Qb reaches the reference voltage V _HCH upper comparator 50, the output of the comparator 50 turns on the transistor Qs is inverted to "0" .. As a result, power supply from the main power supply path 42 is started. The current I _L1 from the main power supply path 42 does not increase at once because of the smoothing coil L1, but gradually increases. Main current I _L1
There is gradually increased, an auxiliary current I _Qb is gradually reduced by I _Qb = I _RL -I _L1. Auxiliary current I _Qb decreases and voltage V
_{When Rs} reaches the lower reference voltage V _HCL of the comparator 50 (t2), the transistor Qs is turned off and the output current I _L1 of the smoothing coil L1 gradually decreases. Then, the auxiliary current I _Qb increases, the voltage V _Rs reaches the upper reference voltage V _HCH (t3), and the transistor Qs is turned on again, the same operation is repeated thereafter. At the first rising portion (immediately after time t1), the output current I _L1 of the smoothing coil L1 cannot catch up immediately, so the auxiliary current I _Qb is the main, but the input V _in
When the load current I _RL changes little, the main current source 42 supplies power. Music signal because the energy of the high frequency component (signal with a large change) is small, the average value of the auxiliary current I _Qb is small, the load current I _RL
Most of the main power line 4 is highly efficient due to switching drive
2 is supplied as the current I _L1 .

Further, in the circuit of FIG. 1, the load current I _RL is all supplied from the transistor Qa (Qa '), but the collector-emitter voltage V _Qa of the transistor Qa is necessary to turn on the transistor Qa as described above. Since the minimum voltage V _Qamin is applied, the loss here is small. Therefore, the efficiency of the circuit as a whole is very high. Moreover, since electric power is supplied from the auxiliary power supply path 48 which has a quick response to the rapid rising of the input V _in , it is possible to supply the necessary power to the load 10 by following the rapid rising. .. Moreover, the auxiliary power line 48
Is driven by a high voltage + B as in the main power supply path 42, so that a predetermined maximum output can be supplied to the load without clipping the output even when the large-amplitude input rapidly rises. ..

Further, according to the circuit of FIG. 1, an effect in terms of acoustic characteristics can be obtained. That is, since the transistor Qb does not cut off, it is always in a small current but in an active state, and operates so that the ripple component of the main current I _L1 due to switching is absorbed by the current I _Qb from the transistor Qb. Therefore, the voltage supplied to the output-stage transistor Qa has no ripple voltage, the influence of the ripple voltage on the output is eliminated, and the distortion factor and S / N deterioration due to switching drive can be prevented.

In the circuit of FIG. 1, the reference voltage V _HCH on the upper side of the hysteresis comparator 50 is set to a sufficiently small value to reduce the average value of the auxiliary current I _Qb and reduce the loss in the transistor Qb. However, the switching frequency of the transistor Qs becomes high and the switching loss increases. Therefore, it is desirable to set the magnitude of the upper reference voltage V _HCH so as to minimize the sum of the loss in the transistor Qb and the switching loss in the transistor Qs. Further, the lower the reference voltage V _HCL , the smaller the loss as long as the transistor Qb is not cut off. The upper limit of the free-running switching frequency of the transistor Qs can be set to, for example, about 100 kHz when a bipolar transistor is used (the free-running frequency varies depending on the output). When the upper limit is set to 100 kHz or more, a power MOS transistor or the like can be used if switching cannot be performed with a bipolar transistor.

FIG. 10 shows an example of the configuration of the hysteresis comparator 50 shown in FIG. A series connection circuit of resistors R3 and R4 is connected between the + B power supply line 56, the connection point 58 of the transistor Qs and the diode D3. Also,
A transistor Qc, a resistor R2, and a resistor R2 are provided between both ends of the resistor Rs.
The series connection circuit of R1 is connected. And the resistance R
A connection point 60 of 3, R4 is connected to the base of the transistor Qc, and a connection point 62 of the resistors R1, R2 is connected to the base of the switching transistor Qs.

According to the above configuration, the transistor Qc is turned on by the voltage generated across the resistor R3 when the transistor Qs is turned off. At this time, a voltage obtained by dividing the voltage V _Rs by the resistors R1 and R2 is applied to the transistor Qs. In this state, the auxiliary current I _Qb increases and the voltage V _Rs rises.
When the emitter-to-emitter voltage rises to about 0.6 V, the transistor Qs turns on and the main current I _L1 flows. Transistor Qs
When is turned on, the potential at the point 58 rises to approximately + BV, so the potential at the point 60 also rises and the transistor Qc is turned off. When the transistor Qs turns on and the main current I _L1 increases, the auxiliary current I _Qb decreases and the voltage V _Rs decreases. Then, when the transistor Qc is turned off, the voltage _VRs is applied as it is between the base and the emitter of the transistor Qs. Therefore, when the voltage _VRs drops below about 0.6V, the transistor Qs is turned off. When the transistor Qs turns off, the potential at the point 58 drops and the transistor Qc
Turns on.

As described above, according to the hysteresis comparator 50 of FIG. 1, when the transistor Qs is off, a voltage obtained by dividing the voltage V _Rs by the resistors R1 and R2 is applied between the base and the emitter of the transistor Qs. The transistor Qs is turned on only when the voltage V _Rs rises to a voltage that causes a voltage of about 0.6 V or more across the resistor R2. When the transistor Qs is on, the voltage V _Rs is applied as it is between the base and the emitter of the transistor Qs, and when the voltage V _Rs becomes about 0.6 V or less, the transistor Qs is turned on.
s turns off. As a result, a comparator operation with hysteresis is realized, and the transistor Qs performs a switching operation by self-oscillation.

The lower reference voltage V _HCL is set by the value of the resistor Rs. Further, the upper reference voltage V _HCH is set by the ratio of the values of the resistors R1 and R2, and V _HCH increases as the value of the resistor R1 becomes larger than that of the resistor R2.

The hysteresis comparator 50 is not limited to the configuration shown in FIG. 10, but may be variously constructed using an operational amplifier or the like. However, the configuration as shown in FIG. 10 has an advantage that it can be easily realized such that a separate power supply is not required for the hysteresis comparator 50.

The switching of the power supply path to the load 10 is performed by detecting the auxiliary current value and the main power supply path 42.
It is also possible to switch the voltage depending on the magnitude of the voltage of the above and the voltage of the auxiliary power supply path 48. That is, the power supply is performed from the main power supply path 42 with high efficiency so that the output voltage of the main power supply path 42 is normally higher, and when the input signal rises rapidly, the output voltage of the main power supply path 42 is increased. Cannot catch up suddenly, so make the output voltage of the auxiliary power line 48 higher.
The transistor Qb is turned on to supply power from the auxiliary power supply path 48 with high speed response.

The output stage transistors Qa and Q shown in FIG.
The power supply path to a'can be conceptually grasped as shown in FIG. That is, a high-efficiency current buffer circuit (switching transistor 42) that can supply current to the main power supply path 42 with high efficiency but has a slow response to a change in input signal.
And a smoothing coil L1) and a high-speed voltage buffer circuit (transistor Qb and a path 51 for controlling the base voltage thereof) which has a fast response to a change in the input signal but a large loss is inserted in the auxiliary power supply path 48. The currents I _L1 and I _Qb of the power supply paths 42 and 48 are added at the addition point 49 to obtain the transistor Q.
The load current I _RL (= I _L1 + I _Qb ) is supplied to the load 10 via a. The current detection circuit (resistor Rs) detects the magnitude of the auxiliary current IQb, and accordingly the high-efficiency current buffer circuit 4 via the control circuit (hysteresis comparator 50).
2. L1 is controlled to adjust the main current I _L1 . Insufficiency at the high speed voltage buffer circuit Qb, 51 is the main current I _L1 (I _RL -
I _L1 ) is supplied as the auxiliary current IQb.

The voltage V at the collector of the output transistor Qa
s is only slightly larger than the output voltage Vout,
This minimizes the loss of the output transistor Qa. In addition, the load current I _RL responds to the abrupt rising of the input by the auxiliary current I _Qb , and gradually the main current I _RL.
L1 becomes dominant. Since the auxiliary current I _Qb rises at an extremely high speed, no loss occurs in the output waveform. The auxiliary current I _Qb is supplied only for a short time, and the energy is mainly supplied by the main current I _L1 from the main power supply path 42 with a small loss, so that the efficiency is high as a whole. As a result, a power amplifier circuit with high efficiency and fast slew rate is realized.

Next, a concrete circuit of the circuit shown in FIG.
Divided into 5 and shown. 12 to 15 show A1 to A8 and B1
B5 and E1 to E6 are connected to each other. The same reference numerals are used for the portions corresponding to those in FIG. It also corresponds to the + side.
The part on the side is shown by adding "'". The configuration of the power supply path to the output stage transistors Qa and Qa 'will be described with respect to the + side circuit (FIG. 14), and the − side circuit (FIG. 15) will be described because it is configured similarly to the + side circuit. Is omitted.

The input signal is input to the transistor Q30 which constitutes the differential amplifier circuit at the first stage in FIG. 12, and is conducted to the transistors Qg and Qg 'via the transistors Q32 and Q32' and the transistors Qf and Qf 'in FIG. Get burned. Then, the signals are output from the transistors Qg and Qg ′, and are output as currents I ₁ and I ₂ from the transistors Tr1 and Tr3 of the current-current conversion circuit 3 via the current mirror circuit 6 via the capacitor Cf and the resistor r3.
The currents I ₁ and I ₂ are converted into voltages by the resistors r1 and r2, and the transistors Q10, Q15 and Q1 forming the drive stage of FIGS.
Output stage transistors Qa, Qa 'through 0', Q15 '
To drive the load (speaker) 10. Further, the output signal is negatively fed back to the transistor Q31 (FIG. 12) and the current-current conversion circuit 3 (FIG. 13) in the first stage.

In FIG. 14, the output stage transistor Qa
And the cascade transistor Qb each have a parallel configuration including three transistors Q23 and Q20. The main power supply path 42 supplies the main current to the output stage transistor Qa via the + 90V power supply → the switching transistor Qs → the coil L1 → the current addition point 49. Further, the auxiliary power supply path 48 supplies an auxiliary current to the output stage transistor Qa via the resistors R10, R11, R12 and the current detection limiter 90 → transistor Qb. The path 51 of FIG. 1 for controlling the base potential of the transistor Qb is, in FIG. 14, the path of the emitter of the transistor Q10 → the Zener diode ZD → the transistor Q11 → the resistor R13 → the diode D10 → the transistor Q14 → the resistor R14 → the base of the transistor Qb. Equivalent to. As a result, when the input rises rapidly, the base potential of the transistor Qb immediately rises to increase and follow the auxiliary current, and then the main current gradually increases and the auxiliary current decreases. I will let you.

The hysteresis comparator 50 is constructed as follows. A series connection circuit of resistors R33 and R34 is connected between the + 90V power supply line 56, the connection point 58 of the transistor Qs (Q22) and the diode D3. In addition, the auxiliary power supply path 48 includes resistors R10 and R12.
Route through, transistor Qc, resistors R11, R1
Routes passing through 2 are provided in parallel. The connection point 60 of the resistors R33 and R34 is connected to the base of the transistor Qc, the connection point 62 of the resistors R11 and R12 is connected to the base of the transistor Q21, and the transistor Q2.
The collector of 1 is connected to the base of the switching transistor Qs via the resistor R21.

According to the above configuration, the transistor Q21
Is off (transistor Qs is off), the transistor Qc is on due to the voltage generated across the resistor R33. At this time, the auxiliary power supply path 48 has a route passing through the resistor R10, the transistor Qc, and the resistor R11.
Routes passing through are formed in parallel, and the voltage across the resistor R10 is applied between the base and emitter of the transistor Q21. In this state, the auxiliary power supply path 4 passing through the resistor R10
8 increases and the base-emitter voltage of the transistor Q21 rises to about 0.6V, the transistor 21
Is turned on, the transistor Qs is turned on, and current is supplied from the main power supply path 42. When the transistor Q21 is turned on, the potential at the point 58 rises to about +90 V, so that the point 6
The potential of 0 also rises and the transistor Qc turns off. When transistor Q21 turns on and the current in main power path 42 increases, the current in auxiliary power path 48 decreases. When the transistor Qc is off, the auxiliary power supply path 48 has only a route passing through the resistor R10. Therefore, all the current in the auxiliary power supply path 48 flows through the resistor R10, and the voltage across the resistor R10 is about 0.6V or less. Transistor Q
21 turns off. When the transistor Q21 turns off, the potential at the point 58 decreases and the transistor Qc turns on.

In this way, depending on the on / off state of the transistor Qc, it is switched whether all the current of the auxiliary power supply path 48 is made to flow through all the resistors R10 or a part thereof is shunted, so that a comparator operation having hysteresis can be performed. When realized, the transistor Qs performs switching operation by self-oscillation.

The output stage transistors Qa and Q shown in FIG.
The following measures have been taken in the power supply path to a '.
The output stage transistor Qa and the cascade transistor Qb have a parallel configuration. In accordance with this, the smoothing coil L1 is also wound in parallel. With this configuration, the output is increased and the emitter of the cascade transistor Qb has a stabilizing resistor (resistance value 0.22Ω).
Etc.) is unnecessary, loss due to this can be prevented, and high efficiency can be maintained. That is, when the smoothing coil L1 is configured by a single winding, the number of current injection points 49 is one, and the emitters of the cascade transistors Qb are inevitably connected in common, so that the operating currents of the plurality of cascade transistors Qb are determined. In order to stabilize as expected, a resistance of 0.22Ω or the like is required for each emitter. On the other hand, when the smoothing coil L1 is wound in parallel, there are two current injection points 49, and there is a small amount of resistance in the coil winding between them, so the operation of each cascade transistor Qb is reduced. Since the mutual interference at is reduced and the problem of stabilization is reduced, the cascade transistor Qb does not need an emitter resistor, and the loss is reduced accordingly.

Between the output of the smoothing coil L1 and the power supply line 56, resistors R15 and R16 and a capacitor C10 are provided.
The ripple absorption circuit 92 (low-pass filter) configured by is connected. As a result, a part of the ripple of the main current due to switching is released to the AC common potential point (power supply line 56), the burden of the operating current of the cascade transistor Qb is alleviated, and the loss increase of this transistor Qb is prevented. ..

Current detection resistors R10, R11, R12
(When the switching transistor Qs is off (transistor Qc is on), the DC resistance of the parallel resistance of the resistors R10 and R11 and the resistor R12, and when the switching transistor Qs is on (transistor Qc is off), the resistance R10 is shown. , R12 series resistance), and a diode D11,
The current detection limiter 90 composed of D12 is connected in parallel. As a result, the upper limit of the current detection voltage transmitted to the transistor Qs side is limited to 1.2V, overdrive between the base and emitter of the transistor Qs is prevented, and current can be supplied from the current detection limiter 90 to the cascade transistor Qb. To improve the current supply capability of the cascade transistor Qb (the current path of the transistor Qb is the resistor Rs (FIG. 1) and the resistor R1).
If the current is regulated by 0, R11, and R12, a large current cannot flow. ).

A hold circuit 94 including a capacitor C11 and a diode D10 is inserted in the middle of a path for linking the base potential of the cascade transistor Qb with the emitter potential (output potential) of the output stage transistor Qa. As a result, the high-frequency signal is subjected to peak detection and the voltage is held in the capacitor C11, so that the base potential of the cascade transistor Qb is made a constant voltage,
The emitter output voltage of the cascade transistor Qb is prevented from becoming insufficient for the output stage transistor Qa due to a phase delay or the like.

Here, the collector-emitter voltage of the output stage transistor Qa is set to about 2V. That is, from the upper end potential of the Zener diode ZD, between the base and the emitter of the transistor Q11, between the base and the emitter of the diode D10, the transistor Q14, and the transistor Q.
4 base-emitter diode junctions (0.
6 × 4 = 2.4 V), the collector potential of the output stage transistor Qa is determined. Further, from the lower end potential of the Zener diode ZD to the base-emitter of the transistor Q15,
The emitter potential of the output stage transistor Qa is determined via two diode junctions (0.6 × 2 = 1.2V) between the base and emitter of the transistor Qa. That is, (3.
2−2.4) − (0−1.2) = 2.0V is the collector-emitter voltage of the output stage transistor Qa.

A switching frequency fine adjustment circuit 96 including a capacitor C12 is inserted in parallel in the current value detection system of the auxiliary current. The switching frequency may become too high (for example, 100 kH or more) when the switching free-running system including the hysteresis comparator 50 and the switching transistor Qs is optimized in design. Can be reduced, which improves design arbitrariness.

Of the power parts shown in FIGS. 14 and 15, for example, three parts surrounded by alternate long and short dash lines 111, 112 and 113 can be respectively constituted by power ICs (hybrid ICs). The power IC 111 includes the output side transistors Qa and Qa '(Q23, Q23') on the + side and the-side, and constitutes an output stage. The power IC 112 includes a + side cascade transistor Qb (Q20) and a switching transistor Qs (Q22), and constitutes a + side cascade stage and a switching stage. The power IC 113 is a minus side cascade transistor Qb '.
(Q20 ') and switching transistor Qs' (Q
22 '), which constitutes the-side cascade stage and switching stage.

As described above, a part of the power section is used as the power IC.
With the configuration, the following effects can be obtained.
Since the output stage is composed of the independent power IC 111, it is not easily affected by switching noise. Also,
This power IC111 can be used as a general-purpose output stage,
It can be compatible with conventional output stages. Since the cascade stage and the switching stage are combined to form one power IC 112 (113), the size is reduced. The power ICs 111 to 113 are easy to manufacture because they are composed almost entirely of resistors and semiconductors. By externally attaching all or part of the control unit of the circuit, adjustment is facilitated and versatility of the IC is ensured. That is, the bias circuit ZD peripheral 4 in the cascade stage
0 and the hysteresis circuit 50 of the switching stage and the like have constants that differ depending on the application, so by incorporating all or part of them externally and incorporating the part that does not change much into the IC,
Versatility is secured.

The circuit of FIG. 13 can also be integrated into an IC. In the circuit of FIG. 13, a portion of the current-voltage conversion circuit 3 and the current mirror circuits 6 and 7 which require temperature compensation,
If the resistors r1, r2, r3 that determine the output stage drive gain and the like are formed on the same IC, variations in characteristics can be improved. FIG. 16 shows an example in which the block indicated by the dotted line in FIG. 13 is integrated. Note that the IC pin numbers used in FIG. 16 are given in FIG. Circuits with external pins other than IC have semi-fixed resistors (2
0k) including an idling current fine adjustment circuit ID, transistors Qh and Qh ', resistors r1, r2 and r3, and a portion Qc for externally adjusting the respective values of the voltage dividing resistors around them.
And the voltage stabilizing circuit Vc. In the above-mentioned embodiment, the whole is driven by the common power supply voltages + B and -B, but it is also possible to drive by different necessary and sufficient power supply voltages for each part as shown in FIG.

[0058]

As described above, according to the present invention, the drive voltage of the output stage element is lowered to the voltage corresponding to the input signal level by the main power supply path which is switching-driven, so that when the input is zero. The applied voltage of the output stage element of becomes small, and even if the idle current of the output stage element is set to a large value,
The loss (heat generation) here is reduced, and the efficiency can be improved. Moreover, since the main power supply path itself is switching-driven, the loss in the main power supply path is small. Also, in case of abrupt rising of the input that cannot be followed by the main power supply line due to the presence of the smoothing circuit, power is supplied from the auxiliary power supply line which has better response than the main power supply line. It can follow the sudden rise of the amplitude input and does not clip the output. Even if the auxiliary power supply line is driven with such a power supply voltage, the power is usually supplied mainly from the main power supply line, so the average current value flowing through the auxiliary power supply line can be small. Losses on the road are small. As a result, an amplifier circuit having high performance, high efficiency, and good followability even for a sharp rise of a large amplitude input is realized.

[Brief description of drawings]

FIG. 1 is a circuit diagram showing an embodiment of the present invention.

FIG. 2 is a circuit diagram showing a conventional class A push-pull amplifier.

3 is an operating characteristic diagram of the circuit of FIG.

FIG. 4 is a circuit diagram of a current-current conversion circuit used in the present invention.

5 is an operating characteristic diagram of the circuit of FIG.

6 is a circuit diagram showing a conventional amplifier circuit using the circuit of FIG.

7 is a characteristic diagram of I _RL of currents I _1out and I _2out in FIG. 1. FIG.

8 is a circuit diagram of FIG. 1 showing transistors Qb and Qs.
It is the figure which extracted and showed the part.

9 is an operation waveform diagram of the circuit of FIG.

FIG. 10 is a hysteresis comparator 5 in FIG.
It is a circuit diagram which shows the specific example of 0.

11 is a block diagram conceptually capturing a power supply path to transistors Qa and Qa ′ in the circuit of FIG. 1. FIG.

12 is a circuit diagram showing a specific example of the circuit shown in FIG.
Some of them constitute an amplifier circuit together with the circuits 3 to 15.

13 is a circuit diagram showing a specific example of the circuit shown in FIG.
2, an amplifier circuit is configured with the circuits of FIGS.

14 is a circuit diagram showing a specific example of the circuit shown in FIG.
2, an amplifier circuit is configured with the circuits of FIGS.

15 is a circuit diagram showing a specific example of the circuit shown in FIG.
An amplifier circuit is configured with the circuits of FIGS.

FIG. 16 is a circuit diagram in which FIG. 13 is partially integrated into an IC.

[Explanation of symbols]

3 current-current conversion circuit 6 first current mirror circuit 7 second current mirror circuit 8 voltage-current conversion circuit 9 voltage synthesis circuit 10 load 21 output circuit 42 main power supply path 44, 44 'power supply 48 auxiliary power supply path 49, 51 Addition point, signal path (auxiliary power supply amount control means) Rs, 50 Auxiliary current detection resistance, hysteresis comparator (switching control means) L1 Smoothing coil (smoothing circuit) Qs Switching transistor (switching element) Qb transistor (auxiliary power) Supply amount adjusting element) Tr1 first transistor Tr2 second transistor Tr3 third transistor Tr4 fourth transistor

Claims (1)

[Claims] 1. A voltage-current conversion circuit for converting an input signal to be amplified into a voltage-current, and both emitters of a first transistor (NPN) and a third transistor (PNP) are commonly connected to each other and a collector thereof. Both emitters of a second transistor (NPN) and a fourth transistor (PNP) whose bases are commonly connected are commonly connected, both bases of the first and second transistors are commonly connected, and the third and third transistors are connected together. The bases of the four transistors are commonly connected, the output current of the voltage-current converter is supplied to the common emitters of the first and third transistors, and a constant current is supplied to the second and fourth transistors, respectively. A current-current conversion circuit configured to supply the current, a first current mirror circuit that outputs a current proportional to the collector current of the first transistor, A second current mirror circuit that outputs a current proportional to the collector current of the third transistor, and a voltage signal corresponding to the output currents of the first and second current mirror circuits and the input signal to be amplified are combined. And a voltage synthesizing circuit that outputs the voltage, an output stage element that amplifies the output voltage of the voltage synthesizing circuit, an output stage element that current-amplifies the output currents of the first and second current mirror circuits, and these output stages. An output circuit for combining the output currents of the elements and supplying it to the load, a main power supply path for supplying driving power to the output stage element, and a main power supply path that is inserted into the main power supply path to switch the main power supply path on and off. A switching element; a smoothing circuit that is inserted into the main power supply path to smooth the output of the switching element and supply the output to the load; and the smoothing circuit according to the level of the input signal. The switching element ratio of the ON period and the OFF period by variably switching,
Switching control means for mainly supplying the power required by the load from the main power supply path, and a power supply path for supplying drive power to the output stage element separately from the main power supply path, Also, an auxiliary power supply line capable of supplying power by following a level change of an input signal with a high-speed response, and an auxiliary power supply line inserted in the auxiliary power supply line in a cascade connection with the output stage element. An amplifier circuit comprising: an auxiliary power supply amount adjustment element for adjusting the power supply amount to the load.