JPH0582154B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION [Technical Field of the Invention] The present invention relates to a current control device that performs feedback control so that the output current of a voltage-type three-phase power converter follows its current command value.

[Technical background of the invention and its problems] Typical voltage-type power conversion devices include sine wave cycloconverters, transistors, and gate turn-off thyristor (GTO) inverters, and devices that drive AC motors as loads are widely known.

When an AC motor is driven by a voltage type power converter that requires quick response, the output current is detected and current feedback control is performed.
This current control amplifies the deviation between the current command value of each phase and the detected current value of each phase through proportional + integral elements, etc., and uses the amplified signal as the output voltage command value of each phase to control the power converter. This is achieved through control. This has been done for a long time and is well known, but one example is
A current control device using a transistor inverter described in Japanese Patent No. 117577 is shown in FIG. 2 and will be briefly described.

In Fig. 2, 1 is a DC current, 2 is an inverter, 3 is an AC motor, 4U, 4V, 4W is a current detector, 5U, 5V, 5W is a subtractor, 6U, 6V,
6W is an amplifier, 7 is a triangular wave generator, 8U, 8V,
8W is a comparator, and 9U, 9V, and 9W are base drive circuits.

Three-phase AC power of variable frequency and variable voltage is supplied from a DC current 1 to an AC motor 3 via an inverter 2 that is controlled by PWM (pulse width modulation). Currents of each phase supplied to this AC motor 3 i _U , i _V ,
i _W is detected by the current detector at 4U, 4V, 4W,
Subtractors 5U, 5V, and 5W calculate deviations from i _U ^* , i _V ^* , and i _W ^* in the current command values of each phase, respectively.
The deviation output of subtractor 5U, 5V, 5W is output from amplifier 6.
It is amplified by U, 6V, and 6W, and outputs voltage command values υ _U ^* , υ _V ^* , υ _W ^* for the output phase voltage of the inverter 2. The output voltage υ _T of the triangular wave generator 7 and the voltage command values υ _U ^* , υ _V ^* , υ _W ^* are determined by comparators 8U and 8, respectively.
V, 8W are compared, a PWM signal is output, and the base drive circuit 9U, 9V, 9W drives the transistor of the inverter 2 to drive the AC motor 3.
Apply three-phase voltages υ _U , υ _V , and υ _W to .

With this configuration, the terminal voltages υ _U , υ _V , υ _W of the AC motor 2 include modulation ripples, but on average have a magnitude proportional to the voltage command value that is the output of the amplifiers 6U to 6W. This voltage command value υ _U ^* , υ _V
^* , υ _W ^* is the subtracter 5U ~ 5W and the amplifier 6U ~
The signal is amplified by the deviation between the current command value and the detected value of each phase due to the 6W operation, so the current of each phase is
i _U , i _V , and i _W are controlled to follow the command values and i _U ^* , i _V ^* , and i _W ^* .

In the conventional current control device configured in this way,
If the components of the control circuit such as current detectors 4U to 4W, subtractors 5U to 5W, or amplifiers 6U to 6W are ideal and do not have offset factors such as temperature drift, the output of amplifiers 6U to 6W is 3.
The phase voltage command values υ _U ^* , υ _V ^* , and υ _W ^* have waveforms that are symmetrical between positive and negative and balanced among the three phases.

However, in reality, it is almost impossible to accept the aforementioned offset factors, and it is more realistic to assume that some kind of offset factor exists.

Before discussing the problems of the conventional device when such an offset factor occurs, let us consider the relationship between the voltage and current of the AC motor 3.

Figure 3 is an armature model of AC motor 3,
U, V, W are the three-phase winding terminals, N is the neutral point of the three-phase winding, υ _U , υ _V , υ _W are the three-phase winding terminals U, V,
The potential of W, υ _N , is the potential of the neutral point N.

When the impedance of each phase winding is Z, the currents i _U , i _V , i _W of each phase are given by the following equations.

i _U =υ _U −υ _N /Z,i _V =υ _V −υ _N /Z,i _W =υ _W −υ _N
/Z...Equation (1) Since the sum of the three-phase currents i _U , i _V , and i _W is 0,
By adding the equations for the three phases in equation (1), the relationship between the neutral point potential υ _N and each phase terminal potential υ _U , υ _V , υ _W can be determined as shown in the following equation.

υ _N = υ _U + υ _V + υ _W /3...Equation (2) By substituting Equation (2) for υ _N in Equation (1), each phase current can be written as follows.

i _U =υ _U −υ _V +υ _U −υ _W /3Z=υ _UV −υ _WU /3Z i _V =υ _V −υ _W +υ _V −υ _U /3Z=υ _VW −υ _UV /3Z i _W =υ _W −υ _U +υ _W −υ _V /3Z = υ _WU −υ _VW /3Z ...(3) In equation (3), υ _UV is the potential difference between the U-phase terminal potential υ _U and the V-phase terminal potential υ _V , i.e. This is the line voltage between U and V. Similarly, υ _VW and υ _WU are the line voltages between V and W and between W and U, respectively.

Equation (3) shows that each phase current of the AC motor 3 is determined by the line voltage between each winding. In other words, the magnitude of each phase current is determined by the potential difference between each winding terminal, and the potentials υ _U , υ _V , υ _W and neutral point potential υ _N of each winding terminal are factors that determine the current magnitude. It means not to be.

Now, let's explain the problems in conventional current control by taking as an example a case where a negative DC component is superimposed on the _U -phase detection current iU due to the offset of the current detector 4U in the conventional device shown in Fig. 2. .

The negative DC component of the U-phase current i _U is amplified by the amplifier 6U, causing the voltage command value υ _U ^* to shift to the positive side. When the U-phase voltage command value υ _U ^* shifts to the positive side, each phase current is determined by the line voltage as described above, so as a result of current control, the voltage command values of other phases υ _V ^* , υ _W
^* also shifts to the positive side. Therefore, amplifier 6U~6W
Current control can be performed with the outputs υ _U ^* , υ _V ^* , υ _W ^* all moving to the positive side, that is, the neutral point current υ _N of the AC motor 3 moving to the positive side.

Normally, since the amplifiers 6U to 6W have an integral element, even an extremely small offset factor is integrated, and the amplifier output changes to one of the polarities. If only one phase changes, the other phases will also change to the same polarity side.

Even if such a potential transition phenomenon occurs, current control is performed because the motor phase currents i _U , i _V , i _W do not exist at the neutral point potential as described above.

However, the fact that the neutral point potential changes easily has the following drawbacks.

If PWM control is performed by comparing the voltage command value at which the neutral point potential changes with the triangular wave, the symmetry of the PWM waveform every 180 degrees within one cycle of 360 degrees disappears. As a result, the ripple component caused by PWM included in the current waveform also becomes asymmetric between positive and negative, resulting in increased noise.

Since there is no control element that determines the neutral point current, the neutral point potential easily fluctuates even due to small disturbance factors such as the ripple component of the feedback current.
Easily affected by external disturbances.

Even when outputting a low voltage, the amplifier output operates in the limit value region depending on the amount of transition of the neutral point potential. When the voltage command value of one phase starts to apply to this limit value or when it comes out of the limit value area, the potential change rate of that phase changes suddenly, which causes a disturbance to the current control of other phases.
The current waveform is disturbed.

The drawbacks of the conventional device shown in FIG. 2 described above are due to the control structure having no element for determining the neutral point potential υ _N of the AC motor 3.

Therefore, the problem can be solved by configuring the neutral point potential υ _N to be 0, or in other words, by configuring the sum of the three-phase current command values υ _U ^* , υ _V ^* , and υ _W ^* to be 0. be.
The configuration shown in FIG. 4 can be considered as one of the solutions.

Components in FIG. 4 having the same reference numerals as those in FIG. 2 have the same functions.

In the configuration shown in Figure 4, current control is performed in two phases: U phase and V phase.
The voltage command value υ _W ^* of the W phase is converted to the voltage command values υ _U ^* , υ _V ^* of the other two phases by the inverting adder 10.
is obtained by inverting the polarity and adding it, that is, by calculating the following equation.

υ _W ^* = −υ _U ^* −υ _V ^* …Equation (4) Therefore, since the sum of the three-phase voltage command values is always 0, the neutral point potential υ _N of the AC motor 3 is also always 0. Therefore, the drawbacks shown in FIG. 2 do not occur.

Further, since the sum of the three-phase currents is 0, by controlling the currents of two of the three phases, the three-phase currents are controlled as a result.

However, this configuration of the device has other drawbacks.
Its drawbacks will be explained next.

Normally, as the operating frequency increases, the induced voltage of the AC motor 3 increases, and the inverter 2 that supplies current is also required to output a voltage with a large amplitude.
Even at the maximum voltage output, it is possible to operate the voltage command value output from the amplifiers 6U and 6V within the positive and negative limit values by increasing the voltage value of the DC power supply 1 to provide a margin.

However, increasing the voltage of the DC power supply 1 causes the following adverse effects.

This increases the ripple component of the current due to PWM control, which also increases noise.

Requires high voltage inverter elements.

The capacity of the DC power supply increases.

For these reasons, in the operating region near the maximum voltage output, the voltage command value is generally operated while being limited by a limit value, and the voltage value of the DC power supply 1 is generally lowered to the minimum necessary level.

Figure 5 shows the three-phase voltage command values υ _U ^* , υ _V ^* , υ _W ^* when the configuration shown in Figure 4 is operated in the voltage limit value region.
(a) is the U-phase voltage command value υ _U ^* which is the output of amplifier 6U, (b) is the V-phase voltage command value υ _V ^* which is the output of amplifier 6V, and (c) is the inverted addition. This is the W-phase voltage command value υ _W ^* which is the output of the device 10. In each waveform, dotted lines represent positive and negative limit voltage values.

As is clear from this figure, the voltage command value υ _W ^* of the W phase has a different waveform from the command values υ _U ^* and υ _V ^* of the other two phases, and the AC motor 3 has an unbalanced waveform between the three phases. A waveform voltage is applied. As a result, the phase current supplied to the AC motor 3 also becomes unbalanced among the three phases,
This causes torque ripple and noise.

As described above, the configuration shown in FIG. 4 has a disadvantage that when the voltage operates in the limit value region, the voltage waveform becomes unbalanced among the three phases.

[Object of the Invention] The present invention was made in view of the drawbacks of the conventional devices as explained above, and provides a current control device that can fix the neutral point potential of the load to 0 even though it is a three-phase current control device. is intended to provide.

[Summary of the Invention] In order to achieve the above object, the present invention adds the current control outputs of each phase for three phases, and subtracts a signal obtained by 1/3 of the added value to the current control output of each phase. The voltage command value shall be As a result, the sum of the three-phase voltage command values becomes 0, and the neutral point potential of the load also becomes 0.

[Embodiments of the Invention] The present invention will be described in more detail below with reference to the drawings.

FIG. 1 is a block diagram showing the configuration of an embodiment of the present invention. Components having the same reference numerals as those in FIG. 2 have the same functions, so a description thereof will be omitted.

In Fig. 1, compared to the configuration of the conventional device shown in Fig. 2,
Adder 11, coefficient unit 12, and subtracter 13U,
13V and 13W are added.

The outputs υ _U ′, υ _V ′, υ _W ′ of the amplifiers 6U to 6W are added by an adder 11, and the added voltage signal υ ₀ is multiplied by 1/3 by a coefficient multiplier 12. The output signal of the coefficient unit 12 is subtracted from the output of the current control amplifier of each phase by subtracters 13U, 13V, and 13W, respectively, and becomes voltage command values υ _U ^* , υ _V ^* , υ _W ^* of each phase. .

Three-phase current-controlled amplifier output υ _U ′, υ _V ′,
When υ _W ' is symmetrical in positive and negative directions and balanced among the three phases, the output υ ₀ of the adder 11 is 0.
The output of the coefficient unit 12 also becomes 0, and the subtracters 13U to 1
The negative side input of 3W is 0. Therefore, in such a case, the current control is performed in the same way as in the conventional device shown in FIG.

Next, the operation in the case where the U-phase detected current i _U includes a negative DC component will be described.

Since the U-phase detection current i _U includes a negative DC component, the output υ _U ′ of the U-phase current control amplifier 6U tends to shift to the positive side by Δυ _U due to the amplification effect of the amplifier 6U. . The outputs of the current-controlled amplifiers υ _U ′, υ _V ′, υ _W ′ for each phase are υ _U0 ′, υ _V0 ′, υ _W0 ′, and υ _U0 ′+Δυ _V , υ _V
0 ′,
υ _W0 ′. Therefore, the output υ ₀ of the adder 11, which is the sum of the outputs of the three-phase current-controlled amplifiers, is expressed by equation (5).

υ ₀ = υ _U0 ′+υ _V0 ′+υ _W0 ′+Δυ _U = Δ _U …(5) Because the sum of the equilibrium components (υ _U0 ′+υ _V0 ′+υ _W0 ′) is 0. The voltage command values for each phase υ _U ^* , υ _V ^* , υ _W ^* are
A signal obtained by multiplying the output υ ₀ (=Δυ _U ) of the adder 11 by 1/3 from the output υ _U ′, υ _V ′, υ _W ′ of the current control amplifier of each phase using the coefficient multiplier 12 is obtained by the subtracter 13U, 13V. ,13W
It is used after being subtracted by . Voltage command value υ _U ^* ,
υ _V ^* and υ _W ^* are expressed by equations (6), (7), and (8), respectively.

υ _U ^* =υ _U0 ′+Δυ _U −1/3Δυ _U =υ _U0 ′+2/3Δυ _U …Equation (6) υ _V ^* =υ _V0 ′−1/3Δυ _U …Equation (7) υ _W ^* =υ _V0 '-1/3Δυ _U ...Equation (8) Therefore, the sum of the three-phase voltage command values is always 0 as shown in Equation (9).

υ _U ^* +υ _V ^* +υ _W ^* =υ _U0 ′+υ _V0 ′+υ _W0 ′=0…(9)
In other words, in the configuration of FIG. 1 of the present invention, even if there is an external offset factor, the sum of the three-phase voltage command values υ _U ^* , υ _V ^* , υ _W ^* always becomes 0. . Therefore, the neutral point potential υ _N of the AC motor 3, which is the load, is also controlled to always be zero.

As a result, unlike in the case of the conventional device shown in FIG. 2, there is no runner-up due to the undetermined neutral point potential.

Furthermore, since the control circuit has exactly the same configuration among the three phases, even when the voltage is operating in the limit value region, the waveform of the limited voltage is the same for all three phases, and the two It also does not cause current imbalance and torque ripple like phase current control.

In Figure 1, three current detectors 4U, 4V, and 4W are used to detect the current for three phases.
Using the relationship that the sum of the three phase currents is 0,
It is also possible to detect only two-phase currents and use a signal obtained by calculating the remaining one-phase current from the detected two-phase currents.

Although the present invention has been described above as an example in which the present invention is applied to a PWM-controlled transistor inverter as a power conversion device, it can be similarly applied even if the power conversion device is a GTO inverter.

In addition, in the case of a sine wave cycloconverter, the triangular wave generator 7 and the comparators 8U, 8V, 8W shown in FIG.
and base drive circuit 9U, 9V, 9W PWM
Although the control section is replaced with a phase control circuit, its current control can be applied with the same configuration as in FIG. 1. That is, the present invention is applicable to a voltage type three-phase power converter that performs feedback control of output current.

[Effects of the Invention] As described above, the present invention is controlled so that the sum of the three-phase voltage command values becomes 0, so stable current control can be achieved even with circuit offset factors and disturbance elements. This makes it possible to operate the device while maintaining three-phase balance even in a region where the output voltage is limited, and it is possible to provide a device with excellent performance and economical efficiency.

[Brief explanation of the drawing]

FIG. 1 is a circuit block diagram showing the configuration of an embodiment of the present invention, FIG. 2 is a block diagram showing an example of the configuration of a conventional device, and FIG. 3 is an armature model diagram of an AC motor.
FIG. 4 is a circuit configuration diagram showing one method for solving the drawbacks of the conventional device shown in FIG. 3, and FIG. 5 is a voltage waveform diagram for explaining the drawbacks of the device shown in FIG. 1...DC power supply, 2...Inverter, 3...AC motor, 4U, 4V, 4W...Current detector, 5
U, 5V, 5W...subtractor, 6U, 6V, 6W...
...Current control amplifier, 7...Triangular wave generator, 8U,
8V, 8W... Comparator, 9U, 9V, 9W... Base drive circuit, 11... Adder, 12... Coefficient unit, 13U, 13V, 13W... Subtractor.

Claims (1)

[Claims] 1 A control amplifier is installed in each phase to detect the output current of each phase of a voltage-type three-phase power converter that outputs an AC voltage that follows a voltage command value, and to amplify the deviation of the detected current for each phase with respect to the current command value. In a current control device for a power conversion device, which is provided for each phase and forms a feedback loop by determining a voltage command value for each phase of the power conversion device from the output of a control amplifier thereof, the current control device controls each phase. The power converter includes an adder that adds the outputs of the amplifiers, a coefficient multiplier that multiplies the output of the adder by a coefficient, and a subtracter that subtracts the output of the coefficient multiplier from the output of the control amplifier of each phase. A current control device for a power conversion device, characterized in that a voltage command value of each phase of the device is used as an output of the subtracter.