JPH0687674B2 - Induction motor speed / flux control device - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION [Industrial application] The present invention relates to an induction motor for operating a voltage and a frequency of the voltage source by connecting a voltage source of a variable voltage and a variable frequency to a stator winding of the induction motor. The present invention relates to a speed / flux control device.

[Conventional technology]

As is well known, today's speed / flux control devices for induction motors are dominated by the so-called "vector control" control method. In Japan, this "vector control" was held at a symposium at the National Conference of the Institute of Electrical Engineers of Japan in 1983. Is summarized. FIG. 2 is a diagram showing an example of a basic circuit configuration of “slip frequency type vector control” summarized and generally discussed in this symposium. In the figure, reference numeral 1 denotes a variable voltage, variable frequency power converter such as a cycloconverter, which converts the AC power of a normal power supply system into a desired voltage and frequency and supplies it to a stator winding of an induction motor 2. . Reference numeral 3 is a rotor angular velocity detector of the induction motor 2. Reference numeral 4 is a current detector for detecting three-phase alternating currents I _1U , I _1V , I _1W flowing into the stator winding, and 5 is a three-phase to two-phase converter. Values in a two-axis rotating coordinate system (dq coordinate system) that rotates the phase alternating currents I _1U , I _1V , I _1W in synchronization with the frequency w _{1 of the} alternating voltage applied to the stator windings, That is, the stator winding current I ₁ d,
Convert to I ₁ q. 6 is the stator winding current I in the dq coordinate system
A magnetic flux calculator that calculates the magnetic flux Φ ₂ d, Φ ₂ q that links the rotor from ₁ d, I ₁ q and the stator winding voltage V ₁ d, V ₁ q, and 7 is the power in the dq coordinate system. It is a two-phase to three-phase converter that converts the voltage command value to be generated by the converter 1 into the actual three-phase instantaneous values _V1U , _V1V , _V1W . 8 is a d-axis current controller, which amplifies the difference between the d-axis component command value 1 ₁ d ^* of the stator winding current and its actual value I ₁ d to try to flow the current according to the command value. belongs to. Similarly, a q-axis current controller 9 controls the q-axis component of the stator winding current. 29 is a magnetic flux controller for controlling the d-axis component Φ ₂ d of the rotor magnetic flux linkage to a desired value Φ ₂ d ^* , 30 is a speed for controlling the rotor angular speed wr to a desired value wr ^* The controller. Reference numeral 31 is a divider, and 32 is a coefficient unit, and the slip frequency command ws ^* is calculated by the divider 31 and the coefficient unit 32.
Further, 22, 24, 26, 27, 28 are adders or subtractors.

Next, the operation will be described. First, the current control system will be described. The three-phase AC currents I _1U , I _1V , I _1W flowing into the stator windings of the induction motor 2 are detected by the current detector 4. Three
The phase-to-two-phase converter 5 has a frequency w _{1 of} a three-phase AC voltage V _1U , V _1V , V _1W applied to the stator winding with a three-phase AC current I _1U , I _1V , I _1W.
Two-axis Cartesian coordinate system that rotates in synchronism with, and converted into stator winding current I ₁ d, I ₁ q viewed from (dq coordinate system), I _1U , I
The conversion from _1V , I _1W to I ₁ d, I ₁ q is θ ₁ = ∫w ₁ dt (1) Done in. The d-axis current controller 8 amplifies the difference between the stator winding d-axis current command value I ₁ d ^* and the stator winding current I ₁ d obtained by the equations (1) and (2) to obtain the stator winding voltage. The d-axis voltage command value V ₁ d of is output. Similarly, for the q-axis component, q
The q-axis voltage command value V ₁ q is output by the axis current controller 9. The d-axis voltage command value V ₁ d and the q-axis voltage command value V ₁ q are 2
The actual three-phase instantaneous voltage V _1U , V _1V ,
Converted to V _1W . The conversion from V ₁ d, V ₁ q to V _1U , V _1V , V _1W is Done in. Three-phase instantaneous voltage V _1U , V obtained by this
_1V and _V1W are actually generated from the power converter 1 so that a desired current can flow. Next, the slip frequency calculation will be described. If the above current control system is operating at a sufficiently high speed, it can be regarded as 1 ₁ d ^* = I ₁ d, I ₁ q ^* = I ₁ q. At this time, the state equation of the system of the induction motor 2 when the stator winding currents I ₁ d and I ₁ q are regarded as inputs is as follows. ₂ d = −αΦ ₂ d + wsΦ ₂ q + βI ₁ d (4) ₂ q = −αΦ ₂ q−wsΦ ₂ d + βI ₁ q (5) r = γ (I ₁ qΦ ₂ d−I ₁ dΦ ₂ q)… ( 6) Here, α, β, γ are constants determined by the induction motor 2. Φ ₂ d is the d-axis component rotor winding linkage magnetic flux (hereinafter referred to as d-axis component magnetic flux), Φ ₂ d is the q-axis component rotor winding linkage magnetic flux (hereinafter referred to as q-axis component magnetic flux), wr is the rotor angular velocity, ws is the slip frequency, and ws = w ₁ −wr (7). Now Then, the equation (5) becomes ₂ q = −αΦ ₂ q (9). Since α> 0, the q-axis component magnetic flux Φ ₂ q approaches zero over time. Thus after a certain time Φ
Regarded as ₂ q = 0. The divider 31 and the coefficient unit 32 calculate the command value w _S ^* of the slip frequency ws based on the equation (8). The slip frequency command value w _S ^* and the rotor angular velocity wr are added by the adder 26, the AC voltage frequency w ₁ applied to the stator winding is calculated, and the two-phase to three-phase converter 7 and the power converter ₁ are calculated. Thus, the AC voltage having the frequency w ₁ is actually applied to the induction motor 2.

Next, the magnetic flux control will be described. When Φ ₂ d = 0 by the above-mentioned slip frequency control, controlling the magnetic flux means controlling the d-axis component magnetic flux Φ ₂ d. (4)
From the formula, from Φ ₂ q = 0, ₂ d = -αΦ ₂ d + βI ₁ d (10) and the d-axis stator winding current I ₁ d is manipulated to set the d-axis component magnetic flux Φ ₂ d to the desired value. You will have control. The magnetic flux controller 29 amplifies the difference between the d-axis component magnetic flux command value Φ ₂ d ^* and the d-axis component magnetic flux Φ ₂ d and outputs the stator winding current command value I ₁ d ^* . The value of the d-axis component magnetic flux Φ ₂ d is obtained by the magnetic flux calculator 6.

Next, speed control will be described. [Phi ₂ q = 0 by slip frequency control, if control is ^{_{Φ 2 d = Φ 2 d *}} ( constant) by the magnetic flux control (6) is q-axis becomes ^{_{r = γΦ 2 d * I 1}} q ... (11) By manipulating the stator winding current I ₁ q, the rotor angular velocity wr can be controlled to a desired value. Command value of the speed controller 30 in the rotor angular velocity wr ^* and outputs a command value I ₁ q ^* of measured values by amplifying the difference between wr q-axis stator winding current I ₁ q.

[Problems to be solved by the invention]

Since the speed / flux control device of the conventional induction motor is configured as described above, if the constant β of the induction motor 2 is unclear, the slip frequency control of the formula (8) becomes incomplete and Φ ₂ q ≠ 0. Then, the magnetic flux and speed control will also be Φ ₂ q
Since ≠ 0, the system becomes non-linear and it is difficult to obtain desired control characteristics. In addition, even if the constant β of the induction motor 2 is clear, Φ due to disturbance to the system, etc.
_{When 2} q ≠ 0, it is returned to Φ ₂ q = 0 depending on the constant α of the induction motor 2 as can be seen from the equation (9), and it returns only with the time constant determined by it. The system between Φ ₂ q ≠ 0 is a non-linear system, and there is a problem in that the characteristics and stability of speed / flux control are not guaranteed.

The present invention has been made in order to solve the above problems. Even if the induction motor constant is unclear or Φ ₂ q ≠ 0 due to disturbance or the like, the speed / speed can be stably maintained with desired characteristics. An object of the present invention is to provide a speed / flux control device for an induction motor capable of controlling magnetic flux.

[Means for solving problems]

The speed / flux control device for an induction motor according to the present invention is the difference between the rotor angular speed of the induction motor detected by the speed detector and the rotor angular speed command value, and the difference and the rotor linkage flux detected by the magnetic flux calculator. And the difference between this rotor interlinkage magnetic flux and the rotor interlinkage magnetic flux command value, and the product of this difference and the rotor angular velocity are used to control the magnitude of the current flowing through the induction motor. The electric current command value of the electric current controller and the operating frequency of the electric power converter that supplies electric power of a variable voltage variable frequency to the induction motor are obtained.

[Action]

The speed / flux control device for an induction motor according to the present invention treats the induction motor as a non-linear multivariable system by introducing a non-linear control theory, and controls Φ ₂ q = 0 by slip frequency control as in the conventional case. Instead of designing, q-axis component magnetic flux Φ ₂ q is also incorporated in the system as the number of states, and fixed so that rotor angular velocity wr, d-axis component magnetic flux Φ ₂ d, and q-axis component magnetic flux Φ ₂ q are controlled to desired values at the same time. The child winding currents I ₁ d, I ₁ q and the slip frequency ws are operated simultaneously.

〔Example〕

An embodiment of the present invention will be described below with reference to the drawings. FIG. 1 is a control block diagram showing an embodiment of the present invention. In FIG. 1, the same parts as those in FIG. 40 is the rotor angular velocity wr detected by the speed detector 3, the rotor angular velocity command value wr ^* , the rotor interlinkage flux Φ ₂ d, Φ ₂ q, the rotor interlinkage flux command detected by the magnetic flux calculator 6. The deviation I ₁ d ^* -I ₁ d, I ₁ between the current command value and the current feedback value based on the value Φ ₂ d ^* and the current I ₁ d, I ₁ q converted by the 3-phase to 2-phase converter 5. q ^* -I ₁ q and calculates the operation frequency w _1, said current deviation _{^{I 1 d * -I 1 d,}} I 1 q * -I 1 q and operating frequency w
It is a command value calculator that outputs ₁ to the current controllers 8 and 9 and the power converter ₁ , respectively. 10,11,12,13,14,15,16 are coefficient units, which are given by constants of the induction motor 2 and coefficients (K _{1 to} K ₆ ) determined by desired control performance and M. 17,18 is a multiplier, 19,
20 is a subtracter, and 21,23,25 are adders. In this figure, the outputs I ₁ d ^* , I ₁ q ^* , ws ^* of the speed / flux controller are given by the following equation.

I ₁ d ^* K (Φ ₂₁ −Φ ₂ d ^* ) −K ₂ Φ ₂ q (wr−wr ^* ) + MΦ ₂ d ^* … (12) I ₁ q ^* = K ₃ Φ ₂ q + K ₄ Φ ₂ d (wr ^{-wr *) ... (13) ws} * = K 5 (wr-wr *) + K 6 Φ 2 q ... (14) (12) (13) is fed back product is between the magnetic flux and speed as can be seen from equation It is a nonlinear control. here,
The meanings of equations (12) to (14) are as follows.

Equation (12) is obtained by subtracting the subtractors 19 and 20, the multiplier 17, the coefficient multipliers 10 and 11,
12, I ₁ d ^* is obtained by the adder 21, and (1
3) is the subtractor 20, multiplier 18, coefficient multipliers 13 and 14, adder 2
I ₁ q ^* is obtained by 3 and equation (14) is ws ^* obtained by the subtractor 20, the coefficient units 15 and 16, and the adder 25.

Then, the subtracter 22 outputs the result of subtracting the current I ₁ d from I ₁ d ^* obtained by the expression (12) to the d-axis current controller 8 as a current deviation.

Further, the subtracter 24 outputs the result of subtracting the current I ₁ q from I ₁ q ^* obtained by the equation (13) to the q-axis current controller 9 as a current deviation.

Further, the adder 25 outputs the result obtained by adding the rotor angular velocity wr to ws ^* obtained by the equation (14) to the power converter 1 as the operating frequency.

If the first term on the right-hand side of equation (13) is accurately described, K
₃ (Φ ₂ q −Φ ₂ q ^* ), but since Φ ₂ q ^* is zero in the steady state, it is simply described as K ₃ Φ ₂ q.

First, the principle will be described. Expressions (4), (5) and (6) are written as follows.

here u ₂ = I ₁ qu ₃ = ws According to Lyapunov's theory of stability, the derivative (time derivative) (x) of the Lyapunov function V (x) V (x) = x ^T Px defined by a certain symmetric positive definite matrix P is (x) <0 (x ≠ 0) (x) = 0 (x = 0), which is globally stable. When (x) is calculated when the system is expressed by equation (15), Becomes Here, the subscript T represents a transposed matrix. Now, as a symmetric positive definite matrix P To take. At this time, PB ₃ + ▲ B ^T ₃ ▼ = 0, and if u ₁ , u ₂ and u ₃ are defined from the equation (16) as follows, (x) can be made negative with x ≠ 0. .

^{_{u 1 = k 1 {x T}} (PB 1 + B 1 T P) x + 2C 1 T Px} k 1 <0
_{... (17) u 2 = k} 2 (x T (PB 2 + B 2 T P) x + 2C 2 T Px} k 2 <0
… (18) u ₃ ＝ k ₃ x ₂ ＋ k ₄ x ₃ … (19) When u ₁ , u ₂ is calculated, equations (17) and (18) are actually calculated, u ₁ ＝ k ₁ (2βx ₁ −2γpx ₂ x ₃ ) ... (20) u ₂ = k ₂ (2β x ₂ + 2γpΦ ₂ d ^* x ₃ + 2γp x ₁ x ₃ ) ... (2
1) Moreover, when (x) is actually calculated, As a result, −α−k ₃ Φ ₂ d ^* <0, k ₁ <0, k ₂ <0 If V is satisfied, V (x) <0 (x ≠ 0) can be satisfied. The number of parameters in the above equation is k ₁ , k ₂ , k ₃ , k ₄ ,
In addition to the condition of (x) <0 (x ≠ 0) with five p, it is possible to specify a desired degree of convergence.

Now, in equation (20), 2βk ₁ = K ₁ , 2γpk ₁ = K ₂
Then, it can be seen that equation (12) is obtained. Further, if 2βk ₂ = K ₃ and 2γpk ₂ = K _{4 in} equation (21), then equation (13) is obtained. Equations (19) and (14) may be set as K ₅ = k ₄ and K ₆ = k ₃ . Also, k ₁ and k ₂ are negative values from the equation (23), and α,
Since β, γ, p are motor constants with positive values, K ₁ ~
The value of K ₅ is always negative. The value of K ₆ is zero or Φ.
If the value has the same polarity as ₂ d ^* , equation (23) holds, so
It becomes zero or a value with the same polarity as Φ ₂ d ^* .

Note that the condition of equation (23) is that if there are errors in the motor constants α, β, and γ, In some cases, however, it is only necessary that the equation (22) becomes negative, so in practice If Here, δ depends on k ₁ , k ₂ , and α, and δ increases as the absolute value of these increases. Since k ₁ and k ₂ can be freely selected as control parameters, stability can be guaranteed by selecting the control parameters k ₁ and k ₂ according to the indefinite accuracy of the constant of the induction motor 2.

Also, the same effect can be obtained by performing the following feedback.

u ₁ = k ₁ (2βx ₁ -2γpx ₂ x ₃ ) + εx ₁ (25) u ₂ = k ₂ (2βx ₂ + 2γpΦ ₂ d ^* x ₃ + 2γpx ₁ x ₃ ) + εx ₂ (26) u ₃ = k ₃ x ₂ + k ₄ x ₃ (27) Here, in order to make the differential value of the Lyapunov function negative, βε−α <0, βε−α−k ₃ Φ ₂ d ^* <0 k ₁ <0, k ₂ < 0 And it is sufficient. Furthermore, by setting k ₁ = k ₂ k ₃ = 0 k ₄ = 2k ₁ βγp, it is possible to prove by simple calculation that the system can be decoupling linearly, and as a result, Φ ₂ d, Φ ₂ q,
Each control of wr can be designed as a single linear loop. In the above embodiment, the symmetric positive definite matrix p is However, other symmetric positive definite matrices can be used to construct a feedback rule that achieves the same effect.

〔The invention's effect〕

As described above, according to the present invention, the difference between the d-axis rotor interlinkage magnetic flux calculated by the magnetic flux calculator and the d-axis rotor interlinkage magnetic flux command value is multiplied by a negative coefficient. Multiplication means,
Second multiplication means for multiplying the difference between the rotor angular velocity detected by the velocity detector and the rotor angular velocity command value by the q-axis rotor interlinkage magnetic flux calculated by the magnetic flux calculator and a negative coefficient, Third multiplication means for multiplying the q-axis rotor interlinkage magnetic flux by a negative coefficient, the difference between the rotor angular velocity and the rotor angular velocity command value being added to the d-axis rotor interlinkage magnetic flux and the negative coefficient 4th multiplication means for multiplying by, and the multiplication result multiplied by the second multiplication means is subtracted from the multiplication result multiplied by the first multiplication means to generate the d-axis current command value of the current controller. The d-axis current command value generating means and the multiplication result multiplied by the third multiplying means add the multiplication result multiplied by the fourth multiplying means to generate the q-axis current command value of the current controller. Q-axis current command value generating means, the rotor angular velocity and the rotor angular velocity Fifth multiplication means for multiplying the difference from the command value by a negative coefficient, and multiplying the q-axis rotor interlinkage magnetic flux by zero or a coefficient having the same polarity as the d-axis rotor interlinkage magnetic flux command value. A command value calculation having a sixth multiplication means and an operation frequency generation means for generating the operation frequency of the power converter by adding the multiplication result of the sixth multiplication means to the multiplication result of the fifth multiplication means. Since the configuration is provided with a rotor, even if an error occurs in the rotor interlinkage magnetic flux which is the calculation result of the magnetic flux calculator due to the influence of disturbance or the uncertainties of the motor constants, the specified current value can be adjusted while compensating for the error. Since it is generated, there is an effect that a control system with high stability can always be configured.

[Brief description of drawings]

FIG. 1 is a block diagram showing a speed / flux control device for an induction motor according to an embodiment of the present invention, and FIG. 2 is a block diagram showing a speed / flux control device for a conventional induction motor. 1 is a power converter, 2 is an induction motor, 3 is a speed detector, 6
Is a magnetic flux calculator, 8 is a d-axis current controller, 9 is a q-axis current controller, and 40 is a command value calculator. In the drawings, the same reference numerals indicate the same parts.

Claims (1)

[Claims] 1. A power converter for supplying electric power of variable voltage and variable frequency to an induction motor, a current controller for controlling the magnitude of a three-phase alternating current flowing through the induction motor, and a rotor angular velocity of the induction motor. A speed detector for detecting the three-phase alternating current is converted into a d-axis current component and a q-axis current component in a two-axis rotating coordinate system that rotates in synchronization with the frequency of the alternating voltage applied to the stator winding. 3 phase to 2 phase converter and the above 3
A magnetic flux calculator for calculating d-axis and q-axis rotor interlinkage magnetic flux in the induction motor based on the d-axis current component and q-axis current component converted by the phase-to-two-phase converter and the stator winding voltage. A first multiplication means for multiplying a difference between the d-axis rotor interlinkage magnetic flux calculated by the magnetic flux calculator and the d-axis rotor interlinkage magnetic flux command value by a negative coefficient, and detected by the speed detector Second multiplying means for multiplying the difference between the rotor angular velocity and the rotor angular velocity command value by the q-axis rotor interlinkage magnetic flux calculated by the magnetic flux calculator and a negative coefficient, the q-axis rotation Third multiplication means for multiplying the interlinkage magnetic flux by a negative coefficient, and Fourth multiplication means for multiplying the difference between the rotor angular velocity and the rotor angular velocity command value by the rotor interlinkage magnetic flux of the d axis and the negative coefficient. Second multiplying means from the multiplication result multiplied by the first multiplying means The d-axis current command value generating means for subtracting the multiplied multiplication result to generate the d-axis current command value of the current controller, and the fourth multiplication result for the multiplication result multiplied by the third multiplication means. Q-axis current command value generation means for generating a q-axis current command value for the current controller by adding the multiplication results multiplied by the multiplication means, and a difference between the rotor angular speed and the rotor angular speed command value is negative. Fifth multiplying coefficient
Means for multiplying by zero or d with respect to the q-axis rotor flux linkage.
The sixth multiplication means for multiplying the rotor interlinkage magnetic flux command value of the shaft by a coefficient having the same polarity, and the multiplication result of the fifth multiplication means, and the multiplication result of the sixth multiplication means are added to add the result. A speed / flux control device for an induction motor, comprising: a command value calculator having an operating frequency generating means for generating an operating frequency of a power converter.