JPH085681A - Frequency conversion circuit - Google Patents

Abstract

PURPOSE:To remove the mixer communication signal from a local oscillator significantly from the output signal of a mixer by providing a variable phase shifter, a variable attenuator, a directional coupler, and a control section. CONSTITUTION:A variable phase shifter 10 outputs the signal from a local oscillator 50 while shifting the phase thus forming a phase angle reverse to that of a mixer communication signal 50a. The phase shifter 10, a variable attenuator 20 and a directional coupler 30 synthesize an inverted signal of the mixer communication signal 50a thus canceling the disturbance signal 50a selectively. A control section 40 receives a frequency setting data signal of sweep step frequency being delivered from a ramp voltage generating section 51 to the oscillator 50. A phase angle data corresponding to the frequency and an attenuation value data are then read out from a parameter memory 42. The data is delivered to corresponding phase shifter 10 and attenuator 20 in order to cancel the signal 50a. When the intermediate frequency f110 is close to the oscillation frequency f100 of the oscillator 50, the signal 50a is canceled thus eliminating the effect of the oscillator 50.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION The frequency signal of the local oscillator is also included in the output signal obtained by converting the input signal to be measured and the local oscillator signal into an intermediate frequency signal by a mixer. The present invention relates to a circuit for removing the frequency component of the included local oscillator.

[0002]

2. Description of the Related Art The prior art will be described with reference to FIGS. 5 and 6 by showing an example of a frequency converter of a spectrum analyzer. The configuration is a mixer 54, a local oscillator 50,
The lamp voltage generator 51 and the band pass filter (BP
F) 58 and a spectrum measuring section 70. The spectrum measuring section 70 includes an IF block 60,
The detector 72, the calculator 74, and the display device 78 are included.

In order to measure the frequency spectrum of the input signal f102, the local oscillator 50 sweeps a desired frequency range corresponding to the sweep range and supplies this output signal f100 to one of the mixers 54. The input signal f102 is
The signal is supplied to the other input of the mixer 54, mixed, converted to the intermediate frequency f110, output, and supplied to the BPF 58. The input frequency range is, for example, f102 = 10 KHz to 3.6.
GHz, and the frequency of the local oscillator 50 is f100 =
4 GHz to 7.6 GHz, intermediate frequency f110 =
It is 4.0 GHz. The BPF 58 has an intermediate frequency f11.
After removing unnecessary frequency components other than 0, the IF block 60
Supply to. The IF block 60 receives this signal, frequency-converts it to a lower intermediate frequency once or plural times, filters it with a desired bandwidth by the variable BPF 66, and then supplies it to the detection unit 72. The detector 72 detects this signal, quantizes it into a digital signal, and then supplies it to the calculator 74. In response to this, the calculation unit 74 converts the measurement data group for each sweep frequency into a display format and displays a spectrum on the display device 78.

Generally, in the output signal after being mixed by the mixer, the oscillation frequency signal f100 of the local oscillator is included.
Is also transmitted to the output side. This transfer level is about −30 to −40 dB even if it is good although it depends on the mixer circuit. Usually, since it is removed by the BPF of the intermediate frequency amplifying section thereafter, it does not cause a problem. The signal of which the oscillation frequency signal f100 is transmitted to the mixer output side will be hereinafter referred to as a mixer passage signal 50a.

By the way, in a usage mode in which the intermediate frequency f110 and the oscillation frequency f100 of the local oscillator 50 are close to each other, such as a spectrum analyzer, it cannot be removed by the BPF, which causes a problem. An example of such a state will be described. The slope waveform 32a shown in FIG. 6 is a spectrum waveform when no signal is input, and this is the local oscillator 5
It is the spectrum waveform of the mixer passing signal 50a from 0. The spectrum waveform 31 shows that the input signal f102 is 10
It is an example of a spectrum waveform in the case of 0 KHz or more. In the case of this example, there is no influence of the slope waveform 32a of the local oscillator 50, and the measurement is performed normally. On the other hand, the spectrum waveform 32b is an example of a waveform when the input signal f102 is 10 KHz. In this case, the waveform overlaps the sloped portion of the slope waveform 32a. With such an input signal, the BPF 58 and the variable BPF 66 may be insufficiently removed or the measurement level error may be large. Alternatively, in the case of a low level input signal, the mixer passing signal 5
There is a case where it is buried in 0a and cannot be measured. By the way, this problem can be mitigated if the variable BPF 66 is set to a narrow band condition and measurement is performed, but a limitation on setting the band condition is given to the user side, which imposes a limitation on the use of the measurement application. Not preferable.

[0006]

As described above, the oscillation frequency f100 and the intermediate frequency f11 of the local oscillator 50.
When 0 approaches, there is a problem that it becomes difficult in principle to remove it by the BPF of the intermediate frequency amplifier. For this reason, the lower limit frequency that can be used is greatly limited, and the measurement level error becomes large or measurement becomes impossible, which is a practical difficulty. Therefore, the problem to be solved by the present invention is to significantly eliminate the mixer passing signal 50a from the local oscillator in the output signal after being mixed by the mixer.

[0007]

In order to solve the above-mentioned problems, in the structure of the present invention, a variable phase shifter 10 for receiving the oscillation frequency signal f100 of the local oscillator 50, converting it to a desired output phase and outputting it is provided. A variable output level means 20 for receiving a signal from the variable phase shifter 10 and attenuating or amplifying the signal to a desired output level and outputting the attenuated or amplified signal is provided. , A combination means 30 for canceling the frequency component of the local oscillator 50 by combining both signals and outputting the combined signal. An example of the synthesizing unit 30 is a synthesizing unit using a directional coupler. Further, as an example of the variable output level means 20, a means using a variable attenuator, a means for operating a gain of 1 or less using a variable amplifier, or a variable amplifier and a fixed attenuator in series. There is a means to connect and realize.

In the above-mentioned solution means, FIG. 2 shows a solution means of the variable phase shifter 10 according to the present invention. That is, the oscillating frequency signal f100 of the local oscillator 50, the gain control signal from the control unit 40, and the variable amplifier 11 that amplifies the signal at a desired amplification degree are received. There is a means of configuring the variable phase shifter 10 with the distortion amplifier 12 that outputs a desired output phase using the nonlinear amplification region.

In addition to the above solving means, a signal for sweeping the oscillation frequency signal f100 of the local oscillator 50 is received, and corresponding parameter data is read from the parameter memory 42 for each oscillation frequency signal f100. The control unit 40 that supplies the variable phase shifter 10 and the variable attenuator 20 is provided, and for each oscillation frequency signal f100 of the local oscillator 50,
A parameter memory 42 for storing corresponding parameter data is provided, and there is an additional configuration means. In this case, even if the oscillation frequency f100 of the local oscillator 50 is swept, the mixer passing signal 50a can be canceled by following the sweep frequency.

Further, in addition to the control section 40 of the above solution means, a spectrum measuring section 70 for detecting the output level signal of the synthesizing section 30, the directional coupler 32, or the dielectric resonator having the coupling section 58c. A control means for receiving a level signal VLBL from 58b and supplying a control parameter to the variable phase shifter 10 and the variable output level means 20 while obtaining a minimum value of the level signal VLBL is added to the control section 40. There is a means. In this case, the mixer passing signal 5 including the current environmental temperature and circuit variation is included.
Offsetting 0a can be realized. For example, if the temperature sensor 46 is provided and the calibration is performed when the temperature changes by a predetermined value or more,
A stable offset can always be realized.

[0011]

The variable phase shifter 10 receives the phase control signal from the control section and can continuously change the phase, and has a function of forming an opposite phase signal for canceling the mixer passing signal 50a.
The variable phase shifter 10, the variable attenuator 20, and the directional coupler 30 have a function of synthesizing in a phase opposite to the mixer passing signal 50a and selectively canceling the disturbing mixer passing signal 50a. The control unit 40 receives the sweep signal from the ramp voltage generation unit 51, reads the corresponding data from the parameter memory 42, and supplies the desired data to the variable phase shifter 10 and the variable attenuator 20. Further, at the time of calibration, the parameter memory 4 receives the signal from the calculation unit 74 or the detection unit 48 and obtains the phase angle of the variable phase shifter 10 and the attenuation amount of the variable attenuator so that the level signal is minimized.
It has the role of storing and saving in 2. In the overall configuration, when the intermediate frequency f110 and the oscillation frequency f100 of the local oscillator 50 are close to each other, the mixer passing signal 50a is canceled to realize a good frequency conversion circuit without the influence of the local oscillator 50. There is.

[0012]

【Example】

(Embodiment 1) Embodiment 1 of the present invention is a configuration diagram in which a signal having an opposite phase is added to the mixer passing signal 50a after the mixer 54. The detection signal at the time of calibration is an example of the configuration that is received from the spectrum measurement unit and executed. This will be described with reference to FIGS. 1, 2A and 2B. The configuration is, as shown in FIG. 1, a variable phase shifter 10 and
Variable attenuator 20, directional coupler 30, and control unit 40
And the parameter memory 42 is added.

The variable phase shifter 10 converts the signal from the local oscillator 50 into a desired phase angle and outputs it. The variable phase shifter 10 is for forming a phase angle opposite to that of the mixer passing signal 50a. It suffices that the phase can be adjusted due to changes and variations in circuit elements, and for example, the phase angle of about 5 to 20 degrees is made variable. An example of this variable phase shifter 10 is shown in FIG.
It is configured as in the example shown in FIG. The configuration includes a variable amplifier 11 and a distortion amplifier 12.

The internal structure of the variable phase shifter 10 is composed of a variable amplifier 11 and a distortion amplifier 12, as shown in FIG. The variable amplifier 11 receives the gain control signal 11a from the control unit 40, amplifies the input signal with an arbitrary amplification degree, and then supplies it to the distortion amplifier 12, and is a general variable amplifier. The distortion amplifier 12 includes a capacitor C
91, C92, C93, C94, L97, L98,
R81, R82, matching circuits (MN) 13, MN15
, A FET 14 and a matching pad 16. The DC bias voltage Vg is supplied to the gate terminal of the FET 14 by R81, C92 and L97. C
Reference numeral 91 denotes an MN that allows the AC signal from the variable amplifier 11 to pass therethrough.
The input impedance is matched at 13 and supplied to the gate terminal of the FET 14. The MN 13 is a matching circuit made up of a distributed constant or lumped constant LC circuit. F
The output signal amplified by ET14 is output from the drain to MN15.
The output impedance is matched with and the AC signal is passed through with C93, and is supplied to the matching pad 16. The matching pad 16 is a 3 dB attenuator with a resistor for matching the impedance between the input and output,
The occurrence of misalignment due to the variation of the non-linear operation of T14 is reduced. The matching pad 16 may be deleted if desired. Further, the DC voltage Vd is supplied to the drain terminal of the FET 14 by R82, C94 and L98. The drain voltage Vd is supplied with a relatively low voltage, for example, 3 v, so that the drain voltage Vd is easily saturated.

The characteristic example shown in FIG. 2B is the distortion amplifier 1
2 shows changes in the output phase change 153 and the output level 152 when the second input level 151 is changed. In this characteristic diagram, there is a region 155 where the phase abruptly changes when the input level is given above a certain level. A region 155 in which this relatively large phase rotation is obtained shows the operating state of the FET 14 before and after the saturation region. In the present invention,
Using this characteristic, input level 1 before and after this saturation condition
By supplying 51 from the variable amplifier 11, the output phase angle is continuously varied. by the way,
A harmonic wave is generated by distortion amplification, but this harmonic wave is not a problem because it is easily filtered by the BPF 58 and the variable BPF 66 thereafter. This output signal is supplied to the variable attenuator 20 in the next stage.

The variable attenuator 20 receives the signal from the variable phase shifter 10 and receives the attenuation amount control signal 20 from the controller 40.
This is a general variable attenuator that receives a and is attenuated to an arbitrary amount of attenuation, and then is supplied to the directional coupler 30, and is continuously variable using a pin diode. Also,
Instead of the variable attenuator 20, a variable amplifier may be used to realize an amplification factor of 1 or less, or a variable amplifier and a fixed attenuator may be connected in series to achieve the same function. The above can be similarly realized as a means for realizing.

The directional coupler 30 synthesizes the mixer passing signal 50a from the mixer 54 and the signal of the opposite phase from the variable attenuator 20 to cancel only the mixer passing signal 50a and B
It is output to the PF 58. As a result, the amount of attenuation at the output point of the directional coupler 30 can be, for example, 30 dB or more by properly controlling the phase angle and the amplitude.

The control unit 40 receives the frequency setting data signal of the sweep step frequency given to the local oscillator 50 from the ramp voltage generating unit 51, and reads the phase angle data and the attenuation value data corresponding to the frequency from the parameter memory 42. However, by supplying this to the corresponding variable phase shifter 10 and the variable attenuator, it is controlled so as to cancel the mixer passing signal 50a. These control operations are operated in the frequency range in which the mixer passing signal 50a has an influence.
The data stored in the parameter memory 42 is stored in advance at the time of manufacturing adjustment for the sweep frequency range that requires the offset control. As another calibration means, there is a means for performing calibration at any time. That is, the calibration start input 44 from the outside is received in the environmental state at the present time and the operation is performed. In this calibration operation procedure, the input terminal f102 is set to a non-signal state,
The level signal is received from the calculation unit 74 of the measurement system, and the values supplied to the variable phase shifter 10 and the variable attenuator are controlled so as to minimize this level signal. That is, firstly, the phase angle of the variable phase shifter 10 is changed to obtain the phase angle that minimizes the level signal from the calculation unit 74, and secondly, the attenuation amount of the variable attenuator 20 is calculated. Is calculated to obtain the attenuation amount that minimizes the level signal from the calculation unit 74. This operation is performed about twice, the minimum level is obtained, and this minimum point is obtained as the calibration data. This calibration data is stored in the parameter memory 42 and stored for each step frequency. Further, a temperature sensor 46 may be provided as a calibration starting means from the outside and automatically executed when the temperature changes by a desired temperature or more.

(Embodiment 2) Embodiment 2 of the present invention is a configuration diagram in which a signal having an opposite phase is added to the mixer passing signal 50a after the mixer 54, and the detection signal at the time of calibration is an independent detection means. This is a case where it is provided and is a general-purpose application configuration example. This will be described with reference to FIG. The composition is
The variable phase shifter 10 and the variable attenuator 2 are different from the conventional configuration.
0, a directional coupler 30, a directional coupler 32, a detector 48, a controller 40, and a parameter memory 42 are added.

Variable phase shifter 10, variable attenuator 20, directional coupler 30, controller 40, parameter memory 4
2 is the same as the description of the first embodiment. Directional coupler 32
Is a coupler for detecting the signal after canceling the mixer passing signal 50 a by the directional coupler 30, detects a frequency signal from this, and supplies it to the detection unit 48. The detector 48 detects the signal from the directional coupler 32, quantizes it with an AD converter, and then supplies it to the controller. After that, this signal level is used at the time of calibration in the same manner as in the first embodiment.

(Embodiment 3) Embodiment 3 of the present invention is a configuration diagram in which a signal of an opposite phase is added to the mixer passing signal 50a after the mixer 54, and a case where a dielectric resonator is used for the BPF. 2 is an example of the configuration in the case where a coupling portion is provided in this dielectric resonance portion for detection during calibration. This will be described with reference to FIG. The configuration is different from the conventional configuration in that the variable phase shifter 10, the variable attenuator 20, the directional coupler 30, the detection unit 48, the control unit 40, the parameter memory 42, and the dielectric resonator 58b portion. To the joint 58
c is provided.

In the case of a bandpass filter using the dielectric resonator 58b as the BPF, the dielectric resonator 58b is used.
By forming the signal detecting coupling portion 58c adjacent to, the signal after the mixer passing signal 50a is canceled by the directional coupler 30 can be detected. This detection signal is supplied to the detection unit 48 and used at the time of calibration. The detection unit 48 is the second embodiment.
Is the same as

[0023]

Since the present invention is configured as described above, it has the following effects. By supplying the signal of the local oscillator 50 to the directional coupler 30 at a desired phase angle and amplitude level by the variable phase shifter 10, the variable attenuator 20, and the controller 40, the output signal after being mixed by the mixer 54 is output. From the inside, the effect of synthesizing with the disturbing mixer passing signal 50a and selectively canceling only the mixer passing signal 50a can be obtained. For example, an attenuation effect of 30 dB or more can be obtained. As a result, as shown in FIG. 6, it is possible to largely eliminate the influence of the slope waveform 32a of the conventional local oscillator 50 on the slope waveform 33a. As a result, the spectrum waveform 32b of the usable input signal is improved to the original spectrum waveform 33b. Accordingly, the oscillation frequency f100 of the local oscillator 50 is
Signals in the vicinity can be measured with higher accuracy, or the lower limit frequency that can be used expands. This has the effect of realizing a good frequency conversion circuit.

[0024]

[Brief description of drawings]

FIG. 1 is a circuit configuration diagram of frequency conversion for adding a signal of an opposite phase to a mixer passing signal 50a after a mixer 54 according to the first embodiment of the present invention.

FIG. 2A is a circuit configuration diagram of a variable phase shifter 10 of the present invention. (B) A characteristic diagram showing changes in the output phase change 153 and the output level 152 when the input level 151 of the distortion amplifier 12 is changed.

FIG. 3 is a circuit configuration diagram of frequency conversion for adding a signal of an opposite phase to the mixer passage signal 50a after the mixer 54 according to the second embodiment of the present invention, and the detection signal at the time of calibration is an independent detection unit. This is an example of the case where it is provided.

FIG. 4 is a circuit configuration diagram of frequency conversion in which a signal of an opposite phase is added to the mixer passage signal 50a after the mixer 54 according to the third embodiment of the present invention, and a detection signal at the time of calibration is a dielectric resonance in the BPF. This is an example of a case where a detector is used and a coupling portion for detection is provided in this portion.

FIG. 5 is a circuit configuration example in the case of a conventional frequency converter of a spectrum analyzer.

6 is a waveform diagram example of frequency spectrum characteristics when the intermediate frequency f110 and the frequency f100 of the local oscillator 50 are close to each other. FIG.

[Explanation of symbols]

 10 variable phaser 11 variable amplifier 11a gain control signal 12 distortion amplifier 13, 15 MN (matching circuit) 14 FET 16 matching pad 20 variable attenuator (variable output level means) 20a attenuation amount control signal 30 directional coupler (combining means) 32) Directional coupler 31, 32b, 33b Spectral waveform 32a, 33a Slope waveform 40 Control unit 42 Parameter memory 44 Calibration start input 46 Temperature sensor 48 Detection unit 50 Local oscillator 50a Mixer passing signal 51 Lamp voltage generation unit 54 Mixer 58 Bandpass filter (BPF) 58b Dielectric resonator 60 IF block 66 Variable BPF 70 Spectrum measurement unit 72 Detection unit 74 Calculation unit 78 Display device 81, 82 R (resistance) 91, 92, 93, 94 C (capacitor) 97, 98 L ( Yl)

Claims (4)

[Claims] 1. A frequency conversion circuit for receiving an input frequency signal (f102) and an oscillating frequency signal (f100) of a local oscillator (50) and mixing and outputting the mixed signal in a mixer (54). A variable phase shifter (1 that receives a signal (f100), changes it to a desired output phase, and outputs it.
0) is provided, and variable output level means (20) for receiving the signal from the variable phase shifter (10), attenuating or amplifying to a desired output level and outputting the attenuated output level is provided. Signal and mixer (5
4) The output signal of 4) is received, and a synthesizing unit (30) for synthesizing both signals and canceling and outputting the frequency component of the local oscillator (50) is provided. Frequency conversion circuit. 2. The variable phase shifter (10) according to claim 1, further comprising an oscillation frequency signal (f100) of a local oscillator (50),
A variable amplifier (11) that receives a gain control signal from a control unit (40) and amplifies it with a desired amplification degree, and a signal from the variable amplifier (11), and a nonlinear amplification region of a distortion amplifier (12). Variable phase shifter (10) with a distortion amplifier (12) that outputs a desired output phase using
And a frequency conversion circuit comprising the above. 3. In addition to the constituent means according to claim 1, a signal for sweeping an oscillation frequency signal (f100) of the local oscillator (50) is received, and a frequency component of the local oscillator (50) is received for each sweep frequency. The data for canceling the variable is read from the corresponding parameter memory (42), and the variable phase shifter (1
0) and a controller (40) for supplying the variable output level means (20), and a parameter memory (42) for storing the corresponding data for each sweep frequency of the local oscillator (50). Frequency conversion circuit characterized by doing. 4. A spectrum measuring section (70) for detecting an output level signal of the synthesizing section (30) or a directional coupler (3) in addition to the constituent means according to claim 3.
2) or receives a level signal (VLBL) from a dielectric resonator (58b) having a coupling section (58c) and supplies a control parameter to the variable phase shifter (10) and the variable output level means (20). However, this level signal (VLBL)
A frequency conversion circuit characterized in that the control unit (40) is provided with control means for obtaining a value that minimizes the above.