JPH0965679A - Brushless motor controller - Google Patents

Abstract

PROBLEM TO BE SOLVED: To provide a brushless motor controller which assures highspeed operation with higher accuracy, although there is a problem that phase of a coil current of a brushless motor is reduced by the time constant of a coil and operation delay of a control microcomputer, etc. SOLUTION: A magnetic pole detector is mounted to a brushless motor to detect the rising and falling of the magnetic pole detecting signals Cu, Cv, Cw with a signal processor and output a signal Cp. Gravity of waveform of a current command value may be changed by changing peak value of the current command value of each Cp and it becomes equivalent to shift of the current command value by shifting the gravity of the current command value. Moreover, with these detected signal, amplitude I* and lead angle β of the current command value, a current command may be generated by a motor current command arithmethic unit 14.

Description

Detailed Description of the Invention

[0001]

BACKGROUND OF THE INVENTION 1. Field of the Invention The present invention relates to a controller for a brushless motor, and more particularly to a controller capable of controlling the phase of current in a stator winding.

[0002]

2. Description of the Related Art In recent years, in association with the rapid growth of FAs, OAs, and information devices, motors as drive sources thereof and control methods thereof have made remarkable progress. Among them, brushless motors are widely used in various fields.

On the other hand, as a kind of brushless motor, there is a control method in which the rotation angle of the rotor of a stepping motor is detected by a rotary encoder and the excitation is switched according to the displacement of the rotor. This control method is being widely used because it can surely avoid stepping out of the stepping motor and missteps. FIG. 4 shows a conceptual configuration of a three-phase brushless motor, where 41 is a stator, u-u'v-v ',
The three windings of ww 'are arranged at a position of an electrical angle of about 120 degrees, and the rotor 42 including the stator 41 and the permanent magnet magnetized to the two poles through the air gap is rotatable. It is supported, and θ is the rotation angle of the rotor. In order to consider the output torque of the three-phase brushless motor shown in FIG. 4, it is common to study the permeance P of the air gap between the stator 41 and the rotor 42 and the current flowing through the windings of each phase of the stator 41. Is. The permeance P (θ) of the air gap between the stator 41 and the rotor 42 of the three-phase brushless motor can be approximated by the equation (1).
Here, P0 and k1 are constants determined by the structure.

P (θ) = P0k1sinθ (1)

When the currents flowing through the windings u, v, w of the stator 41 are iu, iv, iw and the amplitude of the current is I, then iu = Isinωt (2) iv = Isin ( ωt-2π / 3)
(3) iw = Isin (ωt-4π / 3)
It can be expressed as (4).

Therefore, the output torque T of the motor is T = K [P (θ) iu + P (θ-2π / 3) iv + P (θ
-4π / 3) iw] is expressed by (5), the equations (1) to (4) are substituted into the equation (5),

[0007]

[Equation 1] (6) K is a constant.

In order to operate the motor with high efficiency, the rotation angle θ of the rotor 42 of the brushless motor is detected and
Each current value must be controlled according to equations (2) to (4). Therefore, if ωt = θ,

[Equation 2] (7) However, the current flowing through the stator 41 of the brushless motor causes a phase delay due to the time constant of the winding and a delay in calculation by the microcomputer. If there is a phase delay φ in the actual current detected from each winding, ωt = If θ−φ, the currents iu, iv, iw are

Iu = Isin (θ−φ) (8) iv = Isin (θ−2π / 3−φ) (9) iw = Isin (θ−4π / 3−φ) ) (10) Therefore, the output torque T is

[0010]

(Equation 3) (11) As a result, the torque decreases and it is necessary to correct the current phase delay. In order to compensate the above-mentioned drawbacks of the brushless motor, there is also a method of mounting an encoder on the outside and correcting the delay of the current phase according to the rotation angle of the rotor of the motor.

This method will be described with reference to FIG. In the figure, 91 is the waveform of the current command value controlled by the encoder, where N is the number of encoder pulses between one pitch angle of the rotor magnetic pole and 2π radians, the sine wave changes in steps every 2π / N radians. The wave 92 can be approximated.

Therefore, when the phase of the current command value is moved by the device for changing the phase of the control signal by the encoder, the waveform of the current command value becomes as shown in FIG. 10 and 2π / N.
The current phase can be shifted every time, but the number of pulses N of the encoder needs to be increased in order to increase the accuracy of the phase shift, and the high pulse encoder has a drawback that it is expensive.

[0013]

DISCLOSURE OF THE INVENTION Problems to be Solved by the Invention As described above, the conventional brushless motor control device has the drawbacks that the output torque is reduced due to the influence of the delay of the current phase, the high speed rotation is impossible, and the efficiency is deteriorated. Further, the brushless motor with the high pulse encoder attached has a drawback that the encoder is expensive.

[0014]

In order to solve the above problems, in a brushless motor control apparatus according to the present invention, a magnetic pole detector is attached to the brushless motor, and a command value of the winding current is determined by a magnetic pole detection signal. This is solved by a method of controlling the phase of the winding current by calculating and changing the peak value of the command value of the winding current for each magnetic pole detection signal.

[0015]

The controller of the brushless motor according to the present invention comprises:
By the above means, the phase of the winding current is phase-shifted, the phase delay of the winding current is corrected, and the maximum output torque is obtained, whereby high-speed operation can be performed and efficiency can be increased.

[0016]

Embodiments of the present invention will be described below with reference to the drawings. FIG. 1 is a circuit diagram showing a configuration of an embodiment of a brushless motor control device according to the present invention. In FIG. 1, 11 is a three-phase brushless motor, 12 is a magnetic pole detector attached to the rotor of the three-phase brushless motor 11,
Reference numeral 13 is a signal processor of the magnetic pole detector, 14 is a current command calculator, 15 is a current control device, and 16 is a drive circuit. Magnetic pole detection signals Cu, Cv, detected by the magnetic pole detector 12
Cw is sent to the magnetic pole signal processor 13 to generate a signal Cp for each π / 3 radian, and sent to the current command calculator 14 together with the magnetic pole detection signals Cu, Cv, Cw, and the amplitudes of the magnetic pole detection signal and the current command value. A three-phase AC current command value i * u, i * v, i * w is generated by the current command calculator 14 from I * and the advance angle command β of the current command. The amplitude I * of the current command value and the advance angle command β of the current command are input to the current command calculator 14 from the host controller.

FIG. 3 shows the waveforms of the magnetic pole detection signals Cu, Cv, Cw in the magnetic pole signal processor 13, 31 is a magnetic pole detection signal, and these signals are three-phase rectangles with a phase difference of 2π / 3 radians. It is a wave signal. The waveform of the signal Cp generated every .pi. / 3 radian by detecting the rising and falling edges of all phases is 32.

The current command calculator 14 has a magnetic pole detection signal Cu, Cv, Cw of a rectangular wave having a phase difference of 2π / 3 radians, which is the basis of the three-phase current command value, in the interval of 2π radians. Cp generated by the signal processor 13 changes the current amplitude I * to a predetermined value for each π / 3 radian to generate the three-phase current commands iu, iv, iw. The value of the angle β allows the amplitude of the current to be changed.

FIG. 2 shows a signal output from the current command calculator 14, which is one of the three-phase current commands (advance angle command β = 0) generated based on the magnetic pole detection signals Cu, Cc, Cw and Cp. (For example, U phase) 21, the current amplitude is h1I * between 0 and π / 3 radians, which is 2π from π / 3 radians.
It is h2I * for / 3 radians, h3I * for 2π / 3 radians and π radians, and 2π for π radians.
During the radian, the amplitude from 0 to π radian is repeated with the sign reversed. If the harmonics of the current command value are ignored, the current command value is almost sinusoidal Sin θ as shown by 22.
Is close to.

In the waveform 21 shown in FIG. 2, the peak value of the current command value for each π / 3 radian is h1I * (0≤θ <π /
3), h2I * (π / 3 ≦ θ <2π / 3), h3I * (2π
/ 3 ≦ θ <π), h4I * (π ≦ θ <4π / 3), h5I *
(4π / 3 ≦ θ <5π / 3), h6I * (5π / ≦ θ / 3
<2π), an example of the coefficient of the peak value of the current command value when the lead angle command β = 0

It is assumed that h1 = 0.5 h2 = 1.0 h3 = 0.5 h4 = -h1 h5 = -h2 h6 = -h3.

The center of gravity g of the waveform of the current command shown in FIG. 2 between θ = 0 and π can be calculated by the following equation.

[0023]

(Equation 4) (12) Here, h (θ) is a function representing the height of the waveform. h (θ)
Is constant every π / 3, so h1I *, h2I *, h above
Substituting 3I * into equation (12), the center of gravity g of the waveform is

[0024]

(Equation 5) It can be expressed by (13). Here, when the peak value of the current command value when the lead angle command β = 0 is substituted into the equation (13), the center of gravity g of the waveform at this time is at a position of π / 2.

According to the equation (13), the waveform gravity center g of the current command value can be moved by changing the peak value of the current command value for each π / 3 radian. By moving the center of gravity g of the current command value, the waveform of the current command value becomes equivalent to the case of moving by the advance angle β. For example, the waveform of the current command value is π /
If you want to proceed by 12 (β = π / 12) radians, h1 =
Substituting 0.75, h2 = 1, h3 = 0.25 into equation (12), the waveform centroid g of the current command value is 5 as shown in FIG.
The position is π / 12. That is, the waveform centroid g of the current command value has moved from π / 2 to the left of π / 12 radians. The waveform of the current command value in this case is 51 in FIG.
The current command value ignoring harmonics is a substantially sine wave I such as 52.
* sin (θ + β)

From equation (13), the peak value h of the current command
Several examples will be shown below as examples for changing the center of gravity g of the waveform from 1I *, h2I *, and h3I *.

In order to set the advance angle β between 0 and π / 6, h1 = 0.5 + Δh h2 = 1.0 h3 = 0.5-Δh h4 = -h1 h5 = -h2 h6 = -h3. However, Δh = 3β / π.

In order to set the advance angle β between π / 6 and π / 3, h1 = 1.0 h2 = 1.0-Δh h3 = -Δh h4 = -h1 h5 = -h2 h6 = -h3. However, Δh = 3 (β−π / 6) / π.

The waveform of the current command value and the substantially sine wave when the lead angles β = π / 6, π / 4, π / 3 are 6 in FIG. 6, respectively.
1, 62, 71 and 72 in FIG. 7, and 81 and 82 in FIG.

Although the above embodiment has been described with respect to the case where the three-phase brushless motor is used, the present invention is not limited to this and can be applied to those having other phases.

[0031]

The brushless motor control apparatus according to the present invention has the above-described structure, is equipped with a magnetic pole detector, and calculates the winding current command value from the rotor magnetic pole detection signal to detect the magnetic pole. The phase of the winding current can be controlled by changing the peak value of the winding current command value for each signal, the phase delay of the winding current can be corrected, and high-speed operation can be performed. It has the effect of being able to drive effectively.

[Brief description of drawings]

FIG. 1 is a circuit diagram showing an embodiment of a brushless motor control device according to the present invention.

FIG. 2 is a current command waveform (phase shift β = 0) diagram of an embodiment of a controller for a brushless motor according to the present invention.

FIG. 3 is a waveform diagram of magnetic pole detection signals of an embodiment of the controller of the brushless motor according to the present invention.

FIG. 4 is a schematic diagram showing the structure of a brushless motor embodying the present invention.

FIG. 5 is a current command waveform (phase shift β = π / 12) diagram of an embodiment of the brushless motor control device according to the present invention.

FIG. 6 is a current command waveform (phase shift β = π / 6) diagram of an embodiment of a brushless motor control device according to the present invention.

FIG. 7 is a current command waveform (phase shift β = π / 4) diagram of an embodiment of the controller of the brushless motor according to the present invention.

FIG. 8 is a current command waveform (phase shift β = π / 3) diagram of an embodiment of a brushless motor control device according to the present invention.

FIG. 9 is a current command waveform diagram (phase shift β = 0) by a conventional high pulse encoder.

FIG. 10 is a current command waveform diagram (phase shift β = 2π / N) by a conventional high pulse encoder. [Brief Description of Drawings] 11: Three-phase brushless motor 12: Magnetic pole detector 13: Magnetic pole signal processor 14: Current command calculator 15: Current control device 16: Drive circuit 21: Current command value waveform (phase shift β = 0) 22: Almost sine wave of current command value (phase shift β = 0) 31: Magnetic pole detection signal 32: Signal every π / 3 41: Stator of brushless motor 42: Rotor of brushless motor 51: Current command value Waveform (phase shift β = π / 12) 52: sine wave of current command value (phase shift β = π / 12) 61: current command value waveform (phase shift β = π / 6) 62: abbreviation of current command value Sine wave (phase shift β = π / 6) 71: Current command value waveform (phase shift β = π / 4) 72: Approximate sine wave of current command value (phase shift β = π / 4) 81: Current command value waveform (Phase shift β = π / 3) 82: Approximate sine wave of current command value (Phase shift β = π / 3) 91: For high pulse encoder Current command value waveform (phase shift β = 0) 92: Almost sine wave of current command value (phase shift β = 0) 101: Current command value waveform by high pulse encoder (phase shift β = 2π / N) 102: Current Approximate sine wave of command value (phase shift β = 2π / N)

Claims (1)

[Claims] 1. A stator provided with a three-phase stator winding, a rotor provided with a permanent magnet having a plurality of magnetic poles facing the stator with a gap, and the rotor being rotatable. A bearing device for supporting, means for detecting the magnetic pole position of the rotor,
A controller for a brushless motor, comprising: means for converting the change in the magnetic pole position into an electric signal; and energization control means for sequentially energizing the stator windings of the three phases by the electric signal converted by the means, Means for detecting rising and falling edges of the magnetic pole detection signal and calculating a current command value to be passed through the three-phase winding for each edge signal;
By changing the magnitude of the current command value for each edge signal to move the waveform center of gravity of the current command value, a control means for shifting the phase of the current command value is provided. Control device.