JPS5963977A - Rectifier - Google Patents

Abstract

PURPOSE:To reduce the harmonic component of a polyphase AC input current and to obtain a constant DC output having small ripple by controlling the conducting time of switching elements by a specific method so that the polyphase AC input current flows in the sinusoidal waveform. CONSTITUTION:A 3-phase AC input is applied through inductors Lu-Lw, capacitors Cu-Cw and rectifiers Rec1-Rec3 to converters G1-G3, and the outputs are applied through rectifying and smoothing circuits to a load F. Switching transistors Q1, Q'1 in the converters G1-G3 are respectively switched in high conversion frequency by the frequency of 3-phase AC. The energy produced at the output side at every period of the conversion frequency is controlled to be proportional to the product of the square of the 3-phase AC and the control amount, and proportional in the total energy produced from the respective phases of the control amount.

Description

[Detailed description of the invention] There are many failures and illuminations, and the period of the phase open current input is also a value proportional to the square of the instantaneous voltage value of the multiphase current transformation every sufficiently short period.
? It concerns an AC-DC conversion that obtains at the output an energy that depends on a constant D1 relationship with Al m.

A conventional typical switching mode rectifier for obtaining a desired value of DC voltage from a multi-phase AC input, for example a commercial six-phase AC input, is shown in FIG. This conventional rectifier has an inductor connected in series to each 6-phase AC input terminal, and 6 rectifiers D1 to D6 to '&J), which performs full-wave rectification of the 3-phase AC input. Circuit ReC1, a smoothing circuit t for smoothing the output of this six-phase full-wave rectifier circuit, a smoothing inductor Ll, a capacitor C1, and a primary winding 1lli of the transformer T.
Switching transistors Ql s Qx are connected to each other in an if column via lNl, and diodes D7 and D conduct between the off-states of these switching transistors to allow energy in the transformer to flow into the capacitor C1.
B, a rectifier D9 that rectifies the voltage of the secondary winding N2 of the transformer T, and a flywheel diode D. , an inductor L2 and a capacitor C2, forming an output filter circuit (d),
Switching transistors Q1 and Q are used to keep the DC output voltage between output terminal 010' and output terminal 0.0' constant.
It consists of a control circuit Con that controls the pulse width of i.

In such a conventional rectifier, the 6-phase AC input sound is monitored by a 3-phase full-wave rectifier circuit (1eci) (after IL, the DC current is smoothed by a smoothing circuit consisting of an inductor L1 and a capacitor C1, so the highest voltage Since the phase T pulse current flows through the phase diode, it contains many harmonic components.Even with the inductor Lu%Lv1Lw provided in the rectifier circuit 1tea1, at least the low-order high W3
Wave components cannot be removed. These harmonic components not only cause single-conductivity interference to the communication circuit and cause cwt-m operation, but also have an adverse effect on surrounding equipment connected to the same feed line, such as 'fc-c' and multi-phase. This results in an increase in power loss in the variable current feeder line, and in combination with a generator, an increase in capacity due to an increase in power loss in the generator. Another drawback is that a smoothing inductor L1 and a capacitor C1 are required.

The present invention eliminates these conventional drawbacks,
Symmetrical polyphase current transformation and 2-phase 3i type input side and its constant load? Based on the knowledge that the instantaneous value of the energy flow is constant on the DC output side with I-, the conduction period of each switching element is controlled in a specific manner so that the multiphase AC input current flows in a sinusoidal waveform. In particular, it is characterized in that it is possible to completely reduce the high wave component of the multiphase AC input current, and to obtain a constant DC output with a small ripple component.

In a symmetrical n-phase AC device (n≧6) with a symmetrical load, the instantaneous voltage V on the input side is:
・Represented by (1). However, j=1.2.6,...
・-n.

7m is the effective value of the input voltage vj.

Here, if the loads connected between each phase are resistive loads each having a resistance value R, the instantaneous power Pi supplied to the input section
is, P・=n−vm2/R・・・・・・・・・・・・・・・
・・・・・・・・・・・・・・・・・・(ri) becomes.

This equation 6) shows that the energy flowing in the symmetric n-phase AC device is one foot. Therefore, if a constant energy t- is obtained on the DC output side, a sinusoidal polyphase AC input current flows on the input side, so that the high t1 wave component can be sufficiently reduced.

Also, in order to extract a certain amount of energy to the DC output side, it is clear from the above equation that it is sufficient to extract energy proportional to the square of the instantaneous voltage value of each phase at each point in time of the +3C current input. Ru.

The present invention provides a specific AC-D based on the above-mentioned findings.
It provides C-variant mk.

DESCRIPTION OF THE PREFERRED EMBODIMENTS Each embodiment of ADC conversion according to the present invention will be described in detail below with reference to the drawings.

First, one embodiment of the present invention will be fully explained with reference to FIGS. 2 to 5.

The main circuit of the rectifier shown in Fig. 2 is for each phase U of the three-phase AC input.
, V, and W lines are inductors connected in series.
A rectifier circuit 1ec1, Ree2, Iζec) with the same configuration as the capacitor C and the capacitor C1 connected between the Lv s Lw s phases, and a pair of switching transistors Q that perform switching operations at the same time.
1, a transformer T having a primary winding Nl and a secondary winding N2, and a switching transistor Q1.
and Ql are conductive when Ql is off to release the excitation energy stored in the transformer T via the capacitor C4.
1, G2, G3, rectifiers D9, D9, Dg connected to combine the outputs of each conversion section in parallel, 7 life-wheeling diodes, inductors L2, L2 and capacitors c2, c2 that constitute a 010% smoothing circuit. ′
A load F is connected to the output terminal of the capacitor C3.

The rectifier in this embodiment does not require a smoothing circuit for the same input wavenumber between the rectifier circuit that rectifies each phase voltage and the conversion section. Using a completely new control method, each switching transistor Q1 and Q1' is switched to open and close a rectified sinusoidal voltage, thereby significantly reducing the harmonic components of the input current and reducing the ripple component. This provides extremely low and stabilized DC output.

Fig. 6 shows an example of the block configuration of a control circuit that performs this control method, Fig. 4 shows signals indicating the operation timing of each part, and Fig. FIG. 3 is a diagram for explaining the output waveform of a person. In Fig. 23, the reference (g oscillator 1 generates a reference pulse signal with a frequency sufficiently higher than the frequency of the polyphase AC input, for example, 201G (z). This reference pulse signal is generated at time t1 in Fig. 4. This is indicated by signal 5. At the rising edge of No. 4T a, the reset pulse forming circuit 2 generates a reset pulse with a predetermined pulse width, for example, 1 μsec. This reset pulse is passed through the OR circuit 5 to the reset FET.
The voltage is applied to the gate of FET 6, and this FET 6 k is turned on by the pulse width, thereby discharging the charge in the capacitor 7 to j·1 zero. The delay circuit 3 is a latch circuit that drives an ON signal at a time t2 when the fall of the reset pulse is delayed by a time approximately equal to the carrier accumulation time of the switching transistors Q1 and Q1', that is, at a time t2 delayed by a time τ from the signal a. Give to 4. Due to this, the drive latch circuit 4 is connected to the switching transistor Q of the first conversion section G1.
1. Give the drive signal 5lt- to the base of Q, and then apply the signal d.
Turn on as shown.

A capacitor 7 connected in parallel with FET6K is charged by a constant current from a controllable constant current rA9 whose constant current value is controlled by an error No. 46 from an error amplifier 8. The error signal output by the error amplifier B8 depends on the difference between the detection signal Sd, which is proportional to the pressure of the DC output in the rectifier, and a reference value. Therefore, the charge voltage of capacitor 7 is the reference signal a
K 14 = After the reset pulse is applied and the voltage drops to almost zero, the control variable, in this example, the output voltage detection 4
．． -5@Sd It increases in proportion to the size of the difference between it and the reference value. The charge voltage of the capacitor 7 is determined by each comparator 10.10.
', 10''. The negative terminals of these comparators are connected to the DC side terminals of full-wave rectifiers 11, 11', 11'' and resistors 12, 12', 12'', respectively. A voltage proportional to the U-V interphase voltage of the three-phase AC input is applied between the AC side terminals 13 and 14 of the full-wave rectifier 11, and a voltage V - W is applied between the AC side terminals 16' and 14' of the full-wave rectifier 11'. A voltage proportional to the mutual voltage is also applied to the full-wave rectifier 11'' (D alternating current (j
Since a voltage proportional to the W-U phase voltage is applied between the Q terminals 16'' and 14'', the negative terminal of the comparator 10 receives a full-wave rectified sine wave of the U-W phase alternating current voltage. Similarly, at the negative terminals of comparators 1, 0' and 10'', a rectified sinusoidal voltage appears on the AC voltage between the ■-W phase and the AC voltage between the W-U phase. Voltage is applied. The comparators 10゜10' and 10'' compare the voltages as mentioned above applied to the positive and negative terminals, and when the voltage on the positive terminal becomes equal to the voltage on the negative terminal, Output off No. 16. The off signal C of the comparator 10 is input to the drive latch circuit 4 at time t3, and accordingly, the drive latch circuit 4 stops supplying the pace drive signal S10 #J to the switching transistor Q1%Q1 in the conversion section G1.

Therefore, the transistors Q1 and Q1' turn off the storage time 7 at a later time t4 as shown by the signal d in FIG. 4.Then, the output signal C from the comparator 10 is It is input to the reset pulse forming circuit 2'.

Accordingly, the circuit 2' generates a reset pulse k similar to the reset pulse generated by the circuit 2.
is applied to the gate of the capacitor 7 and turned on to discharge the charge of the capacitor 7. In addition, a delay circuit 6 having the same configuration as circuit 6
' is applied to the drive latch circuit 4' at time t5, which is delayed by a time τ approximately equal to the carrier accumulation time of the signal C in response to the reset pulse. Accordingly, the circuit 4' applies a drive signal 82' (l-) to the switching transistor of the converter G2 to turn on the lever (15
No. g). (+) and the capacitor 7 are slightly charged by the current depending on the magnitude of the output voltage detection signal Sd during the on-operation period of the converter G2. Is this charging pressure comparable? ,i
10', the voltage proportional to the v-W interphase voltage is compared with the full-wave rectified sinusoidal voltage as described above, and at the time t6 when both voltages become quiet, the signal is applied to the ft drive latch circuit 4'. Together with the reset pulse forming circuit 2''
give to Immediately upon receiving the signal ft, the drive latch circuit 4' stops sending out the base drive signal S2, and accordingly, the converter G2 starts at time 1. to turn off.

Next, the reset pulse forming circuit 2'' receives the signal f from the comparison surface section 10' and outputs the reset pulse to F via the OR circuit 5.
It is applied to the gate of ET6, turning it on and instantly discharging the charge in capacitor 7.

The delay circuit 6'' of the circuit 6 receives the reset pulse from the circuit 2'' and applies it to the No. 2 ht-drive latte circuit 4''e at time t8 delayed by the time τ from the signal f.

Accordingly, the circuit 4'' applies the output signal S3 to the switching transistor of the converter G3 to turn it on (signal j).The capacitor 7 is discharged and discharged by turning on the FET 6, and the output voltage detection signal Sd is again turned on. The charging voltage of the capacitor 7 is compared with the sinusoidal voltage obtained by full-wave rectification of the voltage proportional to the W-U phase sending pressure of the white phase or current input by the comparator 10''. The comparator 10'' is turned off at the ninth point t9 when both voltages become equal (K11 is applied to the drive latch circuit 4''. At this point, the circuit 4'' stops supplying the drive signal S3, and the converter G
3 turns off after the storage time of its switching transistor has elapsed (signal j). In the section where both the rectifiers 9' and 9'' are non-conducting, the energy stored in the inductor L2 and LiK flows through the diode D1o and the load Ft-.In this way, each part operates for one period T. is completed.In addition, in Fig. 6, 15 indicates each full-wave rectifier 11.11.
This is a compensating diode to cancel the negative effect of forward drop of 11''. The switching operation is performed within a cycle that is sufficiently smaller than one cycle, and as the switching transistor that is on is turned off, the next switching transistor is turned on, and the switching transistor in each conversion section is is controlled by a pulse width proportional to the output voltage detection signal and a sinusoidal voltage proportional to each phase ill voltage, and each switching element opens and closes the rectified sinusoidal voltage of each phase voltage. The switching transistor of each conversion unit has a value in which the energy taken out from each phase of the polyphase AC to the output side at a sufficiently short cycle than the cycle of the polyphase AC is proportional to the square of the instantaneous voltage value of the polyphase AC. It can be seen that the switching is controlled in proportion to the product of (Kain" θt: is a constant) and the control amount.

Furthermore, in order to make this control method easier to understand, FIG.
Full explanation of phase-to-phase input waveforms and output side waveforms.

In Fig. 5 (2), vl is the UV-V phase 1ti pressure waveform, v2 is the v-W phase voltage waveform, ■3 is the W-U phase voltage waveform, and in the same figure (J3), Ul, U2 , ty
, , is time 1 corresponding to 60° intervals of the 6-phase AC input waveform
(, -15 to G6 operate at a conversion frequency of, for example, 201(l(z), and each conversion unit 01 to G6 opens and closes at a conversion frequency of the corresponding phase-to-phase voltage t'' 20 K)Iz of the six-phase AC input. As shown by the chain line in FIG. 5, each output voltage waveform in one cycle starting from time to when the UV interphase voltage v1 is zero value depends on the output voltage detection signal Sd as detailed in the previous embodiment. The switching transistors of each conversion unit 01 to G6 are controlled by the pulse width determined by comparing the charging voltage of the capacitor 70 with the instantaneous value of the corresponding inter-phase voltage, and these switching transistors completely control the corresponding inter-phase voltage. The wave-rectified sinusoidal voltage opening/closing ti combination is that the output U1 corresponding to the human power V at time 1 (,) is very small in both pulse width and amplitude, and the output U2 corresponding to the input v2 and the input Both outputs U3 corresponding to v3 are almost stationary, and the output U
It is easy to understand that both the pulse width and amplitude are sufficiently large compared to 1.

Next, due to the illustration, O (time 11 corresponding to 30' of U40 of 1 AC input (of course, 2 between times to- and -t)
(Each converter is operating at a conversion frequency of 0 KHz), the output U1 is the input v1
As the voltage rises in a sinusoidal waveform, both the pulse width and amplitude increase, and as the human power v2 reaches the maximum amplitude, both the pulse width and the amplitude of the output U2 reach the maximum. Further, as the input v3 becomes approximately equal to V□ at time t, the output U3 has a pulse width and amplitude approximately equal to the output U1. Each period starting from time t2, t3, . . . t5 will be similarly explained below.

In this example, the frequency of the polyphase AC input is 50H.
z, then each cycle j is divided into 20 Kl (400 equal parts by the exchange frequency of z, and the 400 equal parts are divided into 'fc' and the converting parts 01 to G3?c in each cycle are time-divided using the control method as described above. The sum of the outputs U1 to U3 in each cycle is proportional to the control amount, and is proportional to the error signal that is the difference between the output voltage detection signal vd and the reference 11, and this detection As long as the signal vd is constant, the above formula (6)
It can be seen that the sums of the outputs U1 to U3 in each cycle are all equal.

Therefore, according to equation (6) above, since the instantaneous power Pi on the input side is constant, if the instantaneous power obtained on the output side is constant, a sinusoidal current will flow on the input side, but in this example However, the frequency is sufficiently higher than the frequency of the polyphase AC input.
Since the sum of the outputs Ul to U3 in each period due to the conversion frequency of KHz is approximately constant, it can be easily understood that a sinusoidal current with extremely few harmonic components flows on the input side.

Furthermore, according to this A C-DC conversion, it is possible to obtain a DC output voltage with small ripples, and as can be seen from the above, the switching transistor of the converting section G2 is turned on when the switching transistor of the converting section G1 is turned off. ,
Since the switching transistor of the converter G6 can be controlled to turn on as it turns off, it is easy to increase the frequency, and the utilization rate can be increased to 1/2, which is the maximum duty of the converter used in the embodiment.

In order to make this kind of follow-up control possible, it is necessary to delay the on signal by approximately the same amount of time in order to eliminate the influence of the carrier accumulation time of the switching transistor in the Q section. At the same time, in order to improve the proportionality accuracy of the pulse width with the input voltage, it is desirable to start charging the capacitor 7 before this delay time begins.

Next, a specific example of characteristics obtained by such AC-DC conversion will be described. However, the harmonics range from the 2nd order to the 45th order.

Input and power are 6-phase AC (50Hz) (D200V, output 1E)
In the case of DC48V, 30A: Harmonic component of input voltage -U-V: 1.67%, v-w: 1
．． 41%, W-U: 2,22 input current harmonic component -U: 5.02%, V: 4.07
%, W: 4.75chi, power factor -0,982, efficiency -88.6chi.

FIG. 6 shows a rectifier device for explaining another example of the invention, in which the outputs of the rectifiers D9, Dd and D9'' are connected in series through diodes Dll, Dil' and D1%l, etc. There are two ways to control this rectifier.
Therefore, the first control method is exactly the same as the method of the above embodiment, and the second control method performs all switching of converters 01 to G3 without time division because the outputs are in series. It is possible to control the pulse width by operating the devices so that they are turned on at the same time or part of the conduction period is gravitational. This 20th control method also departs from the basic technical idea of the present invention, that is, the switching element is switched at a conversion frequency that is sufficiently higher than the frequency of the three-phase AC input, and for each cycle of this conversion frequency, 6
The electric power taken out from each phase of the phase AC input to the output side is proportional to the product of the value proportional to the square of the instantaneous voltage value of the three-phase AC and the control voltage, and the electric power is taken out from each of the polyphase AC inputs. The control is performed in proportion to the total force of the force, which is exactly the same as in the previous embodiment.

Therefore, to briefly explain this second control method using FIG. It is controlled with a respective pulse width proportional to the product of the signal Sd and the instantaneous value of the polyphase flow input. Therefore, the outputs LL1, U2 of each phase in each period
．． 03 is the corresponding human power phase v at the beginning time of each cycle
1, the amplitude of V2, v3 and the output voltage detection (i
It is determined by a value proportional to the product of the difference between the T number Sd and the reference voltage and the instantaneous value of the multiphase current input. In this control method, the outputs U, U2, US of the alternating current ``Cr Dg, D9', D9'' of the output li+ are multiplied by M.

Next, the off-line diagram shows a case in which the converters 01 to G6 are configured as a bridge VT by switching elements, and a setter-tap double-wave rectifier circuit is connected in series.

Since the basic Sti control method in this rectifier is exactly the same as the method in the previous embodiment, it will not be described in detail.

Next, in the rectifier shown in FIG. 8, the source circuit of the output τlft is a full-wave rectifier bridge D9, D9', D9'', and one of the rectifying arms d1, dl, dl is connected in parallel to form a first inductor. 2 and the other rectifier arm d2, d2
, d21f - another second intalter in parallel, i
is in series with The basic control method of this device is also the same as in the first embodiment, but each switching section has a 1 flW inductor and the maximum duty of the voltage waveform supplied to 2 or L2' is limited to 1. The current is taken out from separate inductors L2 and L2 so that the maximum duty of 172 of the output voltage waveform of each converter can be used by dividing it into positive and negative cycles. It is possible to improve the rate.

Next, FIG. 9 shows six bidirectional switching elements S1 to S.
A basic on-on type six-phase rectifier R4 connected to an eight-phase six-phase full-wave rectifier configuration is shown, and the smoothing inductor L2 has an inductance large enough to maintain a constant current over the conversion period.

The basic control technology is the same as the control method described in connection with Fig. 2, so it will not be described in detail. An example of a sequence created by Le0 is shown below.

To explain and explain the above sequence, the UV phase til
In the phase of l voltage from 0 to π/3, first switching elements S1 and 85, then 85 and 86, and then S3 and S4.
The 11-member switching operation is repeated at a frequency sufficiently higher than the frequency of the 6-phase AC input and according to the control method described above. During the period of π/6 to 2π/3 of the UV phase voltage, the switching elements S1 and S5, S5 and S6, and S6 and Sl repeat the switching operation in the same manner. Similarly, corresponding switching elements perform crab switching operations in other phases as well.

Next, FIG. 10 shows an example of an on-off type 6th Mar device, and since the method of controlling the switching elements S1 to S6 and the switching sequence are the same as those of the device of FIG. 9, a description thereof will be omitted. In this device, energy is stored in the inductor L3 when one pair of switching elements 81 to S6 is on, and energy is stored in the inductor L3 when all of the switching elements 81 to S6 are off.

/I L3 O'Ili product energy is supplied.

In FIG. 11, each of the bidirectional switching elements 81 to S6 in FIG.
and 86', and the input and output sides are DC-insulated by a transformer T. The method for formula 11 is to add an operation to obtain a negative output (180-phase shifted operation) to the operation to obtain a positive output in the itr++ control in the case of Fig. 9,
In the above table, the sequences which are out of phase by 180'' are controlled by zero, and positive and negative outputs are applied to the transformer every /6 conversion cycle.

Although we have described a simple example of the 1i21+ control, if the secondary side of the transformer Z is connected in series to the 7k of the fc rectifier as shown in Figure 6 and the off-line diagram, the switching elements of each converter will be Pulse width control is achieved by fully controlling the pulse width before and after the center of the pulse width as a reference, and when connected in parallel on the output side as in the rectifier shown in Figures 2 and 8. By turning on the switching element when the signal has a large or small pulse width, the symmetry of the control improves, and the power factor can be improved and the harmonics divided into the input current can be reduced. , the characteristics can be significantly improved.

Furthermore, in the embodiment described above, the switching element of the first converter is turned on based on the basic signal generated at a fixed period, and when the switching element of the next converter is turned off, the switching element of the next converter starts the switching operation. The 111st @conversion unit performed a switching operation as described above, but in order to further reduce the waveform distortion of the rickshaw style, the input voltage of each phase is detected and the phase with a large detected voltage or the small 8 It is also possible to operate the f switching section in the order of the phases.

In the above explanation, the input voltage was used as the reference sine waveform applied to the 10,000 input terminals of the comparator, but instead of this, the PLL
It is also possible to use a sine wave generator that generates a sine wave signal synchronized with the input frequency using a circuit or the like.

Further, in an actual device, the current flowing from the input to the output via the switching element varies within the conversion period depending on the leakage inductance in each conversion section and the value of the output side inductor. As a control method for reducing this influence, there is a method of controlling so that the value obtained by integrating the current t flowing through the switching element is proportional to the absolute value of the instantaneous value of the input voltage of the corresponding phase. However, since the integral value depends on the pulse width and the current value, the value fluctuates considerably, and control may become difficult. In this case, the value k corresponding to the fall of the current flowing through the switching element from human power and the control amount
By dividing the value by nt, fluctuations in the 8(min) value can be made small enough, making fli control easier.

The controlled amount in this case is a value whose reciprocal is proportional to the output surface force, and is an opposite pressure to the controlled amount in the case of +Iil.

According to Mr. Bare'C, the switching element is switched at a conversion frequency considerably higher than the frequency of the polyphase AC, and the energy taken out from each 111 to the output iji for each cycle by the conversion frequency is the same as that of the polyphase AC. Since it is controlled so that it is proportional to the product of the square of the control amount and the sum of the energy taken out from each phase is proportional to the control amount, the input of high μ-frequency components is very small. In order to be able to flow current, it is possible to prevent d-conducting disturbances due to harmonic components in single-signal circuits, etc., and to have a constant color power voltage with small high-frequency ripples. You can get it. Furthermore, according to the present invention, the h-school style circuit at the input end and the change! There is a smooth line between the city and the capital.
Since there is no need to provide a circuit, the device can be made smaller.

Of course, it is also possible to use a Zyristor or the like in addition to a transistor as a switching element, and although the above embodiments have described a symmetrical 6-phase AC input, other polyphase inputs or 2-phase 3-phase inputs can also be used. Similarly, even if the input waveform is slightly deformed, ripples at the output can be reduced by controlling the energy flow at the input to be constant using the feedback loop.

[Brief explanation of drawings]

Fig. 1 is a diagram for explaining the conventional rectification value, Fig. 2 is a diagram showing an example of the rectifier according to the present invention, and Fig. 6 is a block diagram of its control circuit. Figure 4 is a diagram showing signals indicating the operation timing of each part, Figure 5 is a diagram to explain waveforms on the output side, Figures 6 to 8, and Figure 9. FIGS. 1 to 11 respectively show "1" - according to the present invention.
It is a figure which shows the different example of a rectifier. Mea1-Re c5... Rectifier circuit 01-G6... Conversion section F... Load Con... Control circuit 1... Leveling signal generator 2.2', 2''-!J upper settle 2 formation circuit 3.6',
6''...delay circuit 4.4', 4''...cardiac latch times 1゛65...OR
1g1 path 8...Error amplifier 9...FliIJ controlled constant current source 10.10', 10''...Comparator patent applicant Sho Origin Electric Co., Ltd. 11 Figure procedure amendment December 27, 1981 Mr. Commissioner of the Japan Patent Office1, Indication of the case: Patent Application No. 173647 of 19822, Title of the invention: Rectifier device 3, Relationship with the person making the amendment case Patent applicant address: 1-18-1 Takada, Toshima-ku, Tokyo 〒17
1 Telephone (983) 7111 (Main) 4, Date of amendment order Voluntary issue 5, Subject of amendment ] Detailed explanation of the invention column 6, Contents of amendment (1) Page 6, line 12, line 1 of the present specification Direct current...
・Correct "slip" as shown below. "If the load on the AC input is linear and the DC output side is controlled so as to obtain a constant energy," 9 amended as follows. "...Returns to C1'..." (6)
Same book, page 16, line 4 F...It rises in proportion to the size. ' shall be corrected as follows. "...rises at a rate of increase proportional to its size." (4) Page 16 of the same book, lines 14 to 16: "...
. . . Inductor L2, . . . conducts current. ' shall be corrected as follows. "...The energy stored in the inductor L2 flows through the diode D10 'i:." (5) If F is constant on page 17 of the same book, line O 8 and line O 9, then・−
``It flows, but'' is corrected as follows. If the above-described pulse width control is applied to the input frequency and the conversion frequency is sufficiently high, a sinusoidal current will naturally flow at the input end simply by removing the high frequency component. (6) On the same page of the same book, line 11, ``...is constant, so enter'' is amended as follows. "...Since it is constant, it is possible to obtain a DC output voltage with small ripples." V) Same ■ Lines 14 and 15 of the same page I...
・Delete "Ripple ゛°・°・... can do it." (8) On the same page of the same book, line 20, ``...can be controlled/high frequency...'' is corrected as follows. 1" ... can be controlled, so the output side can be connected in parallel using an on-on type controller, and high frequency can be achieved at low voltage and large current..." (9 ) On page 23 of the same edition, line 6 of ``...sequence and figure 9...'' wo [...sequence is figure 9...
...” is corrected.

Claims (1)

[Claims] (1) A plurality of rectifier circuits that rectify the voltage between each phase of the multiphase AC input, and directly convert the rectified sinusoidal voltage from each rectifier circuit at a conversion frequency that is sufficiently higher than the frequency of the multiphase AC input. A plurality of transformers each having a switching element for switching, each primary winding connected in series to each of the switching elements, and a plurality of secondary windings connected in series or parallel to each other, each of the secondary windings A rectifier comprising: a rectifier that rectifies each of the voltages produced; an inductor and a capacitor that smooth the output of the rectifier; and a control circuit that controls the switching element. Q) A plurality of rectifier circuits that rectify the phase-to-phase voltage of the polyphase AC input, and the rectified sinusoidal voltages from the respective rectifier circuits are directly switched at a conversion frequency sufficiently higher than the frequency of the polyphase AC input. A bridge-connected switching element, a plurality of transformers each having a similar primary winding and a secondary winding connected to the switching element, and a pair of transformers for double-wave rectification of the voltage between the secondary windings of the transformer. □ A plurality of double-wave type rectifier circuits each having a rectifying arm, separate smoothing inductors and smoothing capacitors connected in series between each of the pair of rectifying arms and one output terminal, and the switching element. A rectifier device characterized by being fully equipped with a control circuit for controlling the device.