JPS643373B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a thyristor drive circuit.

A thyristor has three or more built-in PN junctions.
A semiconductor device that has two stable states, on and off, and can be switched from off to on or vice versa;
Includes TRIAC etc.

When turning on a thyristor, by supplying a large-level gate current with a steep rise, the thyristor will rise sharply from off to on, and the loss when turned on can be reduced. In addition, when a load including reactance is connected to the thyristor, unless a deep negative bias is always applied between the gate and cathode during the off period, (dV/dt) (critical off-voltage rise rate during commutation) There is a risk that the anode voltage may turn on even outside the normal on period due to a change in the anode voltage exceeding .
Because of these characteristics of the thyristor, conventional drive circuits have had the disadvantage of requiring a large level of gate current to be supplied in both positive and negative directions, resulting in large drive power.

Further, as a specific conventional drive circuit, a configuration in which a parallel circuit of a capacitor and a resistor is connected to the gate is known. For example, in the case of GCS, it is necessary to supply a gate current with a peak value of 2 [A], and as the above-mentioned resistor, one with a small resistance value must be used. In order to ensure negative bias when off, a capacitor with a large capacity of several tens of μF is required, and in reality, a conventional drive circuit that connects a parallel circuit of a capacitor and a resistor is inappropriate.

The present invention is capable of supplying a large-level gate current with a steep rise to the thyristor when it is on, and has the function of always applying a deep reverse bias between the gate and cathode during the off period, while reducing drive power. The purpose is to realize a circuit.

An embodiment of the present invention will be described below with reference to the drawings. In this example, the present invention is applied to an induction heating device that generates high-frequency magnetic flux through a work coil and heats cooking utensils (pots, frying pans) placed close to the work coil through eddy current loss. This is what I did.

FIG. 1 shows the configuration of an induction heating device. In the same figure, 1 indicates a GCS (Gate Controlled Switch), and a power supply voltage formed by a rectifier circuit 2 having a diode bridge configuration and a smoothing capacitor 3 is connected to a It is applied between the anode and cathode of GCS1 via. 5 is a power switch.
Further, a series circuit of a work coil 6 and a capacitor 7 and a damper diode 8 are inserted in parallel between the anode and cathode of the GCS 1.

The work coil 6 has a spiral planar pattern, and metal cooking utensils are placed close to the work coil 6.

The oscillation output of a VCO (voltage controlled oscillator) indicated by 9 is supplied to a drive circuit 10 to which the present invention is applied, and the output of the drive circuit 10 is supplied to the gate of GCS1. An output control circuit 11 is provided in connection with the VCO 9, and by moving an output adjustment volume 12, the oscillation frequency of the VCO 9 is changed.
The output power is made variable. Increasing the oscillation frequency of VCO9 increases the output power.

The operation of the output circuit including GCS1 will be explained with reference to FIGS. 2 and 3. A gate voltage V _G having a period T shown in FIG. 3A and formed as described below is applied from the drive circuit 10 between the gate and cathode of the GCS1. Of this period T, t ₁ to _{t 3} are GCS1
The on section is the on section, the off section is from _t3 to _t5 , and the damper section is from _t3 to _t4 in this off section.

The equivalent circuit of the on section is shown in FIG. 2A, and when the direction of the current is determined as shown, the current I _L (FIG. 2B) flows through the chiyoke coil 4.
On the other hand, since the capacitor 7 is charged in the direction shown, a current I _W (FIG. 3E) having a resonance frequency determined by the work coil 6 and the capacitor 7 flows through the work coil 6. The interval from t ₁ to _{t 2} in FIG. 3 is a half period of the resonant frequency. Therefore, in the on state
The anode current I _A of GCS1 flows as shown in Fig. 3C, and the anode voltage V _A is approximately 0 [V] as shown in Fig. 3F (strictly speaking, between the anode and cathode when GCS1 is on) voltage (3V) is present)
becomes.

When the gate voltage V _G falls to negative polarity at t ₃ ,
GCS1 is turned off and the damper section _t3 to _t4 begins. The equivalent circuit at this time is shown in FIG. 2B. The damper diode 8 is turned on by the charge in the capacitor 7, and the damper _current I _D shown in FIG. Then, a current _IW shown in FIG. 3E flows through the work coil 6.
The anode voltage V _A of the GCS 1 is made negative by the voltage drop across the diode 8 .

Capacitor 7 discharges and damper diode 8
When GCS1 is turned off at _t4 , the electromagnetic energy stored in work coil 6 generates a large-level anode voltage V _A as shown in FIG. 3F. The whisker-like voltage at the rise of the anode voltage V _A is the damper diode 8
caused by the accumulated charge of The equivalent circuit at this time is shown in FIG. 2C, and the work coil 6
A current I _W flows through the capacitor 7, and the capacitor 7 is discharged to the polarity shown. At t ₅ , when the forward gate voltage V _G is applied to GCS1 again,
Repeat the above action.

Now, if the oscillation frequency of VCO 9 is increased, the period T becomes shorter, and the period in which both GCS 1 and damper diode 8 are turned off becomes shorter, and the anode voltage decreases as shown by the dashed line in Fig. 3. The level of V _A becomes higher. For this reason,
The level of the anode current I _A and the level of the current I _W flowing through the work coil 6 become larger as shown by the dashed line. Further, the number of GCS1s turned on per unit time also increases by shortening the period. The output power can be increased by these two effects.

FIG. 4 shows an example configuration in which the present invention is applied to the drive circuit 10 described above. In the same figure,
Reference numeral 13 denotes a pulse transformer whose primary and secondary sides are tightly coupled; one end of its primary coil 14a is connected to a positive power supply terminal, and the other end is connected to the collector of a switching transistor 15.
The emitter of the transistor 15 is grounded, and the output pulse of the VCO 9 (FIG. 5A) is supplied to the terminal 16 connected to its base. Trance 13-2
One end of the secondary coil 14b is grounded, and the other end is connected to the gate of the GCS 1 via a resistor 17 with a small value. A capacitor 18 is connected in parallel with the secondary coil 14b, and a capacitor 19 and a resistor 20 are inserted in parallel between the gate of the GCS 1 and the ground. Resistor 20 is for discharging capacitor 19.

In the above configuration, when a positive pulse is applied to terminal 16 as shown in FIG. 5A, transistor 15 is turned on. When the transistor 15 is on, when the pulse transformer 13 is viewed from the secondary side, there is almost only leakage inductance. The leakage inductance is small due to the tight coupling, and the gate current I _G flows into the gate of the GCS1 via this leakage inductance and the resistor 17. At this time,
The leakage inductance and the capacitor 18 form a resonant circuit, and ringing occurs in the gate current _IG . In other words, as shown in FIG. 5B, the gate current I _G
The rise is steep and reaches a large level, and after that it becomes a damped oscillation. The value of the capacitor 18 is set so that the period of this ringing is 1/10 or less of the period T of the pulse applied to the terminal 16. In this way, the gate current I _G rises steeply, so that the GCS 1 is turned on reliably. In addition, the gate voltage V _G when on is shown in Figure 5 C.
The voltage between the gate and cathode of GCS1 is as shown by _t1 to _t3 .

Next, when the transistor 15 is turned off, a back electromotive force is generated between both ends of the secondary coil 14b. Therefore, the gate voltage V _G decreases to a negative value immediately after t ₃ in FIG. 5C. At this time, the minority carriers accumulated in the gate of GCS1 are discharged, so the gate current
I _G becomes negative. Because of this discharge current, GCS1
The gate voltage V _G is kept constant for a predetermined period. When the discharge ends, the gate current I _G becomes zero.

Further, in the section where the transistor 15 is turned off, the primary side of the pulse transformer 13 is open, and the inductance seen from the secondary side becomes the inductance of the secondary coil 14b. This secondary coil 14b, resistor 17, and capacitors 18 and 19 constitute a resonant circuit when transistor 15 is off. Therefore, after the gate voltage V _G of the GCS 1 is kept constant for a predetermined period as described above, the secondary coil _14b and the capacitors ₁₈ and 1 are
It resonates at the resonant frequency determined by 9. This resonance period is set so that the gate voltage V _G does not become positive during the off period of the transistor 15, and so that the bias in the negative direction is deepest at t ₄ and immediately after the end of the damper period as described below. It is selected so that it is 1/3T or more with respect to the period T.
This resonant waveform is a combination of the resonant circuit of the secondary coil 14b and the capacitor 18, and the resonant waveform of the resonant circuit of the secondary coil 14b, the resistor 17, and the capacitor 19.
acts as a damping resistance and suppresses the peak value, so the slopes of the section t ₃ -t ₄ and the section t ₄ -t ₅ in FIG. 5C are different. At _t4 , when the damper section ends, the anode voltage V _A of GCS1 rises sharply to a large level as described above.
A noise voltage as shown in an enlarged view in FIG. 5D appears at the gate of GCS1 via the junction capacitance of GCS1. The level of this noise voltage is made small because the impedance between the gate and cathode is lowered by the capacitor 19 inserted between the gate and cathode of GCS1, and at the same time, the gate voltage V _G is lowered in the negative direction. This prevents the rate of increase in the anode voltage V _A from exceeding (dV/dt) and turning on the GCS1.
In other words, as mentioned above, the secondary coil 14 is adjusted so that the gate voltage V _G becomes the deepest negative bias at t ₄ .
b, the resonant frequency of the resonant circuit composed of the resistor 17 and capacitors 18 and 19 is selected.

Although the resonant period of this resonant circuit is selected to be 1/3T or more as described above, it is also not preferable that it be extremely long.

As described above, according to the present invention, when the GCS 1 is on, the resonance period of the resonant circuit composed of the leakage inductance of the pulse transformer 13 and the capacitor 18 is set to 1/10T with respect to the period T of the pulse supplied to the transistor 15. Since the setting is as follows, the large level gate current that rises sharply can be suppressed.
It can be supplied to GCS1 and GCS1 can be turned on reliably. In addition, when GCS1 is turned off, the resonance period of the resonant circuit consisting of the secondary coil 2b of the pulse transformer, the resistor 17, and the capacitors 18 and 19 is set to be 1/3T or more with respect to the above-mentioned T. Since the gate voltage V _G is set to be the deepest bias in the negative direction at the time t ₄ when the anode voltage V _A of the gate voltage V A rises sharply, it is possible to prevent GCS1 from being turned on by mistake. Therefore, it is no longer necessary to constantly flow a large level of gate current in the ON period, and since resonance is used to generate a deep bias in the negative direction, loss does not increase, and overall the drive power is reduced compared to the conventional one. It can be reduced to about (1/3 to 1/2) more.

FIG. 6 shows another example of a drive circuit to which the present invention is applied. The difference from FIG. 4 is that one end of the primary coil 14a of the pulse transformer 13 is connected to the collector of the PNP transistor 21, its emitter is connected to the power supply terminal +B, and a diode is connected in parallel with the resistor 17. 22 and GCS1
A resonant current is caused to flow through the diode 22 when the diode 22 is turned off. By bypassing the resistor 17 with the diode 22, the loss during off-time can be further reduced.

An example of values of circuit elements of a drive circuit to which the present invention is applied is shown below.

Capacitor 18; 0.015 [μF] Capacitor 1
9; 0.01 [μF] Resistor 17; 6.8 [Ω] Resistor 20; 56 [Ω]

[Brief explanation of drawings]

FIG. 1 is an overall connection diagram of an embodiment of the present invention, FIGS. 2 and 3 are equivalent circuit diagrams and waveform diagrams of each part used to explain the operation of the output circuit, and FIGS. 4 and 5 are diagrams of the present invention. A connection diagram of an example of a drive circuit to which the present invention is applied and a waveform diagram used for its explanation. FIG. 6 is a connection diagram of another example of a drive circuit to which the present invention is applied. 1 is a GCS, 4 is a chiyoke coil, 6 is a work coil, 8 is a damper diode, 10 is a drive circuit, 13 is a pulse transformer, and 18 and 19 are capacitors.

Claims (1)

[Claims] 1. A switching element is connected to the primary coil of the transformer, and a thyristor is connected to the secondary coil of the transformer, and the thyristor is turned on by turning the switching element on and off.
In the thyristor drive circuit configured to turn off, one end of the secondary coil is connected to the gate of the thyristor via a resistor, and the first
is connected to a reference potential via a capacitor, the gate is connected to the reference potential via a second capacitor, and the switching element is configured by the leakage inductance of the transformer when the switching element is turned on and the first capacitor. The resonant circuit has a resonance period of 1/10 or less of the on-off period of the switching element, and is composed of the secondary coil, the resistor, and the first and second capacitors when the switching element is off. A thyristor drive circuit characterized in that the resonant period of the resonant circuit is 1/3 or more of the on/off period of the switching element.