WO2024255895A9 - 信号发射、校准、补偿及收发链路，集成电路、电磁波传感器及设备 - Google Patents

信号发射、校准、补偿及收发链路，集成电路、电磁波传感器及设备 Download PDF

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Publication number
WO2024255895A9
WO2024255895A9 PCT/CN2024/099438 CN2024099438W WO2024255895A9 WO 2024255895 A9 WO2024255895 A9 WO 2024255895A9 CN 2024099438 W CN2024099438 W CN 2024099438W WO 2024255895 A9 WO2024255895 A9 WO 2024255895A9
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Prior art keywords
signal
link
compensation
digital
analog
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PCT/CN2024/099438
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English (en)
French (fr)
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WO2024255895A1 (zh
Inventor
张展
陈嘉澍
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Calterah Semiconductor Technology Shanghai Co Ltd
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Calterah Semiconductor Technology Shanghai Co Ltd
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Priority claimed from CN202410599441.1A external-priority patent/CN119154895B/zh
Application filed by Calterah Semiconductor Technology Shanghai Co Ltd filed Critical Calterah Semiconductor Technology Shanghai Co Ltd
Priority to KR1020247043516A priority Critical patent/KR20260020021A/ko
Priority to EP24822842.1A priority patent/EP4531291A1/en
Priority to US18/879,597 priority patent/US20250385742A1/en
Publication of WO2024255895A1 publication Critical patent/WO2024255895A1/zh
Publication of WO2024255895A9 publication Critical patent/WO2024255895A9/zh
Anticipated expiration legal-status Critical
Ceased legal-status Critical Current

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Classifications

    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B17/00—Monitoring; Testing
    • H04B17/10—Monitoring; Testing of transmitters
    • H04B17/11—Monitoring; Testing of transmitters for calibration
    • H04B17/12—Monitoring; Testing of transmitters for calibration of transmit antennas, e.g. of the amplitude or phase
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B17/00—Monitoring; Testing
    • H04B17/10—Monitoring; Testing of transmitters
    • H04B17/11—Monitoring; Testing of transmitters for calibration
    • H04B17/14—Monitoring; Testing of transmitters for calibration of the whole transmission and reception path, e.g. self-test loop-back
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S7/00—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
    • G01S7/02—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00
    • G01S7/35—Details of non-pulse systems
    • G01S7/352—Receivers
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S7/00—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
    • G01S7/02—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00
    • G01S7/28—Details of pulse systems
    • G01S7/285—Receivers
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S7/00—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
    • G01S7/02—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00
    • G01S7/35—Details of non-pulse systems
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S7/00—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
    • G01S7/02—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00
    • G01S7/40—Means for monitoring or calibrating
    • G01S7/4004—Means for monitoring or calibrating of parts of a radar system
    • G01S7/4008—Means for monitoring or calibrating of parts of a radar system of transmitters
    • G—PHYSICS
    • G01—MEASURING; TESTING
    • G01S—RADIO DIRECTION-FINDING; RADIO NAVIGATION; DETERMINING DISTANCE OR VELOCITY BY USE OF RADIO WAVES; LOCATING OR PRESENCE-DETECTING BY USE OF THE REFLECTION OR RERADIATION OF RADIO WAVES; ANALOGOUS ARRANGEMENTS USING OTHER WAVES
    • G01S7/00—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00
    • G01S7/02—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00
    • G01S7/41—Details of systems according to groups G01S13/00, G01S15/00, G01S17/00 of systems according to group G01S13/00 using analysis of echo signal for target characterisation; Target signature; Target cross-section
    • G01S7/418—Theoretical aspects
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B1/00—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission
    • H04B1/005—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission adapting radio receivers, transmitters andtransceivers for operation on two or more bands, i.e. frequency ranges
    • H04B1/0096—Details of transmission systems, not covered by a single one of groups H04B3/00 - H04B13/00; Details of transmission systems not characterised by the medium used for transmission adapting radio receivers, transmitters andtransceivers for operation on two or more bands, i.e. frequency ranges where a full band is frequency converted into another full band
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B17/00—Monitoring; Testing
    • H04B17/10—Monitoring; Testing of transmitters
    • H04B17/15—Performance testing
    • H—ELECTRICITY
    • H04—ELECTRIC COMMUNICATION TECHNIQUE
    • H04B—TRANSMISSION
    • H04B17/00—Monitoring; Testing
    • H04B17/20—Monitoring; Testing of receivers
    • H04B17/21—Monitoring; Testing of receivers for calibration; for correcting measurements

Definitions

  • the embodiments of the present disclosure relate to, but are not limited to, the technical field of electromagnetic wave sensors, and specifically relate to a signal transmission, calibration, compensation and transceiver link, an integrated circuit, an electromagnetic wave sensor and equipment.
  • the signal transmission link adopts the analog phase shifter architecture, it has problems such as low phase modulation precision (Resolution) and phase modulation accuracy (Accuracy), which makes it unable to meet various requirements of high precision and accuracy.
  • Resolution phase modulation precision
  • Acuracy phase modulation accuracy
  • the embodiments of the present disclosure provide a signal transmission, calibration, compensation and transceiver link, IQ mixer, integrated circuit, electromagnetic wave sensor and equipment, etc., based on the digital phase shifter architecture to form a signal transceiver link, and corresponding calibration and compensation link and other technologies, so as to effectively improve the phase adjustment precision (Resolution) and phase adjustment accuracy (Accuracy), and at the same time avoid the operation of off-line calibration of links and devices such as phase shifters in the transmission link, thereby reducing the complexity and difficulty of engineering implementation.
  • it can also effectively reduce the transmission link area and loss of the phase shift architecture, improve the stability of the system, and reduce the channel coupling degree.
  • An embodiment of the present disclosure provides a signal transmission link, which is applied to an electromagnetic wave sensor.
  • the transmission link includes an analog signal source and a digital phase shifter.
  • the analog signal source can be configured to provide an initial analog signal
  • the digital phase shifter can be configured to provide a phase-shifted signal generated in a digital domain, and to phase-shift the initial analog signal based on the phase-shifted signal, so as to perform a preset phase shift operation on the initial analog signal.
  • the signal transmission link further includes a transmitting antenna, which may be configured to radiate the phase-shifted initial analog signal to a preset spatial region.
  • the digital phase shifter includes a digital phase shift signal source, a digital-to-analog converter and a mixer.
  • the digital phase shifter can be configured to generate a digital phase shift signal.
  • the digital-to-analog converter can be configured to convert the received digital phase shift signal into an analog phase shift signal.
  • the mixer can be configured to perform a mixing operation on the received initial analog signal using the received analog phase shift signal so as to perform a preset phase shift operation on the initial analog signal.
  • the digital phase-shifted signal source includes a direct digital frequency synthesizer
  • the digital-to-analog converter is an IQ digital-to-analog converter
  • the mixer is an IQ mixer.
  • the digital phase-shifted signal is a single-tone signal, and the initial analog signal is a swept-frequency signal; or, the digital phase-shifted signal is a swept-frequency signal, and the initial analog signal is a single-tone signal.
  • the signal transmission link transmits a frequency modulated continuous wave signal.
  • An embodiment of the present disclosure also provides a signal transmission link, comprising a signal transmission main path and a signal calibration link integrated in the same integrated circuit; wherein: the signal calibration link is configured to calibrate the signal transmission main path to obtain compensation information; and the signal transmission main path is configured to generate a radio frequency transmission signal after performing a compensation operation according to the compensation information, so as to achieve target detection and/or communication.
  • the compensation information includes at least one of a harmonic distortion compensation parameter, a local oscillator leakage compensation parameter, and an orthogonal imbalance compensation parameter.
  • the main signal transmission path includes a first signal source and a phase shifter; wherein the first signal source is configured to generate a first analog signal; and the phase shifter is configured to perform frequency shifting and/or phase shifting on the first analog signal to form a radio frequency transmission signal.
  • the phase shifter when the phase shifter is a non-orthogonal architecture, the phase shifter includes a second signal source and a transmitting end mixer, wherein the second signal source is configured to generate a second analog signal, and the transmitting end mixer is configured to perform mixing processing on the first analog signal and the second analog signal to form the radio frequency transmission signal;
  • the phase shifter includes: a second signal source, a digital-to-analog conversion module and a transmitting-end mixer; wherein the second signal source is configured to generate a first digital signal; the digital-to-analog conversion module is configured to convert the first digital signal into a second analog signal; and the transmitting-end mixer is configured to perform frequency shifting and/or phase shifting on the first analog signal based on the second analog signal to form the RF transmission signal.
  • the main transmitting path also includes a compensation circuit, wherein the signal input end of the compensation circuit is connected to the second signal source, and the signal input end is connected to the phase shifter, and the compensation circuit is used to combine the compensation signal and the signal output by the second signal source and then output it.
  • the compensation signal used by the compensation circuit is two orthogonal signals; when the signal output by the main signal transmission path is not two orthogonal signals, the compensation signal used by the compensation circuit is a signal of the same type as the signal output by the second signal source, wherein the signal type is a digital signal or an analog signal.
  • the compensation circuit includes a compensation signal generator and an adder, wherein: the compensation signal generator is configured to generate the compensation signal; the adder is connected to the compensation signal generator and the second signal source, and is used to perform a signal superposition operation on the signal output by the second signal source and the compensation signal output by the compensation signal generator.
  • the compensation signal generator when the signal output by the main signal transmission path is a two-way orthogonal signal, the compensation signal generator includes at least one of a harmonic compensation signal unit, an orthogonal imbalance compensation signal unit and a local oscillator leakage compensation signal unit; when the signal output by the main signal transmission path is not a two-way orthogonal signal, the compensation circuit includes at least one of a harmonic compensation signal unit and a local oscillator leakage compensation signal unit; wherein the compensation signal generated by the harmonic compensation signal unit is used to eliminate the harmonic signal of the main frequency signal in the main signal transmission path; the compensation signal of the local oscillator leakage compensation signal unit is used to compensate for the leakage signal generated by the transmitting end local oscillator signal in the main signal transmission path; and the compensation signal generated by the orthogonal imbalance compensation signal unit is used to compensate for the mirror signal of the main frequency signal in the main signal transmission path.
  • the compensation signal generated by the harmonic compensation signal unit has the same frequency and amplitude as the harmonic signal in the signal transmission main path, and has an opposite phase.
  • the harmonic compensation signal unit includes an n-th power module or an n-fold frequency signal generator; an n-th power module, wherein the signal input end of the n-th power module is connected to the signal output end of the second signal source, and is used to generate a signal with a frequency n times the frequency of the signal output by the second signal source, thereby obtaining the compensation signal; an n-fold frequency signal generator, which is used to generate a signal with a frequency n times the frequency of the signal output by the second signal source, thereby obtaining the compensation signal, wherein the value of n is a positive integer.
  • the value of n is an odd number.
  • the value of n is 3.
  • the compensation signal generated by the local oscillator leakage compensation signal unit is generated based on a leakage signal corresponding to the first analog signal used by the phase shifter.
  • the compensation signal generated by the local oscillator leakage compensation signal unit has the same frequency and amplitude as the leakage signal but an opposite phase.
  • the compensation signal used by the orthogonal imbalance compensation circuit has the same frequency, the same amplitude, and the opposite phase between the image signal corresponding to the signal transmission main path and the image signal corresponding to the desired signal generated by the signal transmission main path.
  • the compensation signal used by the orthogonal imbalance compensation circuit is determined according to a signal output by the second signal source and a frequency-inverted complex conjugate signal of the signal output by the second signal source.
  • a method for obtaining the compensation signal generated by the orthogonal imbalance compensation circuit includes: obtaining the product of a preset pre-compensation coefficient and the complex conjugate signal to obtain an adjustment signal corresponding to the complex conjugate signal; and calculating the difference between the signal output by the second signal source and the adjustment signal to obtain the compensation signal generated by the orthogonal imbalance compensation circuit.
  • the pre-compensation coefficient is determined based on a ratio of an amplitude of a desired signal to an amplitude of an image signal corresponding to the desired signal.
  • the signal calibration link includes a signal receiving link corresponding to the signal transmission main path, wherein the signal receiving link is used to receive and process the echo signal corresponding to the FMCW RF transmission signal; wherein: the signal receiving link is configured to obtain a sampling signal from the signal transmission main path, and obtain configuration information of the compensation signal based on the sampling signal; wherein the frequency of the transmitting end local oscillator signal used in the signal transmission main path and the receiving end local oscillator signal used in the signal receiving link are different.
  • the signal calibration link further includes: a frequency adjustment circuit configured to adjust the frequency of at least one of the sampling signal and the receiving end local oscillator signal.
  • the signal input end of the frequency adjustment circuit is connected between the signal transmission main path and the transmitting antenna, and the signal output end of the frequency adjustment circuit is connected between the signal receiving link and the receiving antenna, wherein the frequency adjustment circuit is configured to adjust the frequency of the RF transmission signal output by the signal transmission main path; correspondingly, the signal receiving link is configured to obtain a sampling signal from the frequency adjustment circuit, and obtain configuration information of the compensation signal according to the sampling signal; or, the frequency adjustment circuit is connected between a receiving end mixer and a receiving end local oscillator in the signal receiving link, and is configured to adjust the frequency of the receiving end local oscillator signal, wherein the receiving end local oscillator is used to generate a receiving end local oscillator signal, and the receiving end mixer is used to use the receiving end local oscillator signal to demodulate the received signal; the signal receiving link is connected between the signal transmission main path and the transmitting antenna, and is configured to obtain a sampling signal from the signal transmission main path, and process the sampling signal using the signal output by the frequency adjustment circuit to obtain configuration
  • the signal calibration link is a signal receiving link that supports processing of echo signals including the two orthogonal signals.
  • the signal calibration link is provided with an orthogonal processing circuit and two signal receiving links, wherein each signal receiving link does not support processing of orthogonal echo signals; wherein the orthogonal processing circuit is configured to perform RF processing on the received orthogonal signal to obtain two signals, and send the two signals to the two signal receiving links respectively.
  • the signal calibration link the first input end of the signal calibration link is connected between the voltage-to-current converter and the current switch in the transmitting end mixer in the signal transmission main path, the second input end is between the signal transmission main path and the transmitting antenna, and the signal output end is connected to the compensation circuit in the signal transmission main path, and is used to obtain the signal in the signal transmission main path from at least one of the first input end and the second input end, and determine the compensation information based on the obtained signal.
  • the signal calibration link includes a calibration demodulator, a multiplexer and a calibration module; wherein: the calibration demodulator is configured to obtain the signal in the signal transmission main path from the second input terminal and perform demodulation processing; the multiplexer has two signal input terminals and one signal input terminal, one of which is connected between the voltage-current converter and the current switch in the transmitting mixer, and the other is connected to the signal input terminal of the calibration demodulator, for outputting a signal corresponding to one of the two signal input terminals; the calibration module is configured to determine the compensation information according to the signal output by the multiplexer.
  • the acquisition signal transmitted by the signal calibration link is two orthogonal signals.
  • the acquisition signals transmitted by the signal calibration link are two orthogonal signals
  • the signal calibration link also includes: an analog-to-digital converter, connected between the signal acquisition circuit and the calibration module, and configured to perform analog-to-digital conversion processing on the acquisition signals output by the signal acquisition circuit.
  • the embodiments of the present disclosure also provide a signal transceiver link, including a signal transmission link as described in any embodiment of the present disclosure, and a signal receiving link;
  • the signal receiving link includes a receiving end mixer, an analog-to-digital converter and a digital signal processing module; wherein the receiving end mixer can be configured to down-convert the received echo signal based on the received receiving end local oscillator signal to obtain an analog intermediate frequency signal, the analog-to-digital converter can be configured to perform analog-to-digital conversion on the received intermediate frequency signal to obtain a digital intermediate frequency signal, and the digital signal processing module can be configured to process the digital intermediate frequency signal to obtain a target parameter; the echo signal is a signal formed when the signal transmitted by the signal transmission link is reflected and/or scattered by a target object.
  • the receiving mixer is a real mixer, and the analog-to-digital converter is a real analog-to-digital converter; or, the receiving mixer is an orthogonal mixer, and the analog-to-digital converter is an orthogonal analog-to-digital converter.
  • the receiving end local oscillator signal is a swept frequency signal; or, the receiving end local oscillator signal is a single tone signal.
  • An embodiment of the present disclosure also provides a signal calibration link, comprising a signal transceiver link as described in any embodiment of the present disclosure; the receiving antenna connection port of the signal receiving link is connected to the transmitting antenna connection port of the signal transmitting link, and the signal receiving link can be configured to calibrate the signal transmitting link.
  • the signal calibration link also includes a BIST module, which is arranged between the local oscillator signal source and the receiving end mixer; wherein the BIST template can be configured to mix the received local oscillator signal based on a preset frequency deviation signal so that there is a preset difference frequency between the local oscillator signal received by the receiving end mixer and the local oscillator signal of the signal transmission link.
  • a BIST module which is arranged between the local oscillator signal source and the receiving end mixer; wherein the BIST template can be configured to mix the received local oscillator signal based on a preset frequency deviation signal so that there is a preset difference frequency between the local oscillator signal received by the receiving end mixer and the local oscillator signal of the signal transmission link.
  • An embodiment of the present disclosure also provides a signal calibration link, comprising a signal transceiver link as described in any embodiment of the present disclosure, and a BIST module; the receiving antenna connection port of the signal receiving link is connected to the transmitting antenna connection port of the signal transmitting link through the BIST module, and the signal receiving link can be configured to calibrate the signal transmitting link.
  • the embodiment of the present disclosure also provides a signal calibration link, comprising two signal receiving links, a BIST module, an auxiliary circuit unit and a signal transmitting link as described in any embodiment of the present disclosure, and; any of the signal receiving links comprises a real mixer, a real analog-to-digital converter and a digital signal processing module; the real mixer can be configured to down-convert the received echo signal based on the received local oscillator signal to obtain an analog intermediate frequency signal, the real analog-to-digital converter can be configured to perform analog-to-digital conversion on the received intermediate frequency signal to obtain a digital intermediate frequency signal, and the digital signal processing module can be configured to process the digital intermediate frequency signal to obtain a target parameter; the echo signal is a signal formed by the signal transmitted by the signal transmitting link being reflected and/or scattered by the target object; the receiving antenna connection ports of the two signal receiving links are respectively connected to the transmitting antenna connection ports of the signal transmitting link through the auxiliary circuit unit and the BIST module in turn, and the signal receiving link can be configured to calibr
  • An embodiment of the present disclosure also provides a signal calibration link of a main signal transmission path, wherein the main signal transmission path is used to generate a radio frequency transmission signal after performing a compensation operation on the generated signal according to a compensation coefficient, so as to achieve target detection and/or communication, wherein: the signal calibration link is configured to obtain current observation information of the main signal transmission path under the current compensation coefficient; and when the current observation information meets the iteration condition, the current compensation coefficient is used as the compensation coefficient for the compensation operation of the signal transmission link; otherwise, the current compensation coefficient is iterated until the obtained observation information meets the iteration condition.
  • the compensation coefficient includes at least one of a harmonic distortion compensation parameter, a local oscillator leakage compensation parameter, and an orthogonal imbalance compensation parameter.
  • the signal transmission main path and the signal calibration link are integrated into the same integrated circuit.
  • the integrated circuit is a millimeter wave radar chip, and/or the radio frequency transmission signal is a FMCW signal.
  • the current compensation coefficient includes the initial compensation coefficient h(0), the first compensation coefficient h(1),..., the k-1th compensation coefficient h(k-1) and the k-th compensation coefficient h(k), in sequence, wherein the k-1th compensation coefficient h(k) is determined based on the k-1th compensation coefficient h(k-1) and the k-1th observation information O(k-1); wherein k is an integer greater than or equal to 2.
  • the kth compensation coefficient h(k) is determined based on the k-1th compensation coefficient h(k-1) and the k-1th observation information O(k-1); when the difference value between the initial observation information O(0) and the first observation information O(1) is less than the preset difference threshold, the initial observation information O(0) is adjusted according to a preset phase adjustment amount to obtain an adjusted first compensation coefficient h(1), and then new first observation information O(1) is obtained based on the adjusted first compensation coefficient h(1), and the kth compensation coefficient h(k) is determined based on the new first observation information O(1); wherein the difference value between the initial observation information O(0) and the new first observation information O(1) is greater than the difference threshold.
  • the method of determining the kth compensation coefficient h(k) based on the k-1th compensation coefficient h(k-1) and the k-1th observation information O(k-1) includes: when the difference between the absolute value of the initial observation information O(0) and the absolute value of the first observation information O(1) is greater than a preset difference threshold, based on updating the first compensation coefficient h(1) to the initial observation value O(0), iteratively calculating the sum between the k-1th compensation coefficient h(k-1) and the k-1 observation information O(k-1) to determine the kth compensation coefficient h(k); when the difference between the absolute value of the first observation information O(1) and the absolute value of the initial observation information O(0) is greater than a preset difference threshold, based on updating the first compensation coefficient h(1) to the inverse of the initial observation value O(0), iteratively calculating the difference between the k-1th compensation coefficient h(k-1) and the k-1 observation information O(k-1) to determine the kth compensation coefficient h(k)
  • determining the kth compensation coefficient h(k) based on the new first observation information O(1) includes: when the difference between the absolute value of the initial observation information O(0) and the absolute value of the new first observation information O(1) is greater than the difference threshold, iteratively calculating the sum of the k-1th observation information O(k-1) and the adjusted k-1th observation information O(k-1) obtained after the k-1th observation information O(k-1) is processed by the phase adjustment amount, and determining the kth compensation coefficient h(k); when the difference between the absolute value of the new first observation information O(1) and the absolute value of the initial observation information O(0) is greater than the difference threshold, updating the first compensation coefficient h(1) to the inverse of the adjusted first compensation coefficient h(1), iteratively calculating the difference between the k-1th compensation coefficient h(k-1) and the adjusted k-1th observation information O(k-1) obtained after the k-1th observation information O(k-1) is processed by the phase adjustment amount, and determining the kth compensation coefficient
  • the first compensation coefficient h(1) is determined based on initial observation information O(0).
  • the value of the initial compensation coefficient h(0) is 0.
  • the main signal transmission path includes a baseband module and a radio frequency module, wherein the baseband module is used to generate a baseband signal; the radio frequency module is used to frequency shift and/or phase shift the baseband signal using a transmitting end local oscillator signal to form a radio frequency transmission signal; wherein the observation information is obtained from the radio frequency module.
  • the RF module includes a transmitting mixer and a power amplifier; wherein the transmitting mixer uses a transmitting local oscillator signal to frequency shift and/or phase shift the baseband signal; the power amplifier is used to perform power amplification processing on the signal output by the transmitting mixer; wherein the observation information is obtained from at least one of the signal output end of the transmitting mixer, the signal output end of the power amplifier, and the signal output end of the voltage-to-current converter in the transmitting mixer.
  • the embodiment of the present disclosure also provides a signal calibration link for a main signal transmission path, wherein the main signal transmission path is used to generate a radio frequency transmission signal after compensating the generated signal according to a compensation coefficient, so as to achieve target detection and/or communication, wherein: the signal calibration link is configured to determine initial observation information O(0), first observation information O(1) and second observation information O(2) corresponding to the main signal transmission path under the conditions of different values of an initial compensation coefficient h(0), a first compensation coefficient h(1) and a second compensation coefficient h(2); and to determine a third compensation coefficient h(3) using the initial observation information O(0), the first observation information O(1) and the second observation information O(2) as the compensation coefficient used in the compensation operation of the signal transmission link.
  • the compensation coefficient includes at least one of a harmonic distortion compensation parameter, a local oscillator leakage compensation parameter, and an orthogonal imbalance compensation parameter.
  • the signal transmission main path and the signal calibration link are integrated into the same integrated circuit.
  • the integrated circuit is a millimeter wave radar chip, and/or the radio frequency transmission signal is a FMCW signal.
  • the method of using initial observation information O(0), first observation information O(1) and second observation information O(2) to determine the third compensation coefficient h(3) includes: normalizing the first observation information O(1) and the second observation information O(2) using the initial observation information O(0); and determining the third compensation coefficient h(3) using the result of the normalization process, with the third observation information O(3) being 0.
  • the method of determining the third compensation coefficient h(3) based on the condition that the third observation information O(3) is 0 and using the result of normalization processing includes: based on the condition that the third observation information O(3) is 0, determining the first coefficient x1 corresponding to the first compensation coefficient h(1) and the second coefficient x2 corresponding to the second compensation coefficient h(2) according to the first ratio d1 and the second ratio d2; wherein the first ratio d1 is the ratio between the first difference and the initial observation information O(0), and the first difference is the difference between the initial observation information O(0) and the first observation information O(1); the second ratio d2 is the ratio between the second difference and the initial observation information O(0), and the second difference is the difference between the initial observation information O(0) and the second observation information O(2); calculating the product between the first coefficient x1 and the first compensation coefficient h(1) to obtain a first multiplication result; and calculating the product between the second coefficient x2 and the second compensation coefficient h(2) to obtain a second multiplication result; and determining the third
  • determining the first coefficient x1 corresponding to the first compensation coefficient h(1) and the second coefficient x2 corresponding to the second compensation coefficient h(2) according to the first ratio d1 and the second ratio d2 includes: constructing an observation equation to determine the first coefficient x1 and the second coefficient x2 in the observation equation; wherein the calculation expression of the observation equation is the product of the inverse matrix of a 2*2 matrix and a first matrix of 1 row and 2 columns to obtain a second matrix of 1 row and 2 columns; wherein the first row of the 2*2 matrix records the numerical value of the real part and the numerical value of the imaginary part of the first ratio d1, and the second row records the numerical value of the real part and the numerical value of the imaginary part of the second ratio d2; the value of the first row in the first matrix is 1, and the value of the second row is 0; the first row of the second matrix is the value of the first coefficient x1, and the second row is the value of the second coefficient x2.
  • determining a third compensation coefficient h(3) according to the first multiplication result and the second multiplication result includes: calculating a sum of the first multiplication result and the second multiplication result to obtain the third compensation coefficient h(3).
  • the first compensation coefficient h(1) and the second compensation coefficient h(2) are determined according to initial observation information O(0).
  • the value of the initial compensation coefficient h(0) is 0.
  • the main signal transmission path includes a baseband module and a radio frequency module, wherein the baseband module is used to generate a baseband signal; the radio frequency module is used to frequency shift and/or phase shift the baseband signal using a transmitting end local oscillator signal to form a radio frequency transmission signal; wherein the observation information is obtained from the radio frequency module.
  • the RF module includes a transmitting mixer and a power amplifier; wherein the transmitting mixer uses a transmitting local oscillator signal to frequency shift and/or phase shift the baseband signal; the power amplifier is used to perform power amplification processing on the signal output by the transmitting mixer; wherein the observation information is obtained from at least one of the signal output end of the transmitting mixer, the signal output end of the power amplifier, and the signal output end of the voltage-to-current converter in the transmitting mixer.
  • An embodiment of the present disclosure also provides a signal compensation link, comprising a signal transmission link as described in any embodiment of the present disclosure, and a compensation unit, wherein the compensation unit can be configured to compensate for at least one of IQ mismatch, IQ imbalance, signal leakage, and harmonic distortion defects of the signal transmission link.
  • the compensation unit may be configured to compensate the signal transmission link based on calibration data obtained from the signal calibration link as described in any embodiment of the present disclosure.
  • the signal receiving link in the signal calibration link is integrated as an auxiliary calibration module in a vicinity of the transmission link to be calibrated, so as to perform real-time calibration operations during intervals when the transmission link is operating.
  • An embodiment of the present disclosure also provides a method for compensating for unequal feeder lengths, which is applied to an antenna array of an electromagnetic wave sensor having at least two signal links.
  • the method includes: taking the one with the shortest feeder line among the at least two signal links as a reference link, obtaining the delay difference of the remaining transmission links relative to the reference link; and compensating for the unequal feeder lengths of the antenna array in the digital domain based on the delay difference.
  • the unequal feeder lengths include unequal feeder lengths between different transmitting links or between different receiving links, between the local oscillator LO and the mixers in each transmitting channel, and/or unequal feeder lengths between the PA of each transmitting channel and its transmitting antenna.
  • An embodiment of the present disclosure also provides a signal calibration system, which is applied to an electromagnetic wave sensor.
  • the signal calibration system includes a signal transmission link and an auxiliary link.
  • the auxiliary link is integrated in the electromagnetic wave sensor adjacent to the signal transmission link.
  • the auxiliary link can be configured to perform real-time calibration on the signal transmission link.
  • the signal transmission link includes a signal transmission link and/or a signal receiving link;
  • the auxiliary link corresponds to the signal transmission link and includes an auxiliary receiving link and/or an auxiliary signal transmission link; wherein the auxiliary receiving link can be configured to calibrate the transmission link, and the auxiliary transmission link can be configured to calibrate the receiving link.
  • the signal receiving link includes a radio frequency part and a baseband part
  • the auxiliary transmission link corresponds to the signal receiving link and includes a radio frequency auxiliary transmission unit and a baseband auxiliary transmission unit
  • the radio frequency auxiliary transmission unit can be configured to calibrate the radio frequency part
  • the baseband auxiliary transmission unit can be configured to calibrate the baseband part
  • the RF auxiliary transmitting unit includes an IQ device; and wherein, when the signal receiving link is calibrated using the auxiliary transmitting link, the baseband part is first calibrated using the baseband auxiliary transmitting unit, and then the calibrated baseband part is used to calibrate the RF auxiliary transmitting unit, and then the calibrated RF auxiliary transmitting unit is used to calibrate the RF part.
  • the auxiliary receiving link includes an auxiliary receiving unit and a calibration receiving unit, wherein the auxiliary receiving unit includes an IQ device; wherein the calibration receiving unit may be configured to calibrate the auxiliary receiving unit, The calibrated auxiliary receiving unit may be configured to calibrate the transmission chain.
  • An embodiment of the present disclosure also provides an IQ mixer, which includes an I-branch mixing unit, a Q-branch mixing unit and a transformer unit, wherein the I-branch mixing unit can be configured to output an I-branch signal, the Q-branch mixing unit can be configured to output a Q-branch signal, and the transformer unit can be configured to magnetically couple the I-branch signal and the Q-branch signal to synthesize the IQ mixing output signal.
  • the transformer unit is a three-turn transformer structure, which includes two branch inductors and one magnetic coupling inductor; wherein, a branch inductor is connected in series between the output ends of the I branch mixing unit and the output ends of the Q branch mixing unit, respectively, and the magnetic coupling inductor is arranged between the two branch inductors.
  • the IQ mixer is applied to a digital phase shifter in a signal transmission link described in any embodiment of the present disclosure, and/or, to a signal calibration link described in any embodiment of the present disclosure, and/or, to a signal calibration system described in any embodiment of the present disclosure.
  • An embodiment of the present disclosure also provides an integrated circuit, comprising a radio frequency module, an analog signal processing module and a digital signal processing module connected in sequence; the radio frequency module is used to generate a radio frequency transmission signal and receive a radio frequency reception signal; the analog signal processing module is used to down-convert the radio frequency reception signal to obtain an intermediate frequency signal; the digital signal processing module is used to perform analog-to-digital conversion on the intermediate frequency signal to obtain a digital signal; and the radio frequency module includes the signal transmission link described in any embodiment of the present disclosure, the signal transceiver link described in any embodiment of the present disclosure, the signal calibration link described in any embodiment of the present disclosure, the signal compensation link described in any embodiment of the present disclosure, the signal calibration system described in any embodiment of the present disclosure, and/or the IQ mixer described in any embodiment of the present disclosure; and/or, the digital signal processing module performs compensation in the digital domain based on the feeder unequal length compensation method described in any embodiment of the present disclosure.
  • the integrated circuit further includes a data processing module, and the data processing module is used to process the digital signal to achieve target detection and/or wireless communication.
  • the integrated circuit is a chip such as millimeter wave or UWB.
  • the radio frequency receiving signal is an echo signal formed when the radio frequency transmitting signal is emitted and/or scattered by a target
  • the integrated circuit is a sensor chip.
  • the embodiments of the present disclosure further provide an electromagnetic wave sensor, comprising: a carrier; the integrated circuit described in any embodiment of the present disclosure, which is arranged on the carrier; an antenna, which is arranged on the carrier, or the antenna and the integrated circuit are integrated into an integral device and arranged on the carrier; wherein the integrated circuit is connected to the antenna for transmitting the radio frequency transmission signal and/or receiving the radio frequency reception signal.
  • An embodiment of the present disclosure also provides a device, comprising: a device body; and an electromagnetic wave sensor as described in any embodiment of the present disclosure, which is arranged on the device body; wherein the electromagnetic wave sensor is used for target detection and/or communication to provide reference information for the operation of the device body.
  • the embodiments of the present disclosure also provide a non-transitory computer-readable storage medium having computer-readable instructions stored thereon.
  • the processor executes the feeder unequal length compensation method described in any embodiment of the present disclosure.
  • FIG1A is a simplified schematic diagram of a signal transmission link of an analog phase shifter architecture
  • FIG1B is a simplified schematic diagram of an analog phase shifter in the signal transmission link shown in FIG1A ;
  • FIG2A is a simplified schematic diagram of a signal transmission link of a digital phase shifter architecture according to an embodiment of the present disclosure
  • FIG2B is a simplified schematic diagram of another signal transmission link of the digital phase shifter architecture in an embodiment of the present disclosure.
  • FIG2C is a waveform diagram of an FMCW transmission signal and an echo signal using sawtooth wave modulation
  • FIG3A is a schematic diagram of a signal model of a transmitter quadrature modulation when the quadrature is unbalanced
  • FIG3B is a schematic diagram of a signal generated by a real mixer
  • FIG3C is a schematic diagram of third-order harmonic distortion
  • FIG4A is a schematic diagram of the structure of a signal transmission link provided by an embodiment of the present disclosure.
  • FIG4B is a schematic diagram of quadrature imbalance compensation provided by an embodiment of the present disclosure.
  • FIG4C is a first schematic diagram of harmonic distortion compensation provided by an embodiment of the present disclosure.
  • FIG4D is a second schematic diagram of harmonic distortion compensation provided by an embodiment of the present disclosure.
  • FIG5A is a schematic diagram of a digital phase shifter architecture in a signal transmission link according to an embodiment of the present disclosure
  • FIG5B is a schematic diagram of a transmission link including a specific compensation unit in an embodiment of the present disclosure
  • FIG6A is a schematic diagram of the structure of a signal transceiver link provided by an embodiment of the present disclosure
  • FIG6B is a schematic diagram of the structure of another signal transceiver link provided by an exemplary embodiment of the present disclosure.
  • FIG7A is a schematic diagram of a transceiver link including a TX IQ Mod, a BIST IQ Mod and a RX Real Mixer in an embodiment of the present disclosure
  • FIG7B is a schematic diagram of a transceiver link including TX IQ Mod, RX IQ De-Mod and LO Freq Diff in an embodiment of the present disclosure
  • FIG8 is a schematic diagram of a transceiver link based on the structure shown in FIG7B combined with BIST in an embodiment of the present disclosure
  • FIG9 is a schematic diagram of a transceiver link including TX IQ Mod, BIST IQ Mod and RX IQ De-Mod in an embodiment of the present disclosure
  • FIG10 is a schematic diagram of a transceiver link including an auxiliary circuit and a BIST IQ Mod in an embodiment of the present disclosure
  • FIG11 is a schematic diagram of another transceiver link including an auxiliary circuit and a BIST IQ Mod in an embodiment of the present disclosure
  • FIG12A is a schematic diagram of the structure of a mixer in an embodiment of the present disclosure.
  • FIG12B is a schematic diagram of the structure of a compensation unit in a transmitter according to an embodiment of the present disclosure.
  • FIG13A is a schematic diagram of a digital pre-compensation HD3 architecture based on a cubic module in an embodiment of the present disclosure
  • FIG13B is a schematic diagram of a digital pre-compensation HD3 architecture based on a frequency doubling waveform generator module in an embodiment of the present disclosure
  • FIG13C is a schematic diagram of calibration compensation of a transmission link based on a digital phase shifter architecture in an embodiment of the present disclosure
  • FIG14A is a schematic diagram of a flow chart of a signal transmission method provided in an embodiment of the present disclosure.
  • FIG14B is a schematic diagram illustrating the principle of determining a compensation coefficient according to an embodiment of the present disclosure
  • FIG15 is a schematic diagram of a transmission link having at least two transmission channels in an embodiment of the present disclosure.
  • FIG16 is a schematic diagram of the structure of a digital LO signal generator in an embodiment of the present disclosure.
  • FIG17 is a schematic diagram of the structure of a transmission link including a feeder unequal length compensation module in an embodiment of the present disclosure
  • FIG18 is a schematic diagram of calibrating and compensating a transceiver link using an auxiliary circuit in an embodiment of the present disclosure
  • FIG19 is a schematic diagram of calibrating and compensating a receiving link using an auxiliary transmitting circuit in an embodiment of the present disclosure
  • FIG20 is a schematic diagram of calibrating and compensating a transmission link using an auxiliary receiving circuit in an embodiment of the present disclosure
  • FIG21 is a schematic diagram of the structure of an auxiliary circuit in an embodiment of the present disclosure.
  • FIG22 is a schematic diagram of the structure of another auxiliary circuit in an embodiment of the present disclosure.
  • FIG23 is a schematic diagram of a circuit module of an IQ Mixer in an embodiment of the present disclosure.
  • FIG24 is a schematic diagram of the structure of an IQ Mixer in an embodiment of the present disclosure.
  • FIG25 is a schematic diagram corresponding to the structure shown in FIG24;
  • FIG26 is a schematic diagram of the physical structure of another IQ Mixer in an embodiment of the present disclosure.
  • FIG27 is a schematic diagram of a flow chart of another signal transmission method provided in an embodiment of the present disclosure.
  • FIG28 is a schematic diagram of a flow chart of another signal transmission method provided in an embodiment of the present disclosure.
  • FIG29 is a flow chart of another signal transmission method provided in an embodiment of the present disclosure.
  • FIG. 1A is a simplified schematic diagram of a signal transmission link of an analog phase shifter architecture
  • FIG. 1B is a simplified schematic diagram of an analog phase shifter in the signal transmission link shown in FIG. 1A .
  • a signal generator 11 such as a phase-locked loop (PLL) generates an LO signal (such as a swept frequency signal in the 77 GHz band), which can be, for example, an FMCW signal;
  • an analog phase shifter (Analog PS) 12 performs a phase shift operation on the received LO signal, and then radiates it to a predetermined spatial area through a transmitting antenna 13 to perform operations such as target detection and measurement.
  • the corresponding analog phase shifter architecture may be as shown in FIG. 1B
  • the specific phase shifting principle may be as shown in the following formula:
  • phase shifter architecture can also be implemented by means of a delay line unit, that is, by using the narrow-band assumption of the signal to perform phase shifting by means of time delay.
  • delay line unit that is, by using the narrow-band assumption of the signal to perform phase shifting by means of time delay.
  • ⁇ is the delay time of the delay line.
  • phase adjustment precision and phase adjustment accuracy can be improved through calibration, the phase shifters of the analog architecture need to be calibrated off-line, which will greatly increase the difficulty and complexity of engineering implementation and product mass production. When working in different states (temperatures), the results of offline calibration may not be accurate.
  • analog phase shifters also have serious problems such as large area, large loss, stability and channel coupling. When multiple antenna phase shifters work together, the performance of the phase shifters will affect each other, further deteriorating the performance of the system.
  • the inventors of the present disclosure creatively proposed a signal transmission link of a digital phase shifter architecture to effectively improve the phase modulation precision and accuracy, while also avoiding the operation of non-online calibration of links and devices such as phase shifters in the transmission link, thereby reducing the complexity and difficulty of engineering implementation.
  • it can also effectively reduce the transmission link area and loss of the phase shifter architecture, improve the stability of the system, and reduce the channel coupling.
  • FIG2A is a simplified schematic diagram of a signal transmission link of a digital phase shifter architecture in an embodiment of the present disclosure
  • FIG2B is a simplified schematic diagram of another signal transmission link of a digital phase shifter architecture in an embodiment of the present disclosure
  • FIG2C is a waveform schematic diagram of an FMCW transmission signal and an echo signal using sawtooth wave modulation.
  • An embodiment of the present disclosure provides a signal transmission link that can be applied to an electromagnetic wave sensor.
  • the transmission link may include an analog signal source and a digital phase shifter.
  • the analog signal source may be configured to provide an initial analog signal (such as an LO signal), and the digital phase shifter may be configured to generate a phase-shifted signal in a digital domain.
  • the digital phase shifter may also shift the phase of the initial analog signal based on the generated phase-shifted signal to perform a preset phase shift operation on the initial analog signal.
  • a signal transmission link of a digital phase shifter architecture may include an analog signal source 21, a digital phase shifter (Digital PS) 22, and a transmitting antenna 23, etc., that is, the analog signal source 21 may be configured to provide an LO signal, and the digital phase shifter 22 may be configured to perform a preset phase shift operation on the received LO signal, so that the phase-shifted LO signal is radiated to a preset spatial region through the transmitting antenna 23.
  • the analog signal source 21 may also be an architecture including a phase-locked loop PLL, which may provide an electromagnetic wave (such as a laser, microwave, etc.) signal.
  • the analog signal source 21, the digital phase shifter 22, and the transmitting antenna 23 may be integrated into an integrated device, or may be separate components; for example, the analog signal source 21 and the digital phase shifter 22 may be integrated into a package body to form a SoC chip, etc., and the transmitting antenna 23 may be connected through the peripheral port of the chip, and formed on a carrier such as a PCB board.
  • the transmitting antenna 23 can also be integrated on the chip package to form AiP or AoP, having a chip structure with a packaged antenna.
  • the digital phase shifter 22 in the embodiment of the present disclosure may include a mixer 221, a digital-to-analog converter (i.e., DAC) 222, and a phase shift signal source (e.g., a digital baseband signal source Baseband) 223, etc., that is, the phase shift signal source 223 may be configured to provide a digital phase shift signal; the digital-to-analog converter 222 may be configured to perform analog-to-digital conversion on the received digital phase shift signal to convert the digital phase shift signal into an analog phase shift signal; the mixer 221 may be configured to perform a mixing operation on the received analog phase shift signal with the received electromagnetic wave signal from the analog signal source 21 to achieve a phase shift operation of setting the above-mentioned electromagnetic wave signal using the digital phase shift signal.
  • DAC digital-to-analog converter
  • a phase shift signal source e.g., a digital baseband signal source Baseband
  • a swept frequency electromagnetic wave signal can be provided based on the analog signal source 21, and/or a swept frequency digital phase shift signal can be provided based on the phase shift signal source 223, so that after mixing by the mixer 221, a swept frequency continuous wave signal is output.
  • the analog signal source 21 may be configured to provide an FMCW signal in a centimeter wave band or a millimeter wave band (such as 3.1 GHz, 24 GHz, 60 GHz, 77 GHz, etc.) in a microwave
  • the phase shift signal source 223 may be configured to provide a digital phase shift signal at the MHz level (for example, 3 MHz to 5 MHz, such as 3 MHz, 4 MHz, 5 MHz, etc.), that is, the digital-to-analog converter 222 performs digital-to-analog conversion on the MHz-level digital phase shift signal to obtain an analog phase shift signal in the corresponding frequency range
  • the mixer 221 may be configured to perform up-mixing or down-mixing operations on the received millimeter wave band FMCW signal based on the received analog phase shift signal of the fixed frequency band, so as to realize a preset phase shift operation on the FMCW signal.
  • the centimeter wave signal in the 3.1 GHz frequency band may include 3.1 GHz to 10.6 GHz, for example, 3.1 GHz, 5 GHz, 5 GHz, 6 GHz, 8 GHz, 10.6 GHz, etc.;
  • the millimeter wave signal in the 77 GHz frequency band may include 76 GHz to 81 GHz signals, for example, swept frequency signals such as 76 GHz to 77 GHz, 77 GHz to 79 GHz, 79 GHz to 81 GHz, or fixed frequency band signals such as 76 GHz, 77 GHz, 78 GHz, 79 GHz, 80 GHz, 81 GHz, etc.
  • the mixer 221 can be set as an IQ Mixer, and the digital-to-analog converter 222 is an IQ DAC.
  • the phase-shift signal source 223 can be configured to provide a digital baseband signal source (DDFS) for phase shifting and/or provide a corresponding source signal as a waveform controller (Waveform Control).
  • DDFS digital baseband signal source
  • Waveform Control waveform Control
  • an embodiment of the present disclosure provides a signal transmission link, which is applied to a radar system.
  • the signal transmission link includes: a transmitting baseband digital module 201 (including the aforementioned phase-shifted signal source 223), a digital-to-analog converter (DAC) module 202, a transmitting local oscillator 203, and a transmitting orthogonal modulator 204.
  • the DAC module 202 includes two identical digital-to-analog converters, wherein: the transmitting baseband digital module 201 is configured to generate two orthogonal transmitting digital baseband signals.
  • the two orthogonal transmitting digital baseband signals are respectively sent to a digital-to-analog conversion module 202; the digital-to-analog conversion module 202 is configured to convert the two orthogonal transmitting digital baseband signals into two transmitting analog baseband signals (i.e. analog phase-shifted signals); the transmitting local oscillator 203 is configured to provide a transmitting local oscillator signal TX_LO (i.e. initial analog signal); the transmitting orthogonal modulator 204 is configured to perform frequency shifting and phase shifting on the transmitting local oscillator signal TX_LO based on the two transmitting analog baseband signals, so as to form a predetermined phase-shifted FMCW RF transmitting signal.
  • TX_LO i.e. initial analog signal
  • the transmitting orthogonal modulator 204 is configured to perform frequency shifting and phase shifting on the transmitting local oscillator signal TX_LO based on the two transmitting analog baseband signals, so as to form a predetermined phase-shifted FMCW RF transmitting signal.
  • the transmitting digital baseband signal provided by the transmitting baseband digital module 201 includes preset phase information; the digital-to-analog conversion module 202 performs digital-to-analog conversion on the received transmitting digital baseband signal to convert the transmitting digital baseband signal into a transmitting analog baseband signal (such as without changing the phase information); the transmitting orthogonal modulator 204 mixes the received transmitting analog baseband signal with the transmitting local oscillator signal TX_LO generated by the transmitting local oscillator 203 to achieve frequency shifting of the transmitting local oscillator signal based on the transmitting analog baseband signal while performing a preset phase shift operation to form a FMCW RF transmission signal after a predetermined phase shift.
  • the signal transmission link of the embodiment of the present disclosure forms a digital phase shifter architecture by using a transmitting baseband digital module 201, a digital-to-analog conversion module 202 and a transmitting orthogonal modulator 204. Since the baseband signal of the architecture is generated in the digital domain, it has better orthogonality and lower side lobes, so its phase shift phase can be generated very accurately, making the phase modulation accuracy higher, thereby realizing a vehicle-mounted radar system with a high-precision digital phase shifting function, reducing the isolation requirements between antennas, and having the advantages of small link loss, low cost, and no need for offline calibration, and can support more flexible wave transmission schemes, such as high-performance Doppler division multiplexing and frequency division multiplexing, and can support frequency response compensation in the digital domain.
  • the transmitting baseband digital module 201 provides a digital signal
  • the transmitting modulator is set as an orthogonal modulator (IQ Modulator) and the digital-to-analog conversion module 202 is set as an orthogonal digital-to-analog converter (IQ DAC).
  • IQ Modulator orthogonal modulator
  • IQ DAC orthogonal digital-to-analog converter
  • the transmitting local oscillator 203 may include a phase-locked loop (PLL) structure, and may provide an electromagnetic wave (eg, laser, microwave, etc.) signal.
  • PLL phase-locked loop
  • the signal transmission link also includes: a power amplifier (PA) 205, wherein: the power amplifier 205 is configured to amplify the power of the phase-shifted RF signal and output the amplified signal to the transmitting antenna.
  • PA power amplifier
  • the signal transmission link further includes: a transmitting antenna 206, wherein: the transmitting antenna 206 is configured to radiate the amplified signal to a preset spatial area.
  • the signal amplified by the power amplifier 205 can be radiated to a preset spatial region through the packaged integrated or external transmitting antenna 206. That is, the transmitting local oscillator 203, the digital phase shifter and the transmitting antenna 206 can be integrated into a device or can be separate components; for example, the transmitting local oscillator 203 and the digital phase shifter can be integrated into a package to form a SoC chip, while the transmitting antenna 206 can be connected through the peripheral port of the chip and formed on a carrier such as a PCB board. At the same time, in some optional embodiments, the transmitting antenna 206 can also be integrated into the chip package to form AiP or AoP, with a chip structure with a packaged antenna.
  • the bandwidth of the frequency sweep signal is above 2 GHz.
  • the electromagnetic wave of the transmission signal emitted by the transmitting antenna of the frequency modulated continuous wave radar system is a high-frequency frequency modulated continuous wave
  • the echo signal received by the receiving antenna of the frequency modulated continuous wave radar system is the electromagnetic wave reflected/scattered back by the target object.
  • FIG2C shows a waveform diagram of an exemplary FMCW transmission signal and an echo signal. As shown in FIG2C , the frequencies of the transmission signal and the echo signal change regularly over time.
  • the frequency modulated continuous wave is generally sawtooth-shaped, triangular, etc.
  • the present disclosure takes the sawtooth shape as an example for explanation.
  • the electromagnetic wave within each frequency modulation period T is called a chirp, and the frequency of each Chirp signal increases linearly with time.
  • the bandwidth range B of a chirp is greater than or equal to 2 GHz.
  • the transmitting end digital baseband signal is a single-tone signal
  • the transmitting end local oscillator signal is a swept frequency signal.
  • the transmitting end local oscillator 203 may be configured to provide a FMCW signal of a centimeter wave frequency band or a millimeter wave frequency band (such as 3.1 GHz, 24 GHz, 60 GHz, 77 GHz, etc.) in a microwave
  • the transmitting end baseband digital module 201 may be configured to provide a single-tone transmitting end digital baseband signal of a MHz level (for example, 3 MHz to 5 MHz, such as 3 MHz, 4 MHz, 5 MHz, etc.), that is, the digital-to-analog conversion module 202 performs digital-to-analog conversion on the MHz-level single-tone transmitting end digital baseband signal to obtain a single-tone transmitting end analog baseband signal of a corresponding frequency range
  • the transmitting end orthogonal modulator 204 may be configured to
  • the FMCW signal in the 3.1 GHz frequency band may include a swept frequency signal between 3.1 GHz and 10.6 GHz, such as 7.163-8.812 GHz;
  • the FMCW signal in the 77 GHz frequency band may include a swept frequency signal between 76 GHz and 81 GHz, or swept frequency signals such as 76 GHz to 77 GHz, 77 GHz to 79 GHz, or 79 GHz to 81 GHz.
  • the digital baseband signal at the transmitting end may be a swept frequency signal
  • the local oscillator signal at the transmitting end may be a single tone signal
  • the transmitting end local oscillator 203 may be configured to provide a single-tone transmitting end local oscillator signal in a centimeter wave band or a millimeter wave band (such as 3.1 GHz, 24 GHz, 60 GHz, 77 GHz, etc.) in microwaves
  • the transmitting end baseband digital module 201 may be configured to provide a MHz (for example, 3 MHz to 5 MHz, such as 3 MHz, 4 MHz, 5 MHz, etc.) level transmitting end digital baseband FMCW signal, that is, the digital-to-analog conversion module 202 performs digital-to-analog conversion on the MHz level transmitting end digital baseband FMCW signal to obtain a transmitting end analog baseband FMCW signal in a corresponding frequency range
  • the transmitting end orthogonal modulator 204 may be configured to perform up-mixing or down-mixing operations on the received single-tone transmitting end local oscillator signal in the centimeter wave band or the millimeter wave band based on
  • the single-tone transmitting local oscillator signal in the 3.1 GHz frequency band can be a single-tone analog signal in a fixed frequency band such as 3.1 GHz, 5 GHz, 6 GHz, 8 GHz, 10.6 GHz, etc.;
  • the single-tone transmitting local oscillator signal in the 77 GHz frequency band can be a single-tone analog signal in a fixed frequency band such as 76 GHz, 77 GHz, 78 GHz, 79 GHz, 80 GHz, 81 GHz, etc.
  • the signal transmission link also includes: a low pass filter (LPF) 207, which is arranged between the digital-to-analog conversion module 202 and the transmitting end orthogonal modulator 204, and is configured to perform low pass filtering on the transmitting end analog baseband signal output by the digital-to-analog conversion module 202 and output it to the transmitting end orthogonal modulator 204.
  • LPF low pass filter
  • the transmitting end baseband digital module 201 generates two orthogonal digital baseband signals, namely, an I digital baseband signal and a Q digital baseband signal, and sends the generated digital baseband signals to the digital-to-analog conversion module 202 (including two identical DACs, namely, IQ DACs) to obtain two analog baseband signals.
  • the two analog baseband signals are then input into the low-pass filter 207 to filter out the out-of-band noise signals, and are orthogonally modulated by the transmitting end orthogonal modulator 204 to obtain the modulated RF signal, which is then radiated out through the power amplifier 205 and the transmitting antenna 206.
  • the signal transmission link may further include a direct digital frequency synthesizer (DDFS) (not shown in FIG. 2B ), which is disposed between the transmitting baseband digital module 201 and the digital-to-analog conversion module 202.
  • the direct digital frequency synthesizer may be configured to implement at least one of a variety of signal waveforms and transmission modes such as CDM (Code-Division Multiplexing), DDM (Doppler Division Multiplexing), TDM (Time-Division Multiplexing), SDM (Space Division Multiplexing), CSD (Circuit Switch Data), and Digital IF (Digital Intermediate Frequency) based on the received source signal, so as to achieve flexible configuration of the signal transmission form and the transmission waveform.
  • CDM Code-Division Multiplexing
  • DDM Doppler Division Multiplexing
  • TDM Time-Division Multiplexing
  • SDM Space Division Multiplexing
  • CSD Circuit Switch Data
  • Digital IF Digital Intermediate Frequency
  • FIG3A is a schematic diagram of a signal model of quadrature modulation of a transmitter when the quadrature is unbalanced.
  • DSP is a digital signal processor
  • DAC is a digital-to-analog converter
  • LPF is a low-pass filter
  • LO is a local oscillator signal source
  • a I (f) is a frequency domain expression of the output signal of the LPF of channel I
  • E I (f) is a baseband frequency response of channel I
  • A′ I (f) is a frequency domain expression of the baseband output signal of channel I
  • S I (f) is a frequency domain expression of the output signal of the modulator of channel I
  • G I (f) is a radio frequency frequency response of channel I
  • S′ I (f) is a frequency domain expression of the radio frequency output signal of channel I
  • ⁇ I is the amplitude gain of the mixer of channel I, is the phase response of the I channel mixer
  • a Q (f) is the frequency domain expression of the Q
  • the output U RF (f) of the quadrature demodulator can be expressed as:
  • A(ff m )V 1 (ff m ) is the desired signal near frequency f m ;
  • a * (-ff m ) V1 * (-ff m ) is the desired signal near the frequency -f m ;
  • a * (-f+ fm ) V2 ( ffm ) is the image signal near the frequency fm ;
  • A(f+f m )V 2 * (-ff m ) is the image signal near the frequency -f m ;
  • the baseband equivalent model of the quadrature regulator can be: Where A(f)V 1 (f) is the desired signal near frequency f m ; A * (-f)V 2 (f) is the image signal near frequency -f m .
  • the relative image ratio (RIR) is used to measure the ability of the quadrature mixer to suppress unwanted image frequency components. Therefore, the definition of RIR is the ratio of the power of the desired signal and the image signal components, so it can be written as The above equations show that the cascaded gain and phase response of the I and Q channels need to be equal (the cascaded gain and phase response of each channel should be considered), otherwise the channels are unbalanced and the image components are not completely suppressed.
  • FIG3B is a schematic diagram of signals generated by a real mixer.
  • F IN represents the input signal of the mixer
  • F OUT represents the output signal of the mixer
  • F LO represents the local oscillator signal used by the mixer
  • the input signal of the mixer is a baseband signal with a frequency of F BB
  • the local oscillator signal used by the mixer is a high-frequency signal with a frequency of F LO
  • the output signal of the mixer is a signal with a frequency of F LO ⁇ F BB .
  • An ideal mixer will produce an output that is the product of its two inputs. In frequency, the output should be F LO +F BB and F LO -F BB , with nothing else. If either input is not driven, there will be no output. However, a real mixer will also produce some energy at F BB and F LO . The energy produced at F BB can be ignored because it is far from the desired output and will be filtered out by the RF devices after the mixer output. Regardless of the energy produced at F BB , the energy produced at F LO can be a problem. It is very close to or within the desired output signal and is difficult or impossible to remove by filtering, which will also filter out the desired signal. There are other ways that the LO can leak to the system output, such as through the power supply or across the silicon itself. Regardless of how the local oscillator leaks, its leakage can be referred to as LO Leakage (LOL).
  • LLO Leakage LLO Leakage
  • FIG3C is a schematic diagram of third-order harmonic distortion.
  • the baseband signal generated by the baseband processor is a baseband signal with a frequency of BB, and after being processed by the IQ modulator, a desired signal Wanted, an image signal IMG, and a third-order harmonic HD3 can be obtained, and the signal output by the power amplifier PA includes the desired signal Wanted, the image signal IMG, and the third-order harmonic HD3.
  • the main source of HD3 is the third-order nonlinearity of the baseband analog device.
  • the third-order harmonic component HD3 of the baseband BB signal is located at -3BB.
  • the baseband HD3 is up-converted to LO-3BB, that is, C-IM3.
  • the signal generated by the modulation is difficult to distinguish at high frequencies.
  • the present disclosure uses the addition of a digital pre-compensation module at the transmitting end to solve the problems introduced by the non-ideal characteristics of analog devices in the digital orthogonal modulation radar TX.
  • the present disclosure not only includes a pre-compensation module, but also a calibration loop to achieve real-time calibration and compensation.
  • FIG4A is a schematic diagram of the structure of a signal transmission link provided by an embodiment of the present disclosure.
  • the signal transmission link includes a signal transmission main path and a signal calibration link integrated in the same integrated circuit; wherein: the signal calibration link is configured to calibrate the signal transmission main path to obtain compensation information; and the signal transmission main path is configured to generate a radio frequency transmission signal after performing a compensation operation according to the compensation information, so as to achieve target detection and/or communication.
  • the signal calibration link can perform calibration operations on the main transmission path in real time, and the calibration operation of the calibration link can be unaffected by changes in the operating environment of the main transmission path, so that the main transmission path can obtain more accurate calibration information, thereby improving the signal processing performance of the main transmission path.
  • the compensation information includes at least one of a harmonic distortion compensation parameter, a local oscillator leakage compensation parameter, and an orthogonal imbalance compensation parameter. Since the compensation information generated by the signal calibration link can solve the orthogonal imbalance, LO leakage, and harmonic distortion problems, the signal quality in the main transmission path of the digital phase shifter architecture is effectively improved.
  • the integrated circuit is a millimeter wave radar chip or an ultra-wideband (Ultra Wide Band, UWB) chip, etc., and/or the RF transmission signal is an FMCW signal.
  • UWB Ultra Wide Band
  • the main signal transmission path may include a first signal source and a phase shifter; wherein the first signal source is configured to generate a first analog signal; and the phase shifter is configured to perform frequency shifting and/or phase shifting on the first analog signal to form a radio frequency transmission signal.
  • the phase shifter When the phase shifter is a non-orthogonal architecture, the phase shifter includes a second signal source and a transmitting mixer, wherein the second signal source is configured to generate a second analog signal, and the transmitting mixer is configured to mix the first analog signal and the second analog signal to form the RF transmission signal; when the phase shifter is an orthogonal architecture, the phase shifter includes: a second signal source, a digital-to-analog conversion module and a transmitting mixer; wherein the second signal source is configured to generate a first digital signal; the digital-to-analog conversion module is configured to convert the first digital signal into a second analog signal; and the transmitting mixer is configured to frequency shift and/or phase shift the first analog signal based on the second analog signal to form the RF transmission signal.
  • the main transmitting path also includes a compensation circuit, wherein the signal input end of the compensation circuit is connected to the second signal source, and the signal input end is connected to the phase shifter, and the compensation circuit is used to combine the compensation signal and the signal output by the second signal source before outputting it.
  • the compensation signal used by the compensation circuit is two orthogonal signals; when the signal output by the main signal transmission path is not two orthogonal signals, the compensation signal used by the compensation circuit is a signal of the same type as the signal output by the second signal source, wherein the signal type is a digital signal or an analog signal.
  • the compensation circuit includes a compensation signal generator and an adder, the compensation signal generator can be configured to generate the compensation signal; the adder is connected to the compensation signal generator and the second signal source, and is used to perform a signal superposition operation on the signal output by the second signal source and the compensation signal output by the compensation signal generator.
  • the compensation signal generator when the signal output by the main signal transmission path is a two-way orthogonal signal, the compensation signal generator includes at least one of a harmonic compensation signal unit, an orthogonal imbalance compensation signal unit and a local oscillator leakage compensation signal unit; when the signal output by the main signal transmission path is not a two-way orthogonal signal, the compensation circuit includes at least one of a harmonic compensation signal unit and a local oscillator leakage compensation signal unit; wherein the compensation signal generated by the harmonic compensation signal unit is used to eliminate the harmonic signal of the main frequency signal in the main signal transmission path; the compensation signal of the local oscillator leakage compensation signal unit is used to compensate for the leakage signal generated by the transmitting end local oscillator signal in the main signal transmission path; and the compensation signal generated by the orthogonal imbalance compensation signal unit is used to compensate for the mirror signal of the main frequency signal in the main signal transmission path.
  • the compensation signal used by the orthogonal imbalance compensation circuit has the same frequency, the same amplitude, and the opposite phase between the image signal corresponding to the signal transmission main path and the image signal corresponding to the desired signal generated by the signal transmission main path.
  • the compensation signal used by the quadrature imbalance compensation circuit is determined based on a signal output by a second signal source and a complex conjugate signal of a frequency inversion of the signal output by the second signal source.
  • the compensation signal generated by the quadrature imbalance compensation circuit is obtained by: obtaining the product of a preset pre-compensation coefficient and the complex conjugate signal to obtain an adjustment signal corresponding to the complex conjugate signal; and calculating the difference between the signal output by the second signal source and the adjustment signal to obtain the compensation signal generated by the quadrature imbalance compensation circuit.
  • the pre-compensation coefficient is determined based on the ratio of the amplitude of the desired signal to the amplitude of the mirror signal corresponding to the desired signal.
  • FIG4B is a schematic diagram of quadrature imbalance compensation provided by an embodiment of the present disclosure.
  • a pre-compensation method may be used.
  • Let h IQ be the pre-compensation coefficient, acting on the complex conjugate signal A * (-f) of the baseband signal to be transmitted.
  • the frequency domain form of the modulator output can be expressed as:
  • the precompensation coefficient represents the frequency response of the optimal precompensation filter, which will be used in the digital domain of the modulator to predistort (precompensate) the I and Q signals before transmission. It can be seen that only the ratio between V 2 ( -ft ) and V 1 ( ft ) needs to be estimated.
  • the compensation signal generated by the local oscillator leakage compensation signal unit is generated based on the leakage signal corresponding to the first analog signal used by the phase shifter.
  • the compensation signal generated by the local oscillator leakage compensation signal unit has the same frequency and amplitude as the leakage signal but opposite phase.
  • LO Leakage Correction is achieved by generating a signal with equal amplitude but opposite phase to LOL, thereby canceling it.
  • the cancellation signal can be generated by applying a DC offset to the input of the transmitter.
  • the orthogonal mixer structure enables it to generate cancellation signals well. Since there are orthogonal signals of LO in the mixer, signals of any phase and amplitude can be generated at the frequency of LO. It can be geometrically interpreted in the complex plane. The first line is the result of the synthesis of the two IQ channels in the time domain, and the second line is geometrically interpreted from the perspective of the spectrum of the two IQ channels. After applying a DC bias to the transmit signal, the output of the mixer will contain the desired transmit signal as well as the desired LOL cancellation signal. The deliberately generated cancellation signal LOC will cancel the useless LOL, leaving only the desired transmit signal.
  • Tx LOC h DC
  • LOL can be modeled as follows:
  • the ideal IQs are ⁇ I cos(w m t) and ⁇ Q sin(w m t); the actual IQs are ⁇ I cos(w m t+ ⁇ I ) and ⁇ Q sin(w m t+ ⁇ Q ); the LO leakage is Therefore, LO Leakage needs to be eliminated in the following ways:
  • LOL can be calibrated by adding different DC offsets to the I and Q signals.
  • the compensation signal generated by the harmonic compensation signal unit has the same frequency and amplitude as the harmonic signal in the signal transmission main path, and has an opposite phase.
  • the harmonic compensation signal unit includes an n-th power module or an n-fold frequency signal generator; wherein: the n-th power module, wherein the signal input end of the n-th power module is connected to the signal output end of the second signal source, is used to generate a signal with a frequency n times the frequency of the signal output by the second signal source, and obtain the compensation signal; the n-fold frequency signal generator is used to generate a signal with a frequency n times the frequency of the signal output by the second signal source, and obtain the compensation signal, wherein the value of n is a positive integer. wherein the value of n is an odd number. further, the value of n is 3.
  • FIGS. 4C and 4D are schematic diagrams of harmonic distortion compensation provided by an embodiment of the present disclosure.
  • FIG4C and FIG4D similar to the principle of eliminating LOL, there are two compensation methods for third-order harmonic distortion, namely, a compensation architecture based on an n-th power module and a compensation architecture based on an n-fold frequency waveform generator.
  • a pc (n) a(n)-h HD3 ⁇ ( ⁇ * (n)) 3
  • a pc (n) a(n)-h HD3 ⁇ a 3 (n)
  • FIG5A is a schematic diagram of a digital phase shifter architecture in a signal transmission link based on the one shown in FIG2A;
  • FIG5B is a schematic diagram of a transmission link including a specific compensation unit in an embodiment of the present disclosure.
  • a signal transmission link (TX digital phase shifter architecture) of a digital phase shifter architecture may include a digital baseband signal source (Baseband), a direct digital frequency synthesizer (Direct Digital Frequency Synthesizer, referred to as DDFS), an IQ digital-to-analog converter (Digital to Analog Convertor, referred to as DAC), a low-pass filter (Low-Pass Filter, referred to as LPF), an IQ modulator (IQ modulator/IQ Mixer), a power amplifier (Power Amplifier, PA), etc., that is, the baseband signal source is configured to provide a digital phase shift source signal, and the direct digital frequency synthesizer (DDFS) may include a digital baseband signal source (Base) (Direct
  • the synthesizer can be configured to implement at least one of a variety of signal waveforms and wave transmission methods such as CDM (Code-Division Multiplexing), DDM (Doppler Division Multiplexing), TDM (Time-Division Multiplexing), SDM (Space Division Multiplexing), CSD (Circuit Switch Data), Digital IF (Digital Intermediate Frequency) based on the received source signal, so as to realize flexible configuration of signal transmission form and transmission waveform.
  • CDM Code-Division Multiplexing
  • DDM Doppler Division Multiplexing
  • TDM Time-Division Multiplexing
  • SDM Space Division Multiplexing
  • CSD Circuit Switch Data
  • Digital IF Digital Intermediate Frequency
  • the digital phase shifter architecture is configured to generate a baseband signal sequence in the digital domain, and can generate an analog baseband signal through a DAC, and then modulate the transmission signal to a high frequency through an orthogonal mixer, that is, because the baseband signal of the architecture is generated in the digital domain, it has better orthogonality and lower sidelobes, and therefore its phase shift phase can be generated very accurately, resulting in higher phase modulation accuracy.
  • a compensation unit can be added to the signal transmission link to address the TX IQ imbalance (Imbalance), signal leakage (such as TX LO Leakage) and harmonic distortion (HD) caused by IQ mismatch.
  • TX compensation can be set between the TX DDFS and the IQ DAC to perform calibration and compensation operations on the signal transmission link of the digital phase shifter architecture to achieve an operation to solve at least one of the above problems.
  • the HD caused by the third-order nonlinearity of the baseband can be referred to as HD3.
  • the TX compensation unit may include at least one of an LO leakage compensation unit (TX LO leakage compensation), an IQ imbalance compensation unit (TX IQ Imbalance compensation) and an HD3 compensation unit (TX HD3 compensation), wherein the LO leakage compensation unit may be configured to be used for compensation for signal leakage, the IQ imbalance compensation unit may be configured to be used for compensation for IQ imbalance, and the HD3 compensation unit may be configured to be used for compensation for the above-mentioned HD3.
  • the IQ imbalance compensation unit is configured to be used for compensation for at least one of an IQ modulator imbalance (TX IQ Modulator imbalance) and an IQ channel imbalance (IQ channel imbalance).
  • the TX compensation unit includes at least two of the LO leakage compensation unit, the IQ imbalance compensation unit and the HD3 compensation unit
  • compensation can be performed synchronously (such as in parallel) or sequentially (such as in series) according to actual needs and signal characteristics.
  • IQ imbalance compensation can be performed first, then LO leakage compensation, and finally HD3 compensation.
  • the signal transmission link of the digital phase shifter architecture may also include an error correction module for DAC (TX DAC Board Error Correction) and an AWGN (additive white gaussian noise) module for Gaussian white noise, etc., which are not shown in the figure and can be added or deleted according to actual needs.
  • I in IQ mentioned in the embodiments of the present disclosure can be expressed as the abbreviation of In-Phase (i.e., in-phase)
  • Q can be expressed as the abbreviation of Quadrature (i.e., orthogonal)
  • RF can be expressed as the abbreviation of Radio Frequency (i.e., radio frequency).
  • the embodiment of the present disclosure also provides a signal transceiver link, including a signal transmission link and a signal receiving link, as shown in Figure 6A or Figure 6B, the signal transmission link may include: a transmitting baseband digital module 201, a digital-to-analog conversion module 202, a transmitting local oscillator 203 and a transmitting orthogonal modulator 204, wherein: the transmitting baseband digital module 201 is configured to generate two orthogonal transmitting digital baseband signals, and send the generated transmitting digital baseband signals to the digital-to-analog conversion module 202; the digital-to-analog conversion module 202 is configured to convert the transmitting digital baseband signal into a transmitting analog baseband signal; the transmitting local oscillator 203 is configured to provide a transmitting local oscillator signal TX_LO; the transmitting orthogonal modulator 204 is configured to perform a phase shift operation on the transmitting local oscillator signal TX_LO based on the transmitting analog baseband signal to obtain a phase
  • the signal receiving link may include a receiving end local oscillator 302, a receiving end mixer 303, an analog-to-digital converter (ADC) 304 and a receiving end baseband digital module 305; wherein the receiving end local oscillator 302 is configured to provide a receiving end local oscillator signal; the receiving end mixer 303 is configured to perform a mixing operation on the received echo signal based on the receiving end local oscillator signal to obtain a receiving end analog baseband signal; the analog-to-digital converter 304 is configured to convert the receiving end analog baseband signal into a receiving end digital baseband signal; the receiving end baseband digital module 305 is configured to process the receiving end digital baseband signal to achieve target detection and/or wireless communication, for example, to obtain parameter information of the target such as distance, speed, angle, height and micro-motion characteristics.
  • ADC analog-to-digital converter
  • two ideal I-channel digital baseband signals and Q-channel digital baseband signals generated by the transmitting end baseband digital module 201 can obtain a very ideal complex signal after passing through the digital-to-analog conversion module 202, and the phase of the complex signal can be accurately controlled by the transmitting end baseband digital module 201.
  • the phase information of the radio frequency signal of the signal transmission link can be effectively obtained, so that phase modulation of multiple antennas can be realized.
  • the signal transmission link may further include: a power amplifier 205, wherein the power amplifier 205 is configured to amplify the power of the phase-shifted radio frequency signal and output the amplified signal to the transmission antenna.
  • the signal transmission link may further include: a transmission antenna 206, wherein the transmission antenna 206 is configured to radiate the amplified signal to a preset spatial area.
  • the signal receiving link may further include a receiving antenna 301, wherein the receiving antenna 301 is configured to receive an echo signal, where the echo signal is a signal formed when the signal transmitted by the signal transmitting link is reflected and/or scattered by a target object.
  • the local oscillator signal at the receiving end may be a swept frequency signal, or the local oscillator signal at the receiving end may be a single tone signal.
  • the frequency of the TX-LO signal received by the transmitting end orthogonal modulator 204 in the signal transmission link and the frequency of the RX-LO signal received by the receiving end mixer 303 in the signal receiving link may be the same.
  • the signal output by the transmitting end baseband digital module 201 is a sine wave of x MHz
  • the TX-LO signal and the RX-LO signal may both be sine waves of z GHz, wherein x and z are both positive numbers, generally between 0 and 1000.
  • the signal transmission link can have two wave transmission schemes: 1) the transmitting end local oscillator signal is swept, and the transmitting end digital baseband signal is single-tone; 2) the transmitting end local oscillator signal is single-tone, and the transmitting end digital baseband signal is swept.
  • the transmitting end local oscillator signal TX_LO, the transmitting end digital baseband signal, and the modulated transmission signal are represented by TLO(t), BB(t) and TX(t), respectively, and the subscripts I and q are used to represent the I-path signal and the Q-path signal, and the superscript a is used to represent its complex signal form, then under the two wave transmission schemes, the signals at each stage of the signal transmission link can be represented as follows:
  • fbb is the starting frequency of the digital baseband signal at the transmitting end
  • ftlo is the starting frequency of the local oscillator signal at the transmitting end
  • RLO(t) the receiving end local oscillator signal RX_LO in the signal receiving link shown in FIG. 6A.
  • RLO(t) the receiving end local oscillator signal
  • f rlo is the starting frequency of the local oscillator signal at the receiving end
  • ⁇ 0 is the initial phase of the local oscillator signal at the receiving end.
  • the receiving antenna 301 can be connected through the peripheral port of the chip and formed on a carrier such as a PCB board.
  • the receiving antenna can also be integrated on the chip package to form AiP or AoP, that is, a chip structure with a packaged antenna.
  • the signal receiving link may further include a low noise amplifier (Low Noise Amplifier, LNA) 306, which is disposed between the receiving antenna 301 and the receiving end mixer 303, and performs low noise amplification on the echo signal received by the receiving antenna 301 before sending it to the receiving end mixer 303.
  • LNA Low Noise Amplifier
  • the signal receiving link may further include a low pass filter (LPF) 307 and a high pass filter (HPF) 308 connected in series, which are arranged between the receiving mixer 303 and the analog-to-digital converter 304.
  • the low pass filter 307 and the high pass filter 308 constitute a bandpass filter for filtering out-of-band noise.
  • the receiving mixer 303 may be a real mixer
  • the analog-to-digital converter 304 may be a real analog-to-digital converter.
  • the signal transmission link adopts a digital phase-shifting architecture
  • the signal receiving link may include a receiver of an orthogonal receiving architecture or a non-orthogonal receiving architecture. Therefore, it can effectively be compatible with sensors of receiving links of various architectures, effectively reducing the development cost of the entire transceiver link system.
  • the receiving mixer 303 may be an orthogonal mixer
  • the analog-to-digital converter 304 may be an orthogonal analog-to-digital converter.
  • the embodiment of the present disclosure adjusts the receiving mixer 303 in the signal receiving link to an IQ demodulator (IQ Demodulator), and adjusts the analog-to-digital converter 304 to an IQ ADC.
  • the echo signal received by the receiving antenna is processed by the above-mentioned low-noise amplifier 306, receiving mixer 303, low-pass filter 307, high-pass filter 308 and analog-to-digital converter 304 in sequence, and then converted into an IQ digital baseband signal.
  • the subsequent receiving baseband digital module 305 processes the IQ digital baseband signal to obtain parameter information of the target such as distance, speed, angle, altitude and micro-motion characteristics (i.e., micro-Doppler).
  • the receiving end local oscillator signal RX_LO can be expressed as:
  • the local oscillator signal at the receiving end of the signal receiving link may be a single tone signal as shown in formula (12) or formula (13) in addition to being a frequency sweep signal as shown in formula (9) or formula (11).
  • the receiving end mixer 303 is an orthogonal mixer:
  • the local oscillator signal at the receiving end is a single-tone signal
  • the embodiments of the present disclosure can expand various system-level technical solutions according to the combination of different transmission schemes and receiving schemes (for example, whether the transmitting end uses a digital baseband signal single tone, a local oscillator signal sweep frequency or a digital baseband signal sweep frequency, a local oscillator signal single tone; whether the receiving end uses a real mixer, a real analog-to-digital converter or an orthogonal mixer, an orthogonal analog-to-digital converter; whether the receiving end uses a single-tone local oscillator signal or a swept frequency local oscillator signal).
  • Figure 7A is a schematic diagram of a transceiver link including TX IQ modulation (Mod) and RX real demodulation (Real De-Mod) in an embodiment of the present disclosure
  • Figure 7B is a schematic diagram of a transceiver link including TX IQ Mod and RX IQ De-Mod in an embodiment of the present disclosure
  • Figure 8 is a schematic diagram of a transceiver link based on the structure shown in Figure 7B combined with BIST in an embodiment of the present disclosure
  • Figure 9 is a schematic diagram of a transceiver link including TX IQ Mod, BIST IQ Mod and RX IQ De-Mod in an embodiment of the present disclosure.
  • a transceiver link may include a transmitting link and a receiving link, etc.
  • the transmitting link may include a digital baseband signal source (Baseband), a direct digital frequency synthesizer (TX DDFS), an IQ digital-to-analog converter (IQ DAC), a low-pass filter (LPF), an IQ modulator (IQ Modulator), a power amplifier (PA), etc., which are connected in sequence.
  • the signal amplified by the power amplifier is radiated to a preset spatial area through a transmitting antenna.
  • the receiving link may include a low noise amplifier (LNA), a real mixer (Real Mixer), a trans-impedance amplifier (TIA), a low pass filter (LPF), a high pass filter (HPF), a real digital-to-analog converter (Real ADC), etc. connected in sequence, that is, the echo signal received by the receiving antenna is processed by the above-mentioned LNA, Real Mixer, TIA, LPF, HPF and Real ADC in sequence, and then converted into a real digital baseband signal.
  • the subsequent digital signal processing module processes the real digital baseband signal to obtain parameter information such as distance, speed, angle, height and micro-motion characteristics of the target.
  • the frequency of the TX-LO signal received by the IQ modulator in the transmitting link and the RX-LO signal received by the Real Mixer in the receiving link can be the same.
  • the signal output by the Baseband is a sine wave of x MHz
  • the TX-LO signal and the RX-LO signal may both be sine waves of z GHz.
  • the transmitting link adopts a digital phase-shifting architecture
  • the receiving link may adopt components of an analog architecture, i.e., IQ components are not required. Therefore, the sensor of the receiving link of the analog architecture can be effectively compatible, thereby effectively reducing the development cost of the entire transceiver link system.
  • the receiving link may include a receiving antenna, that is, the receiving antenna may be connected through a peripheral port of the chip and formed on a carrier such as a PCB board.
  • the receiving antenna may also be integrated on the chip package to form AiP or AoP, that is, a chip structure with a packaged antenna.
  • the transceiver link shown in FIG7B may include a transmitting link architecture and a receiving link similar to that in FIG7A (in order to avoid redundancy, the same parts will not be described in detail here), that is, the Real Mixer in the receiving link in FIG7A is adjusted to an IQ demodulator (IQ Demodulator), and the Real ADC is adjusted to an IQ ADC.
  • IQ Demodulator IQ Demodulator
  • the receiving link may include a low noise amplifier LNA), an IQ Demodulator (IQ Demodulator), transimpedance amplifier (TIA), low-pass filter (LPF), high-pass filter (HPF), IQ digital-to-analog converter (IQ ADC), etc., that is, the echo signal received by the receiving antenna is processed by the above-mentioned LNA, IQ Demodulator, TIA, LPF, HPF and IQ ADC in sequence and converted into an IQ digital baseband signal.
  • the subsequent digital signal processing module processes the IQ digital baseband signal to obtain parameter information of the target such as distance, speed, angle, altitude and micro-motion characteristics (i.e. micro-Doppler).
  • the transmitting link when performing self-calibration based on the transceiver link shown in FIG. 7B , as long as the signal output port of the transmitting link is directly connected to the signal input port of the receiving link through a transmission line, that is, the transmitting link directly sends the transmitting signal to the receiving link through the transmission line, so as to realize the self-calibration operation of the receiving and/or transmitting link without passing through the transmitting antenna and the receiving antenna.
  • there is a certain frequency deviation between the TX-LO signal received by the IQ modulator in the transmitting link and the RX-LO signal received by the IQ Demodulator in the receiving link For example, as shown in FIG.
  • the TX-LO signal can be a sine wave of z GHz
  • the RX-LO signal can be a sine wave of (z GHz-yMHz), that is, the frequency deviation at this time is y MHz.
  • the transmission link (the transmitter (Transmitter, TX) shown in the figure) can be calibrated by adding a receiving link (the receiver (Receiver, RX) shown in the figure), and the TX IQ imbalance compensation unit in the transmission link performs compensation operations based on the calibrated data.
  • the transmission link (the transmitter (Transmitter, TX) shown in the figure) can also be calibrated by multiplexing the receiving link (the receiver (Receiver, RX) shown in the figure) actually used for signal transmission and reception, and the TX IQ imbalance compensation unit in the transmission link and/or the receiving link performs compensation operations based on the calibrated data. Similar implementations can also be performed in other embodiments, which will not be described in detail later for the sake of simplicity.
  • an internal self-test module (Built-in Self-Test, BIST for short) module may be set at the RX-LO port of the IQ Demodulator of the receiving link shown in FIG. 7B , that is, as shown in FIG. 8 , based on the transceiver link structure shown in FIG. 7B , an IQ BIST architecture is set at the RX-LO port of the IQ Demodulator of the receiving link to achieve accurate calibration of the IQ in the receiving link.
  • the RX-LO port of the Demodulator inputs an LO signal with a preset frequency deviation.
  • an IQ BIST composed of a phase angle converter and an IQ modulator (IQ Modulator) is used to form a frequency-deviation signal based on the frequency deviation signal of another input signal BIST-LO of the IQ modulator, such as a TX-LO signal, through a phase angle converter.
  • the signal is input to the RX-LO port of the IQ Demodulator.
  • the TX-LO signal is a z GHz sine wave
  • the BIST-LO signal is a y MHz sine wave
  • the frequency-deviation signal input to the RX-LO port of the IQ Demodulator is (z GHz-yMHz). It should be noted that between different embodiments, x, y, and z are all schematic values, and the specific values may be the same or different.
  • the transmit link of the digital phase shifter architecture can also be calibrated by multiplexing the receive link in the transceiver link; wherein, in other embodiments, the calibration operation of the transmit link using the receive link, and the calibration operation of the receive link using the transmit link, can be achieved by multiplexing the corresponding receive link or transmit link in the link that actually performs signal transmission and reception, and can also be achieved by adding a corresponding calibration receive link or calibration transmit link to achieve the calibration operation of the corresponding transmit link or receive link in the link that actually performs signal transmission and reception.
  • the IQ BIST may include a phase angle converter and an IQ modulator (IQ Modulator).
  • the phase angle converter is used to realize the separate calibration of the I and Q paths in the transmit link of the digital architecture, while the other input signal BIST-LO of the IQ modulator may be a y MHz sine wave, which is used to simulate the characteristics related to the echo signal formed by the reflection of the transmit signal by the target, wherein x, y, and z in Figure 8 are all positive numbers, and x ⁇ y ⁇ z, and can generally be between 0 and 1000.
  • a TX IQ imbalance compensation unit (TX IQ Imbalance Compensation) may be set in the transmitting link (for example, between the TX DDFS and the IQ DAC), and/or a TX IQ imbalance compensation unit (TX IQ Imbalance Compensation) may be set in the receiving link (for example, after the IQ ADC), so that the transmitted and/or received signals can be supplemented based on the calibration parameters (or coefficients) obtained by the above-mentioned self-calibration operation to solve problems such as IQ imbalance.
  • the above-mentioned IQ BIST module can be set between the signal output port of the transmitting link and the signal input port of the receiving link, that is, the transmitting link directly sends the transmitting signal to the receiving link through the IQ BIST module, so as to realize the self-calibration operation of the receiving link and/or the transmitting link without passing through the transmitting antenna and the receiving antenna.
  • TX IQ Imbalance compensation an LO leakage compensation unit (TX LO leakage compensation) and an HD3 compensation unit (TX HD3 compensation) may be added to the transmission link to form a compensation unit (TX compensation) including an LO leakage compensation unit (TX LO leakage compensation), an IQ imbalance compensation unit (TX IQ Imbalance compensation) and/or an HD3 compensation unit (TX HD3 compensation).
  • TX compensation including an LO leakage compensation unit (TX LO leakage compensation), an IQ imbalance compensation unit (TX IQ Imbalance compensation) and/or an HD3 compensation unit (TX HD3 compensation).
  • Figure 10 is a schematic diagram of a transceiver link including an auxiliary circuit and a BIST IQ Mod in an embodiment of the present disclosure
  • Figure 11 is a schematic diagram of another transceiver link including an auxiliary circuit and a BIST IQ Mod in an embodiment of the present disclosure.
  • a transceiver link may include a transmitting link, a receiving link, and a calibration link.
  • the transmitting link may include a TX digital baseband signal source (TX Baseband), a direct digital frequency synthesizer (TX DDFS), a compensation unit (Compensation), an IQ digital-to-analog converter (IQ DAC), a low-pass filter (LPF), an IQ modulator (IQ Modulator), and a power amplifier (PA), etc., which are connected in sequence.
  • the signal amplified by the power amplifier is radiated to a preset spatial area through a transmitting antenna.
  • the receiving link may include a low noise amplifier (LNA), a real mixer (Real Mixer), a trans-impedance amplifier (TIA), a high-pass filter (HPF), a variable automatic gain amplifier (VGA), a real digital-to-analog converter (Real ADC) and an RX Baseband for TX RF calibration, which are connected in sequence. That is, the echo signal received by the receiving antenna is processed by the above-mentioned LNA, Real Mixer, TIA, HPF, VGA and Real ADC in sequence and converted into a real digital baseband signal. The subsequent digital signal processing module processes the real digital baseband signal to obtain parameter information of the target such as distance, speed, angle, height and micro-motion characteristics.
  • LNA low noise amplifier
  • TIA trans-impedance amplifier
  • HPF high-pass filter
  • VGA variable automatic gain amplifier
  • Real ADC real digital-to-analog converter
  • RX Baseband for TX RF calibration which are connected in sequence. That is, the echo signal
  • the compensation unit (TX compensation) set between the TX DDFS and the IQ DAC may include an LO leakage compensation unit (TX LO leakage compensation), an IQ imbalance compensation unit (TX IQ Imbalance compensation) and/or an HD3 compensation unit (TX HD3 compensation) and other units, so as to realize the corresponding compensation operations such as LO leakage, IQ Imbalance and HD3 in the transmit link of the digital phase shifter architecture.
  • TX LO leakage compensation TX LO leakage compensation
  • TX IQ Imbalance compensation IQ imbalance compensation unit
  • HD3 compensation HD3 compensation unit
  • a calibration module may be provided between the transmitting link and the receiving link, and the calibration module may be configured to multiplex the receiving link to perform operations such as calibration on the transmitting link of the above-mentioned digital phase shifter architecture.
  • the compensation module i.e., the various compensation units in the aforementioned embodiments
  • the compensation module may be based on the parameters or coefficients obtained by the calibration operation of the calibration module, so as to realize the compensation operation of the transmitted signal at the transmitting link end.
  • a corresponding receiving compensation module may be provided in the receiving link simultaneously or separately, that is, at this time, the receiving compensation module may be based on the parameters or coefficients obtained by the above-mentioned calibration operation, so as to realize the compensation of the echo signal at the receiving link end.
  • the calibration module may include a BIST unit and an auxiliary circuit unit. That is, the output port of the transmitting link is connected to any node between the Real Mixer and the Real ADC in the receiving link through the BIST unit and the auxiliary circuit unit.
  • the IQ Modulator in the transmitting link generates a radio frequency signal of (z GHz ⁇ x MHz) based on the digital phase-shifted baseband signal of x MHz and the LO signal of z GHz, and then outputs it to the BIST unit through the output port.
  • the intermediate frequency signal is then input into a preset node in the receiving link to implement the calibration operation in the transmitting link.
  • the auxiliary circuit unit may be an orthogonal demodulator circuit, and the output end of the auxiliary circuit unit may be connected to any node among the nodes between the TIA and the HPF, the HPF and the VGA, and the VGA and the Real ADC in the receiving link.
  • the I and Q branches may be respectively connected to different transmitting links, as shown in FIG10, that is, the transmitting link is calibrated by multiplexing two receiving links.
  • the compensation units such as the LO leakage compensation unit (TX LO leakage compensation), the IQ imbalance compensation unit (TX IQ Imbalance compensation) and/or the HD3 compensation unit (TX HD3 compensation) in the above-mentioned compensation unit (TX compensation) may be used to implement compensation operations corresponding to the problems of LO leakage, IQ Imbalance and HD3 in the transmitting link of the digital phase shifter architecture based on the parameters obtained by calibration.
  • the above-mentioned BIST unit may include a phase angle converter and an IQ modulator (IQ Modulator) connected in sequence
  • the auxiliary circuit unit may include an LNA, an IQ De-Modulator and a TIA connected in sequence, that is, the phase angle converter receives the RF signal output from the transmission link, and one input end of the IQ Modulator is connected to the output end of the phase angle converter, and the other input end receives the BIST-LO signal of y MHz to generate a preset echo signal.
  • the LNA sends the received echo signal to one input end of the IQ De-Modulator after amplification, and the other input end of the IQ De-Modulator is used to receive the RX-LO signal of z GHz.
  • the two output branches (i.e., the I branch and the Q branch) of the IQ De-Modulator are respectively connected to the corresponding nodes in the corresponding receiving links after the TIA, so as to output the generated preset intermediate frequency signal to the two receiving links, thereby realizing a more efficient multiplexing receiving link design while realizing the calibration operation.
  • the TX LO signal can be used as a single-tone signal for point-by-point calibration; at the same time, the TX LO signal can also be used as a swept frequency signal for large-bandwidth calibration operations, and even the swept frequency bandwidth calibration can be used once to implement the calibration operation for the swept frequency signal of the entire frequency band.
  • the compensation coefficient of IQ Imbalance can be obtained in the time domain (Time-Domain) based on spectrum analysis, and can also be obtained in the frequency domain (Frequency-Domain) based on the spectrum peak ratio.
  • the ideal compensation coefficient in order to further improve the accuracy of the IQ Imbalance compensation coefficient, can be approximated by iterative calibration and compensation, or the ideal compensation coefficient can be obtained by multi-observation calibration and compensation.
  • the iterative calibration and compensation method it can be determined whether to stop the iterative operation based on the size relationship between the compensation coefficients of the two calibration compensations, or whether the difference between the compensation coefficients of the two calibration compensations meets the preset iterative conditions, and the compensation coefficient obtained when the iterative operation is stopped is used as the final compensation coefficient in the current scene for subsequent operations.
  • the measurement data obtained from each operation can be subjected to FFT (fast Fourier transform) and the corresponding amplitude and phase information can be obtained.
  • the measurement can be subtracted and normalized to obtain relevant data, and an observation matrix can be constructed; the corresponding compensation coefficient can be reversely solved based on the data obtained by inverting the observation matrix.
  • the compensation coefficient of LO leakage and/or HD3 can be obtained by using methods such as iterative calibration and compensation, or multiple observation calibration and compensation.
  • FIG. 12A is a schematic diagram of the structure of the mixer in the embodiment of the present disclosure.
  • the mixer includes a voltage-current converter (V/I Converter), a current switch (Current Switch) and a current-voltage converter (I/V Converter).
  • V/I Converter voltage-current converter
  • Current Switch Current Switch
  • I/V Converter current-voltage converter
  • the voltage-current converter converts the received voltage signal into a current signal
  • the current switch is connected to the voltage-current converter and the second signal generator, and is used to process the current signal output by the voltage-current converter using the local oscillator signal
  • the current-voltage converter is connected to the current switch, and is used to convert the current signal output by the current switch into a voltage signal.
  • the current signal output by the voltage-current converter contains a harmonic signal corresponding to the baseband signal.
  • the HD caused by the third-order nonlinearity of the baseband can be referred to as HD3.
  • the harmonic caused by the fifth-order nonlinearity is called HD5.
  • the current switch processes the current signal output by the voltage-current converter, the frequency of the harmonic is converted to the RF band after up-conversion. Since the operation complexity of suppressing the harmonic signal in the RF band is high, the hardware cost is high. If the harmonic signal in the RF band is not removed, it will affect the signal quality of the radar reception and transmission, and thus affect the accuracy of the radar measurement.
  • the compensation unit is used to input the generated cancellation signal into the signal transmission link to cancel the harmonic signal in the radio frequency signal.
  • the compensation unit is independent of the first signal generator.
  • the compensation unit may include a cancellation signal generator.
  • the cancellation signal output by the compensation unit can suppress the harmonic signal in the radio frequency signal, reduce the harmonic component in the radio frequency signal, and thus improve the signal quality of the radio frequency signal output by the transmitter.
  • the compensation unit uses feedback or according to the characteristics of the transmission wave to input the generated cancellation signal into the signal transmission link to cancel the harmonic signal in the RF signal output by the signal transmission link.
  • the cancellation signal has the characteristics of opposite phase and similar amplitude to the harmonic signal transmitted in the RF transmission circuit, so as to achieve the purpose of suppressing the harmonic signal.
  • the compensation unit generates a compensation signal including a cancellation effect according to parameters such as the phase, frequency, or amplitude of the baseband signal generated by the first signal generator, or even the path length combined with the LO signal.
  • FIG5A shows an example of a transmitter in which a compensation unit is connected to a signal transmission link.
  • the compensation unit is a TX compensation unit.
  • the TX compensation unit includes a generator (not shown) that can generate a cancellation signal according to the characteristics of the transmission wave.
  • the cancellation signal generator can be exemplified by a TX HD3 compensation unit as shown in FIG5B.
  • the baseband processor controls the orthogonal digital baseband signal generated by the TX DDFS, and the TX compensation unit generates an orthogonal compensation signal according to the parameters of the orthogonal digital signal, and merges the orthogonal compensation signal and the orthogonal digital signal and sends them to the IQ DAC to be converted into an analog baseband signal.
  • the mixer i.e., the IQ modulator in FIG. 5A
  • the PA amplifies the mixed signal and outputs it through the transmitting antenna.
  • the compensation signal cancels out at least part of the harmonic signals in the radio frequency transmission circuit, such as the HD3 harmonic signal. Therefore, the clutter in the transmitted radio frequency signal will be greatly reduced.
  • the radio frequency signal can be an FMCW signal.
  • the compensation unit generates a compensation signal according to the harmonic information obtained through the feedback of the RF transmission circuit.
  • FIG12B is a schematic diagram of the structure of the compensation unit in the transmitter shown in FIG5A. As shown in FIG12B, the compensation unit includes an acquisition circuit and a cancellation signal generator.
  • the acquisition circuit is coupled to the RF transmission circuit, and is used to acquire the signal in the RF transmission circuit to obtain an acquisition signal.
  • the acquisition signal (or sampling signal) can reflect the waveform information (also called harmonic parameters) in the harmonic signal, such as the phase of the main frequency signal, the phase of the harmonic signal, the frequency of the harmonic signal, the frequency of the main frequency signal, the power of the harmonic signal, the power of the main frequency signal, etc.
  • the harmonic parameters reflected by the acquisition signal are related to the information carried by the signal that can be acquired by the acquisition circuit.
  • the acquisition circuit is a power acquisition circuit
  • the corresponding acquisition signal includes the power of the main frequency.
  • the acquisition signal reflects the phase of the main frequency signal, the phase of the harmonic signal, the frequency of the harmonic signal, the frequency of the main frequency signal, the power of the harmonic signal, the power of the main frequency signal, etc.
  • At least one of the above harmonic parameters can be extracted by analog circuits.
  • the power of the main frequency signal is output through a coupler and a power detector.
  • the advantage of the digital circuit in the radar chip in frequency domain calculation is used to extract harmonic parameters.
  • a signal identical to the signal transmitted at the coupling point is obtained as a collection signal.
  • the collection signal carries the main frequency signal and the harmonic signal.
  • the collection signal is converted into a digital signal by an ADC and handed over to the digital circuit for calculation in the frequency domain to obtain more harmonic parameters.
  • the input end of the acquisition circuit is connected to the output end or the signal detection end of the mixer.
  • This method can detect the harmonic signal generated by the voltage-current converter and has a simplified acquisition circuit.
  • the signal calibration link the first input end of the signal calibration link is connected between the voltage-current converter and the current switch in the transmitting mixer in the main signal transmission path, the second input end is between the main signal transmission path and the transmitting antenna, and the signal output end is connected to the compensation circuit in the main signal transmission path, and is used to obtain the signal in the main signal transmission path from at least one of the first input end and the second input end, and determine the compensation information according to the obtained signal.
  • the input end of the acquisition circuit is connected to the detection end between the voltage-current converter and the current switch, and is coupled to the ADC.
  • the signal calibration link includes a calibration demodulator (IQ Demodulator), a multiplexer (MUX) and a calibration module (TX Calbration); wherein: the calibration demodulator is configured to obtain the signal in the signal transmission main path from the second input terminal and perform demodulation processing; the multiplexer has two signal input terminals and one signal input terminal, one of which is connected between the voltage-current converter and the current switch in the transmitting mixer, and the other is connected to the signal output terminal of the calibration demodulator, for outputting a signal corresponding to one of the two signal input terminals; the calibration module is configured to determine the compensation information according to the signal output by the multiplexer.
  • IQ Demodulator IQ Demodulator
  • MUX multiplexer
  • TX Calbration calibration module
  • the input end of the acquisition circuit is connected to the RF output end or RF detection end of the RF transmission circuit.
  • the RF output end is exemplified as the output end of the RF transmission circuit.
  • the RF detection end is exemplified as the input end or output end of at least one PA in the RF transmission circuit. This method can acquire more accurate harmonic parameters in the RF transmission circuit, but has a more complex circuit structure.
  • the acquisition circuit can obtain the acquisition signal through part or all of the circuits in the BIST module.
  • the input end of the acquisition circuit is coupled to the RF output end, which includes a down converter, a filter, etc. in sequence, and is connected to the IQ ADC to output a digital acquisition signal.
  • the down converter, the filter, etc. can reuse the BIST module or the receiver.
  • the collected signal is input to a cancellation signal generator, which is at least one circuit in the compensation unit.
  • the cancellation signal generator is connected to the first signal generator, so that the signal received by the radio frequency transmitting circuit includes both the baseband signal and the cancellation signal.
  • the cancellation signal generator includes the cancellation signal generator mentioned above and a digital circuit for extracting harmonic information, wherein the digital circuit for extracting harmonic information can be configured independently, or at least partially shared with the digital circuit in the radar chip.
  • the digital circuit for extracting harmonic information uses, for example, a digital circuit for processing difference frequency baseband signals in a radar chip to extract harmonic information such as harmonic frequency, main frequency, and main frequency power, and provides the information to a cancellation signal generator.
  • the cancellation signal generator generates a cancellation signal according to the received parameters.
  • the digital circuit for extracting harmonic information extracts the main frequency amplitude in the collected signal, and calculates the harmonic amplitude based on the difference between the preset main frequency amplitude and the harmonic amplitude.
  • the cancellation signal generator generates a harmonic signal compensation signal based on the calculated harmonic amplitude and other pre-configured harmonic parameters.
  • the pre-configured harmonic parameters can be calculated based on the sweep frequency range, phase, etc. of the main frequency signal to be transmitted by the radar chip.
  • the cancellation signal generator in the cancellation signal generator can be configured independently with the first signal generator, or at least partially shared.
  • the cancellation signal generated by the cancellation signal generator is input into the first signal generator, so that the baseband signal output by the first signal generator includes the cancellation signal.
  • the cancellation signal generator may include a third harmonic generator and a fifth harmonic generator.
  • the compensation unit further includes an adder, coupled to the cancellation signal generator and the first signal generator, to combine the baseband signal generated by the first signal generator and the cancellation signal generated by the cancellation signal generator.
  • the cancellation signal includes a cancellation signal Signal_HD3 generated by the third harmonic generator to cancel the third harmonic, and a cancellation signal Signal_HD5 generated by the fifth harmonic generator to cancel the fifth harmonic.
  • the cancellation signals Signal_HD3 and Signal_HD5 and the baseband signal generated by the first signal generator are combined by the adder and output to the RF transmission circuit.
  • the transmitter circuit examples provided in the present disclosure that use feedback to pre-input a cancellation signal into a radio frequency transmitting circuit can ensure that the radio frequency signal transmitted by the chip contains sufficiently low harmonic signals under different environments.
  • the present disclosure also provides a method for canceling harmonic signals in the above transmitter using a feedback mechanism, including: step 10, collecting signals in a signal transmission link to obtain a collection signal; wherein the signal transmission link is used to generate a radio frequency signal for radar detection, wherein the radio frequency signal contains a harmonic signal. Step 20, detecting the collection signal, generating a cancellation signal for canceling the harmonic signal, and outputting the cancellation signal to the signal transmission link.
  • the method provided by the embodiment of the present disclosure performs a collection operation on the signal in the signal transmission link to obtain the collected signal, and uses the collected signal to generate a cancellation signal, and outputs it to the signal transmission link, so as to use the cancellation signal to suppress the harmonic signal in the radio frequency signal and reduce the harmonic component in the radio frequency signal, thereby improving the signal quality of the radio frequency signal output by the transmitter, and further improving the receiving performance of the receiver for the radio frequency signal.
  • FIG7B shows an example of using a feedback mechanism to extract harmonic information in a transmitter so that the compensation unit generates a corresponding cancellation signal.
  • the compensation unit includes a TX HD3 compensation unit as an example.
  • the TX HD3 compensation unit generates a compensation signal according to the waveform characteristics of the received signal.
  • the feedback mechanism can be executed in the calibration mode of the radar chip to prevent the signal transmission power of the radar chip from being weakened during normal detection.
  • the baseband processor in the transmitter controls the orthogonal digital baseband signal generated by the TX DDFS, and sends it to the IQ DAC after merging the orthogonal digital cancellation signal generated by the TX compensation unit to convert it into an analog baseband signal.
  • the analog baseband signal is mixed with an analog cancellation signal used to cancel the harmonic signal in the transmission link.
  • the analog baseband signal enters the first mixer (i.e., the IQ modulator in Figure 7B) after LPF filtering.
  • the first mixer uses TX LO to mix the received filtered signal to obtain a radio frequency signal.
  • the radio frequency signal is coupled and output to the TX HD3 calibration circuit in the compensation unit through the receiver (the TX HD3 calibration box in Figure 7B).
  • the TX HD3 calibration circuit can be regarded as a digital circuit for extracting harmonic information.
  • the LNA amplifies the signal output by the transmitter and outputs it to the second mixer (i.e., the IQ demodulator in FIG7B ) to obtain a demodulated signal, which is then sent to the transimpedance amplifier for amplification, and then passes through the LPF and HPF in sequence, and then passes through the IQ ADC for analog-to-digital conversion before being sent to the TX HD3 calibration circuit.
  • the TX HD3 calibration circuit extracts the harmonic information in the transmitter from the feedback signal, and converts it into the parameters required to generate the cancellation signal through the upper controller, and provides it to the TX HD3 compensation unit.
  • the harmonic information obtained by the TX HD3 calibration circuit includes one or more of the following parameters: the initial phase, starting frequency, cutoff frequency, frequency change duration, center frequency, etc. of the harmonic signal (or main frequency signal).
  • the TX HD3 compensation unit or the upper controller determines the parameters used to generate the cancellation signal in the compensation unit according to the harmonic information, such as the initial phase, the frequency of the cancellation signal, the delay, etc.
  • cancellation operations of the third harmonic and/or the fifth harmonic in the above examples may be determined according to the requirements of the transmitter.
  • FIG. 13A is a schematic diagram of a digital pre-compensation HD3 architecture based on a cubic module in an embodiment of the present disclosure
  • FIG. 13B is a schematic diagram of a digital pre-compensation HD3 architecture based on a frequency doubling waveform generator module in an embodiment of the present disclosure.
  • the main source of HD3 in the active mixer is the third harmonic in the nonlinearity of V/IConverter
  • it can be achieved through a compensation architecture based on a cubic module as shown in FIG13A , or a compensation structure based on a triple frequency waveform generator as shown in FIG13B .
  • FIG13C is a calibration compensation schematic diagram of a transmission link based on a digital phase shifter architecture in an embodiment of the present disclosure.
  • the compensation operation for IQ Imbalance can be achieved by compensating the conjugate signal of the BB (baseband) signal to reversely cancel the image component, and this compensation operation method is not affected by the calibration method of IQ Imbalance.
  • the compensation for LO Leakage can be achieved by adjusting the DC component (i.e., DC bias) of the IQ two-way, and the calibration method of LO Leakage has no effect on its compensation scheme.
  • the compensation methods of HD3 based on the digital pre-compensation architecture of the digital cubic module and the digital pre-compensation architecture based on the frequency doubling waveform generator module will directly affect the subsequent calibration scheme and subsequent compensation process. Specifically:
  • the LO Leakage may be calibrated and compensated first, and then the root cause of the HD3 problem, that is, the compensation coefficient of HD3, may be calibrated under a stable DC bias; then the IQ Imbalance may be calibrated, and after further compensating for the IQ Imbalance, the third harmonic distortion may be compensated for the two IQ paths based on the results of the IQ Imbalance pre-compensation.
  • the compensation coefficient of HD3 can be calibrated, and IQ Imbalance can be calibrated and compensated under a stable DC bias; subsequently, the actual waveforms of the IQ and Q signals and the compensation coefficient of HD3 are calculated respectively through the compensation results, and the waveform information of the 3x and 5x frequencies that need to be pre-compensated can be inversely calculated.
  • LO Leakage may be calibrated and compensated first, and then the pre-compensation coefficients of HD3 and IQ Imbalance may be calibrated simultaneously through multiple (e.g., three) observations, and then the pre-compensation coefficients at the HD3 mirror position may be calibrated again (e.g., twice); finally, the 3x and 5x coefficients that need to be pre-compensated may be inversely calculated through the pre-compensation coefficients at the HD3 and HD3 mirror positions.
  • the observations in the disclosed embodiments are used to represent operations such as testing and comparative analysis of different test results.
  • the acquisition circuit has two acquisition branches, and the two acquisition branches can be switched dynamically; wherein, the input end of one acquisition branch is connected between the voltage-current converter and the current switch; the input end of the other acquisition branch is connected to the output end of the power amplifier, and the acquisition branch is provided with an IQ demodulator; in addition, in the structure shown in FIG13C , the acquisition circuit is also provided with a multiplexer, wherein the input end of the multiplexer is respectively connected to the output end of the two acquisition branches, and the output end is used to output the acquisition signal.
  • the acquisition signal can be an analog signal, that is, the output end of the acquisition circuit is connected to the IQ ADC; or, the acquisition signal can be a digital signal, then the acquisition circuit at least includes an IQ ADC.
  • the acquisition circuits in Figures 13A to 13C can be completed using non-orthogonal components.
  • a single-ended down-conversion mixer can be used instead of an IQ demodulator
  • a single-ended analog-to-digital converter can be used instead of an IQ ADC.
  • an embodiment of the present disclosure further provides a signal transmission method, which is applied to an electromagnetic wave device having at least one signal transmission link, and the signal transmission method includes: step 1401, determining the phase of the radio frequency transmission signal of each signal transmission link; step 1402, determining the initial phase of the transmitting digital baseband signal in each signal transmission link according to the phase of the radio frequency transmission signal; step 1403, generating a transmitting digital baseband signal according to the determined initial phase; step 1404, converting the transmitting digital baseband signal into a transmitting analog baseband signal, and performing a phase shift operation on the transmitting local oscillator signal based on the transmitting analog baseband signal.
  • the signal transmission method of the embodiment of the present disclosure generates a digital baseband signal in the digital domain by using a digital phase shifter architecture, which has better orthogonality and lower sidelobes.
  • the phase shift phase can be generated very accurately, so that the phase modulation accuracy is higher, thereby realizing a vehicle-mounted radar system with a high-precision digital phase shifting function, reducing the isolation requirements between antennas, and having the advantages of small link loss, low cost, and no need for offline calibration. It can also support more flexible wave transmission schemes, such as high-performance Doppler division multiplexing and frequency division multiplexing, and can support frequency response compensation in the digital domain.
  • the digital baseband signal at the transmitting end is a single-tone signal, and the local oscillator signal at the transmitting end is a swept-frequency signal; or, the digital baseband signal at the transmitting end is a swept-frequency signal, and the local oscillator signal at the transmitting end is a single-tone signal.
  • the frequency bandwidth of the swept frequency signal is above 2 GHz.
  • the ideal compensation coefficient in order to further improve the accuracy of the IQ Imbalance compensation coefficient, can be approximated by iterative calibration and compensation, or the compensation coefficient used in the actual working state can be obtained by multi-observation calibration and compensation.
  • the determination of the compensation coefficient needs to additionally consider the influence of the link characteristics on the calculation accuracy of the compensation coefficient.
  • the compensation coefficient h under actual working conditions is The compensation coefficient H under ideal conditions is Due to the obvious difference between the two expressions, the compensation coefficient under actual working conditions is different from that under ideal conditions.
  • f t 5 MHz is a narrowband signal compared to the carrier frequency of 76-81 GHz, they can be approximately considered equal;
  • FIG14B is a schematic diagram for explaining the principle of determining the compensation coefficient provided by the embodiment of the present disclosure.
  • FIG14B (a) represents the phase and amplitude of the compensation coefficient H in the ideal state of the signal transmission link
  • FIG14B (b) represents the phase and amplitude of the compensation coefficient H in the ideal state of the signal transmission link, the phase and amplitude of the compensation coefficient h in the actual working state, and the difference D between the compensation coefficient h in the actual working state and the compensation coefficient H in the ideal state
  • FIG14B (c) represents that the three vectors in FIG14B (b) are not changed by the change of amplitude; therefore, it is only necessary to control the compensation coefficient h to be infinitely close to the compensation coefficient H in phase, see FIG14B (d) for details.
  • (e) in Figure 14B indicates that area 1 indicates that the compensation coefficient h is close to the compensation coefficient H, and area 2 indicates that the compensation coefficient h is not close to the compensation coefficient H; (f) in Figure 14B optimizes the area in (e) in Figure 14B, including reducing the size of area 1, thereby increasing area 2, and further dividing the increased area 2 to obtain area 3 and a new area 2, wherein the compensation coefficient h in area 3 is not close to the compensation coefficient H, and the new area 2 indicates that the compensation coefficient with the opposite value of the phase to the compensation coefficient h is close to the compensation coefficient H; for example, the compensation coefficient h in Figure 14B(g) and Figure 14B(h) are not in area 1, but the compensation coefficient h in Figure 14B(h) is closer to the compensation coefficient H than the compensation coefficient h in Figure 14B(g), which can be clearly known by the length of the difference D in Figure 14B(g) and Figure 14B(h).
  • the embodiments of the present disclosure propose an iterative calibration and compensation method.
  • whether to stop the iterative operation can be determined based on the size relationship between the compensation coefficients of the two calibration compensations, or whether the difference between the compensation coefficients of the two calibration compensations meets the preset iteration conditions, and the compensation coefficient obtained when the iterative operation is stopped is used as the final compensation coefficient in the current scene for subsequent operations.
  • the compensation coefficient of LO leakage and/or HD3 can be obtained by using methods such as iterative calibration and compensation, or multiple observation calibration and compensation.
  • an embodiment of the present disclosure provides a signal calibration link of a main signal transmission path, wherein the main signal transmission path is used to generate a radio frequency transmission signal after performing a compensation operation on the generated signal according to a compensation coefficient, so as to achieve target detection and/or communication, wherein: the signal calibration link is configured to obtain current observation information of the main signal transmission path under the current compensation coefficient; and when the current observation information meets the iteration condition, the current compensation coefficient is used as the compensation coefficient used for the compensation operation of the signal transmission link; otherwise, the current compensation coefficient is iterated until the obtained observation information meets the iteration condition.
  • a signal calibration method may include the following steps: Step A1: With an initial compensation coefficient h(0) of 0, obtain the signal state of the signal transmission link and obtain the initial observation information O(0). Step A2: Perform a first compensation on the signal transmission link using the first compensation coefficient h(1) determined by the initial observation information O(0), obtain the signal state of the signal transmission link and obtain the first observation information O(1). Step A3: If the difference between the initial observation information O(0) and the first observation information O(1) is greater than a preset difference threshold, determine the kth compensation coefficient h(k) according to the k-1th compensation coefficient h(k-1) and the k-1th observation information O(k-1), where k is an integer greater than or equal to 2.
  • the first compensation coefficient h(1) is updated to the initial observation value O(0) as the basis, and the sum between the k-1th compensation coefficient h(k-1) and the k-1 observation information O(k-1) is iteratively calculated to determine the kth compensation coefficient h(k).
  • the first compensation coefficient h(1) is updated to the inverse of the initial observation value O(0) as the basis, and the difference between the k-1th compensation coefficient h(k-1) and the k-1 observation information O(k-1) is iteratively calculated to determine the kth compensation coefficient h(k).
  • Step A4 If the difference between the initial observation information O(0) and the first observation information O(1) is less than a preset difference threshold, the initial observation information O(0) is adjusted according to the preset phase adjustment amount to obtain the adjusted first compensation coefficient h(1), and then the new first observation information O(1) is obtained again according to the adjusted first compensation coefficient h(1), and the kth compensation coefficient h(k) is determined based on the new first observation information O(1); wherein the difference between the initial observation information O(0) and the new first observation information O(1) is greater than the difference threshold.
  • the sum of the k-1th observation information O(k-1) and the adjusted k-1th observation information O(k-1) obtained by processing the k-1th observation information O(k-1) by the phase adjustment amount is iteratively calculated to determine the kth compensation coefficient h(k).
  • the first compensation coefficient h(1) is updated to the inverse of the adjusted first compensation coefficient h(1), and the difference between the k-1th compensation coefficient h(k-1) and the adjusted k-1th observation information O(k-1) obtained by processing the k-1th observation information O(k-1) by the phase adjustment amount is iteratively calculated to determine the k-th compensation coefficient h(k).
  • each iterative operation process in step A3 and step A4 it is determined whether the observation information corresponding to each compensation coefficient is less than a preset iteration threshold; if the observation information is less than the iteration threshold, the iterative operation is stopped, and the compensation coefficient used by the observation information less than the iteration threshold is used as the compensation coefficient.
  • Step B3 Compare the magnitude relationship between O IQ (0) and O IQ (1) to determine the subsequent iteration relationship; if
  • Step B4 If the relationship in step B3 is not satisfied, let Re-obtain O IQ (1): If
  • + ⁇ , where ⁇ represents the threshold, the subsequent iteration expression is as follows: Where k 2, 3, .... If
  • - ⁇ , where ⁇ represents the threshold, the subsequent iteration expression is as follows: Where k 2, 3, ... Step B5: When O IQ (k) is less than a certain threshold (where the threshold value is less than or equal to -50db), the iteration is stopped.
  • Step C3 Compare the magnitude relationship between O IQ (0) and O IQ (1) to determine the subsequent iteration relationship; if
  • Step C4 If the relationship in step C3 is not satisfied, let Re-obtain O DC (1): If
  • the compensation operation of harmonic distortion is taken as an example for description, wherein the symbols mentioned in the above embodiments are distinguished by the subscript HD3.
  • the initial phase of the baseband (BB) signal is x degrees
  • the initial phase of the subsequent 3BB signal needs to be 3x degrees
  • O(n) represents the amplitude phase value of the frequency point corresponding to the 3BB signal ⁇ the amplitude phase value of the frequency point corresponding to the BB signal.
  • step D3 Compare the size relationship between O HD3 (0) and O HD3 (1) to determine the subsequent iteration relationship; if
  • the embodiments of the present disclosure also propose a multi-observation calibration and compensation method.
  • FFT fast Fourier transform
  • the compensation coefficient of LO leakage and/or HD3 can be obtained by using methods such as iterative calibration and compensation, or multiple observation calibration and compensation.
  • a signal calibration link of a main signal transmission path wherein the main signal transmission path is used to generate a radio frequency transmission signal after compensating the generated signal according to a compensation coefficient, so as to achieve target detection and/or communication
  • the signal calibration link is configured to determine initial observation information O(0), first observation information O(1) and second observation information O(2) corresponding to the main signal transmission path under the conditions of different values of an initial compensation coefficient h(0), a first compensation coefficient h(1) and a second compensation coefficient h(2); using the initial observation information O(0), the first observation information O(1) and the second observation information O(2), determine a third compensation coefficient h(3) as the compensation coefficient used in the compensation operation of the signal transmission link.
  • the method for quadrature imbalance compensation may include: step E1: taking the initial compensation coefficient h(0) as 0, obtaining the signal state of the signal transmission link, and obtaining the initial observation information O(0).
  • step E2 performing a first compensation on the signal transmission link using the first compensation coefficient h(1) determined by the initial observation information O(0), obtaining the signal state of the signal transmission link, and obtaining the first observation information O(1); and, performing a second compensation on the signal transmission link using the second compensation coefficient h(2) determined by the initial observation information O(0), obtaining the signal state of the signal transmission link, and obtaining the second observation information O(2); wherein the second compensation coefficient h(2) and the first compensation coefficient h(1) have different values, and the second observation information O(2) and the first observation information O(1) have different values.
  • Step E3 Obtain the ratio between the first difference and the initial observation information O(0) to obtain a first ratio d1; and obtain the ratio between the second difference and the initial observation information O(0) to obtain a second ratio d2; wherein the first difference is the difference between the initial observation information O(0) and the first observation information O(1), and the second difference is the difference between the initial observation information O(0) and the second observation information O(2).
  • Step E4 Under the condition that the third observation information O(3) is 0, determine the first coefficient x1 corresponding to the first compensation coefficient h(1) and the second coefficient x2 corresponding to the second compensation coefficient h(2) according to the first ratio d1 and the second ratio d2; specifically, construct a 2*2 matrix, wherein the first row of the 2*2 matrix records the numerical value of the real part and the numerical value of the imaginary part of the first ratio, and the second row records the numerical value of the real part and the numerical value of the imaginary part of the second ratio; calculate the product between the inverse matrix of the 2*2 matrix and the first matrix of 1 row and 2 columns to obtain a second matrix of 1 row and 2 columns, wherein the value of the first row in the first matrix is 1, the value of the second row is 0, wherein the first row of the second matrix is the value of the first coefficient x1, and the second row is the value of the second coefficient x2.
  • Step E5 Calculate the product between the first coefficient x1 and the first compensation coefficient h(1) to obtain a first multiplication result; and calculate the product between the second coefficient x2 and the second compensation coefficient h(2) to obtain a second multiplication result; use the first multiplication result and the second multiplication result as the third compensation coefficient h(3).
  • Step 5 The Nature of Compensation:
  • the method for LO leakage may include: Step F1: With the initial compensation coefficients of the I-path signal and the Q-path signal both being 0, the signal state of the signal transmission link is obtained to obtain the initial observation information O(0).
  • Step F3 Obtain the ratio between the first difference and the initial observation information O(0) to obtain the in-line ratio CI; and, obtain the ratio between the second difference and the initial observation information O(0) to obtain the orthogonal ratio CQ; wherein the first difference is the difference between the initial observation information O(0) and the first observation information O(1), and the second difference is the difference between the initial observation information O(0) and the second observation information O(2).
  • Step F4 Under the condition that the third observation information O(3) is 0, determine the third compensation coefficient h(3) of the in-phase signal and the third compensation coefficient h(3) of the quadrature signal according to the in-phase ratio CI and the quadrature ratio CQ;
  • a 2*2 matrix is constructed, wherein the first row of the 2*2 matrix records the values of the real part of the in-phase ratio CI and the orthogonal ratio CQ, and the second row records the values of the imaginary part of the in-phase ratio CI and the orthogonal ratio CQ;
  • Step 4 Differencing the measurement and normalizing it to obtain:
  • Step 5 The Nature of Compensation:
  • Step G1 With the initial compensation coefficient h(0) being 0, the signal state of the signal transmission link is obtained to obtain the initial observation information O(0).
  • Step G2 The signal transmission link is compensated for the first time using the first compensation coefficient h(1) determined by the initial observation information O(0), the signal state of the signal transmission link is obtained, and the first observation information O(1); and, the signal transmission link is compensated for the second time using the second compensation coefficient h(2) determined by the initial observation information O(0), the signal state of the signal transmission link is obtained, and the second observation information O(2); wherein the second compensation coefficient h(2) and the first compensation coefficient h(1) have different values, and the second observation information O(2) and the first observation information O(1) have different values.
  • Step G3 Obtain the ratio between the first difference and the initial observation information O(0) to obtain the first ratio d1; and obtain the ratio between the second difference and the initial observation information O(0) to obtain the second ratio d2; wherein the first difference is the difference between the initial observation information O(0) and the first observation information O(1), and the second difference is the difference between the initial observation information O(0) and the second observation information O(2).
  • Step G4 Under the condition that the third observation information O(3) is 0, determine the first coefficient x1 corresponding to the first compensation coefficient h(1) and the second coefficient x2 corresponding to the second compensation coefficient h(2) according to the first ratio d1 and the second ratio d2.
  • a 2*2 matrix is constructed, wherein the first row of the 2*2 matrix records the values of the real part and the imaginary part of the first ratio, and the second row records the values of the real part and the imaginary part of the second ratio; the product between the inverse matrix of the 2*2 matrix and the first matrix of 1 row and 2 columns is calculated to obtain a second matrix of 1 row and 2 columns, wherein the value of the first row in the first matrix is 1, and the value of the second row is 0, wherein the first row of the second matrix is the value of the first coefficient x1, and the second row is the value of the second coefficient x2.
  • Step G5 Calculate the product between the first coefficient x1 and the first compensation coefficient h(1) to obtain a first multiplication result; and, The product of the second coefficient x2 and the second compensation coefficient h(2) is calculated to obtain a second multiplication result; the first multiplication result and the second multiplication result are used as the third compensation coefficient h(3).
  • Step 3 When hour,
  • Step 4 Subtract and normalize the measurements to obtain:
  • the architecture of the transmitting link and/or receiving link of the digital phase shifter architecture recorded in the embodiments of the present disclosure can be used not only for transmitting and/or receiving signals in electromagnetic wave sensors, but also for addressing the problems of unequal lengths of antenna feed lines between different transmitting/receiving links, unequal lengths of feed lines of TXLO (i.e., between LO and the mixers in each transmitting channel), and using the architecture of the transmitting link and/or receiving link of the digital phase shifter architecture recorded in the embodiments of the present disclosure as an auxiliary link to calibrate the receiving link and/or transmitting link of the analog phase shifter architecture.
  • Radio waves include short waves, medium waves, long waves and microwaves, etc.
  • Microwaves include centimeter waves (i.e. electromagnetic waves in the range of 3GHz to 30GHz, such as electromagnetic waves in the range of 3.1GHz to 10.6GHz, electromagnetic waves in the frequency band of 24GHz, etc.) and millimeter waves (i.e.
  • electromagnetic waves in the range of 30GHz to 300GHz such as electromagnetic waves in the frequency band of 60GHz, electromagnetic waves in the frequency band of 77GHz (such as 77GHz to 81GHz, etc.)
  • light waves may include ultraviolet rays, visible light, infrared rays, and lasers, etc., among which the electromagnetic wave frequency band of lasers is (3.846 to 7.895)*10 ⁇ 5GHz, i.e., lasers are included in part of the frequency bands of ultraviolet rays and visible light.
  • Figure 15 is a schematic diagram of a transmission link with at least two transmission channels in an embodiment of the present disclosure
  • Figure 16 is a structural schematic diagram of a digital LO signal generator in an embodiment of the present disclosure
  • Figure 17 is a structural schematic diagram of a transmission link including a feeder unequal length compensation module in an embodiment of the present disclosure.
  • TXLO feeder length between the local oscillator signal and the mixer (Mixer) in each transmission channel
  • the antenna design using unequal length transmission antenna feeders is increasingly widely used, especially for close-range measurement, relatively closed space areas (such as indoor or cabin lights) and other application scenarios, or in various scenarios using packaged antenna chips (AiP), such as wide frequency sweeping application scenarios.
  • AuP packaged antenna chips
  • the embodiment of the present disclosure provides a compensation solution for the relative delay caused by the unequal feeder lengths based on the above-mentioned digital phase shifting architecture, so as to effectively improve the quality of the transmitting and receiving signals.
  • the transmitting antenna array of the digital phase shifter architecture includes four transmitting channels, each transmitting channel shares a waveform control module (waveform control) and a local oscillator source LO, and includes a transmitting direct digital frequency synthesizer TX DDFS, an IQ digital-to-analog converter DAC, an analog low-pass filter Analog LPF, an IQ mixer Mixer, a power amplifier PA and a transmitting antenna, etc., which are connected in sequence.
  • waveform control waveform control
  • LO local oscillator source LO
  • the waveform control module is respectively connected to the TX DDFS in each transmitting channel, and the local oscillator source LO is respectively connected to an input end of the Mixer in each transmitting channel to up-mix the phase-shifted signal output by the LPF in each transmitting channel to form a radio frequency signal for transmission through each transmitting antenna.
  • preset compensation can be performed in the transmitting direct digital frequency synthesizer TX DDFS to reduce the delay problem of the transmitting signal caused by the above-mentioned unequal feeder lengths.
  • the shortest feeder from the local oscillator LO to the Mixer in one of the transmit channels can be used as a benchmark, and then the length difference between the Mixer in the remaining transmit channels and the local oscillator LO compared to the benchmark can be obtained.
  • the corresponding delay value can be obtained by dividing the length difference of the above-mentioned transmit channels by the speed c.
  • the corresponding compensation can be directly performed in TX DDFS to obtain the delay of each transmit channel compared to the benchmark.
  • a similar method can be used to address the problem of unequal feeder lengths between the PA of each transmit channel and its transmit antenna to obtain and compensate for the delay problem caused by the feeder differences between the transmit channels.
  • the delay can be compensated by using a digital LO generator based on the delay ⁇ caused by the feeder length compared to the reference, the sweep period T, the sweep bandwidth ⁇ , the sweep start frequency f c and other parameters, and adapting the clock signal of the subsequent DAC.
  • ⁇ the delay difference of the feeder corresponding to each transmission channel compared to the reference feeder
  • Figures 16 and 17 illustrate the structural diagrams of a transmission channel
  • Figure 15 shows a transmission antenna array with four transmission channels.
  • the transmission antenna array only needs to include at least two (such as two, three or five, etc.) transmission channels, and compensation operations can also be performed during the reception of signals and/or signal processing.
  • a similar idea can also be used to perform compensation directly in the TX DFFS or in the reception of signals and/or signal processing.
  • FIG18 is a schematic diagram of using an auxiliary circuit to calibrate and compensate a transceiver link in an embodiment of the present disclosure
  • FIG19 is a schematic diagram of using an auxiliary transmitting circuit to calibrate and compensate a receiving link in an embodiment of the present disclosure.
  • the signal transceiver link may generally include an antenna, RF analog devices, baseband analog devices, and a baseband digital processor.
  • the RF analog devices may include a phase-locked loop (PLL), a power amplifier (PA), a phase shifter (PS), a low-noise amplifier (LNA), a mixer local oscillator (LO), and a power detector (PD), while the baseband analog devices may include a low-pass filter (LPF), a high-pass filter (HPF), an analog-to-digital converter (ADC), etc.
  • PLL phase-locked loop
  • PA power amplifier
  • PS phase shifter
  • LNA low-noise amplifier
  • LO mixer local oscillator
  • PD power detector
  • the baseband analog devices may include a low-pass filter (LPF), a high-pass filter (HPF), an analog-to-digital converter (ADC), etc.
  • LPF low-pass filter
  • HPF high-pass filter
  • ADC analog-to-digital converter
  • the frequency responses of RF analog devices and circuits in the transmission link (TX), baseband analog devices and circuits in the receiving link (RX), and RF analog devices and circuits in the receiving link (RX) may have unbalanced defects, PD accuracy problems may exist at the PA output port and LNA input port, and there may be local oscillator leakage (LO leakage) in the receiving link (RX).
  • TX transmission link
  • RX baseband analog devices and circuits in the receiving link
  • RX RF analog devices and circuits in the receiving link
  • LO leakage local oscillator leakage
  • the analog device works at different frequencies, its amplitude-frequency response will also be different, which is equivalent to adding an unexpected window function to the received signal, which will adversely affect subsequent operations such as target distance and speed estimation, especially in multi-target scenarios, and may even cause false targets to be detected or real targets to be missed.
  • the analog device works at different frequencies, its amplitude-frequency response will also be different, which is equivalent to adding an unexpected window function to the received signal, which will adversely affect subsequent operations such as target distance and speed estimation, especially in multi-target scenarios, and may even cause false targets to be detected or real targets to be missed.
  • there will be different frequency responses between different receiving links which will introduce additional noise information such as different phases and amplitudes to the receiving link.
  • the introduced phase will directly affect the subsequent estimation of the echo direction of the target, thereby reducing the accuracy of the target angle estimation.
  • the errors of analog devices and circuits can be measured by external equipment, and then certain compensation operations can be performed on them, such as using Bench calibration or ATE calibration to calibrate the performance of analog devices and circuits of the receiving link.
  • the disclosed embodiments also provide a scheme for calibrating the transmit link and/or receive link based on an auxiliary link method, so as to achieve real-time calibration of analog devices and circuits while realizing precise calibration without the involvement of external devices, thereby effectively reducing the impact of changes in RF device parameter indicators due to environmental changes.
  • the electromagnetic wave signal transmission and reception link can be calibrated in real time by setting up an auxiliary calibration circuit (Auxiliary).
  • the auxiliary calibration circuit i.e., auxiliary calibration link
  • the auxiliary calibration circuit can be integrated into the electromagnetic wave sensor, thereby achieving real-time and accurate calibration without the involvement of external devices, thereby effectively reducing the impact of changes in RF device parameter indicators caused by environmental changes.
  • an auxiliary receiving path (Auxiliary Reveiver, referred to as ARX) can be set at a position near the transmission path in the sensor to achieve real-time calibration operation of the transmission path; similarly, for the receiving path, an auxiliary transmitting path (Auxiliary Transmitter, referred to as ATX) can also be set at a position near the receiving path (i.e., the signal receiving link) in the sensor to achieve real-time calibration operation of the receiving path.
  • the auxiliary path can also be set between two paths to be calibrated, so that an auxiliary path can be reused to calibrate at least two different paths.
  • a common ARX is set between the two transmission paths, and a common ATX is set between the two receiving paths, and the problem of wiring or excessively long lines can be further avoided.
  • two ARXs and two ATXs can be set, that is, each ARX is set between the two transmission channels, and each ATX is set between the two receiving channels, so as to effectively reduce the difficulty of wiring and improve the real-time performance of calibration.
  • auxiliary calibration paths can be set for RF Rx and Rx BB respectively, such as setting a RF auxiliary transmission unit (abbreviated as RF ATX) for RF Rx and a baseband auxiliary transmission unit (abbreviated as IF ATX) for Rx BB.
  • RF ATX RF auxiliary transmission unit
  • IF ATX baseband auxiliary transmission unit
  • the ARX can be calibrated by setting a RF tone signal transmitter (RF Tone Generator), and then the calibrated ARX can be used to calibrate the transmission channel (TX).
  • the Rx BB may be calibrated using the IF ATX first, and then the calibrated Rx BB may be calibrated using the calibrated Rx BB, and finally the Rx BB may be calibrated using the calibrated RF ATX, thereby achieving the calibration operation of the entire receiving path.
  • the calibration scheme for the receiving channel (Receiver) is described, that is, the receiving channel may include a receiving antenna (or receiving port), LNA, Mixer, TIA, LPF, HPF, Real ADC and other devices connected in sequence, and a PD connected to the LNA input port, and the receiving channel may be divided into RF Rx and Rx BB based on the Mixer, that is, along the direction of signal transmission, the Mixer and the part before it are defined as RF Rx, and the rest may be defined as Rx BB; as shown in FIG19 , RF Rx may include a receiving antenna (or receiving port), LNA, Real Mixer and PD, etc.
  • the IF ATX corresponding to the Rx BB of the above-mentioned receiving path may include a frequency divider (such as Freq Divider 1/N) and a Real DAC (such as 1-bit Real DAC) connected in sequence, that is, the output end of the Real DAC is connected to the output end of the Real Mixer.
  • a frequency divider such as Freq Divider 1/N
  • a Real DAC such as 1-bit Real DAC
  • the IF ATX may include a frequency divider (such as Freq Divider 1/N) and a bit digital-to-analog converter (such as 1-bit Real DAC) connected in sequence, that is, the IF ATX may stably generate a single-tone signal (single Tone) under various environments through the frequency divider and the digital-to-analog converter, and the single-tone signal may also be freely configured to various frequencies.
  • a frequency divider such as Freq Divider 1/N
  • a bit digital-to-analog converter such as 1-bit Real DAC
  • the IF ATX may be calibrated before the IF ATX calibrates the Rx BB, such as calibration of the intermediate frequency (IF) frequency response and DC calibration in the IF ATX.
  • the calibrated IF ATX may be used to calibrate the Rx BB of the receiving path, such as by transmitting single-tone signals of different frequencies through the calibrated IF ATX to complete calibration and compensation operations on the Rx BB part.
  • the frequency (e.g., tens of MHz) of the single-tone signal transmitted by the IF ATX may be determined based on the sampling frequency of the Real ADC in the receiving path.
  • the RF ATX set for the RF Rx of the corresponding receiving channel may include a TX DDFS, an IQ imbalance compensation unit (TX IQ Comp), an adder, an IQ digital-to-analog converter (IQ DAC), a low-pass filter (LPF), an amplifier (such as PA), a multiplier, a local oscillator (LO, not shown in the figure) and a squarer (x ⁇ 2) connected in sequence.
  • the output of the multiplier is connected to the output of the Mixer in the receiving path through the squarer to form a first calibration branch.
  • the output of the multiplier is also connected to the link between the PD and the transmitting antenna in the receiving path to form a second calibration branch.
  • the IQ imbalance compensation unit can be configured to compensate for the imbalance of IQ in the RF ATX
  • the adder can be configured to compensate for the local oscillator leakage (LO Leakage) in the RF ATX.
  • the squarer can be configured to compensate for the residual sideband effect caused by the imbalance of IQ in the RF ATX.
  • the RF ATX may be calibrated before the RF ATX calibrates the Rx ATX, such as calibration operations for local oscillator leakage, RF frequency response, IQ imbalance, and other issues in the RF ATX.
  • the calibrated RF ATX is used to calibrate the RF Rx of the receiving path.
  • the calibrated RF ATX may be used to transmit single-tone signals of different frequencies, and then the baseband signal processing module in the receiving link may be used to calibrate and compensate the power detector (PD) at the input end of the LNA in the RF part of the receiving path RF Rx, the total gain from the LNA to the ADC in the receiving path, and the auxiliary calibration of the frequency response.
  • PD power detector
  • the IF ATX when calibrating a receiving path (Receiver) using IF ATX and RF ATX, the IF ATX may be calibrated first, and the calibrated IF ATX may be used to calibrate the Rx BB in the receiving path. The calibrated Rx BB may then be used to calibrate the RF ATX through the first calibration branch of the RF ATX (i.e., through a squarer). Finally, the second calibration branch of the calibrated RF ATX may be used to calibrate the RF Rx in the receiving path, thereby achieving calibration and compensation operations for the entire receiving path.
  • the Rx BB and Real DAC can be calibrated by 1-bit Real DAC, and then the LO leakage and IQ imbalance in the RF ATX can be calibrated by the calibrated RxBB. Finally, the calibrated RF ATX transmit signal can be input from the LNA in the receiving path to correct the RF LO leakage and RF frequency response of the RF Rx.
  • FIG20 is a schematic diagram of calibrating and compensating the transmission link using an auxiliary receiving circuit in an embodiment of the present disclosure.
  • the transmission path may include a phase shift module PS, an amplifier PA, a power detector PD, etc.
  • the transmission path may adopt the transmission link of the digital phase shifter architecture (Digital Phase Shifter) described in any embodiment of the present disclosure.
  • Digital Phase Shifter Digital Phase Shifter
  • the transmission path adopts a digital phase shifter architecture, while achieving more accurate phase shifting operations, the transmission channel can simultaneously support multiple modes such as DDM and FDM (Frequency Division Multiplexing) of multiple antennas, and can also save the calibration operation of the RF phase shifter (Phase Shifter), reduce the isolation and coupling in the phase shifting system, and reduce link loss and production costs.
  • the transmit path of the digital phase shifter architecture can also support RF frequency response compensation, IQ imbalance and LO leakage calibration operations in the digital domain.
  • ARX may include a mixer, TIA, LPF, HPF, IQ ADC, adder and RF calibration module (RF Calib) connected in sequence, that is, by receiving the ARX IQ LO signal at one input end of the mixer, and the other input end is connected to the node before the transmission path PD along the signal transmission direction (i.e., the direction of the arrow shown in the figure), or to any node after the phase shifter (module).
  • the output end of the PA (synchronously realizing the calibration of the PA), the input end of the PA, etc., so as to perform calibration operations on the transmission path through the ARX.
  • a corresponding calibration circuit i.e., calibration receiving unit
  • ARX i.e., auxiliary receiving unit
  • RF single-tone signal generating circuit RF Tone Generator
  • BPF Band-pass Filter
  • the adder may be configured to calibrate and compensate for the TX LO leakage (TX LO leakage Waveform)
  • the multiplier may be configured to compensate for the RF Tone Gen LO leakage (leakage)
  • the BPF may be configured to filter out the DC signal generated by the LO leakage of the RF Tone Generator. That is, the RF Tone Generator may be configured to generate multiple stable single-tone signals of different frequencies to achieve the calibration operation of the ARX.
  • the ARX may be calibrated using an RF Tone Generator first, and then the calibrated ARX may be used to calibrate a transmitting channel (Transmitter) including a PA, for example, calibration of the PD at the PA output, the phase shifter in the transmitting channel, the total gain from the DAC to the PA output, and the frequency response and other devices and circuits.
  • a transmitting channel Transmitter
  • PA for example, calibration of the PD at the PA output, the phase shifter in the transmitting channel, the total gain from the DAC to the PA output, and the frequency response and other devices and circuits.
  • the RF Tone Generator can be used to generate multiple stable single-tone signals of different frequencies to assist in calibrating the ARX, and then the calibrated ARX can be used to assist in calibrating the IQ imbalance, local oscillator leakage, inconsistent frequency response and other issues of the transmit path TX.
  • Figure 21 is a schematic diagram of the structure of an auxiliary circuit in an embodiment of the present disclosure
  • Figure 22 is a schematic diagram of the structure of another auxiliary circuit in an embodiment of the present disclosure
  • Figure 23 is a schematic diagram of the circuit module of the IQ Mixer in an embodiment of the present disclosure
  • Figure 24 is a schematic diagram of the structure of an IQ Mixer in an embodiment of the present disclosure
  • Figure 25 is a schematic diagram corresponding to the structure shown in Figure 24.
  • ATX can also be implemented by combining a squarer with a DAC of multiple bits.
  • the receiving path includes an LNA, a mixer, a TIA, a HPF, an ADC and a BB processing module (BB Processor) connected in sequence, and a local oscillator source LO, that is, the mixer uses the first LO signal provided by the local oscillator source to down-convert the echo signal provided by the LNA to obtain an intermediate frequency signal (i.e., an analog baseband signal).
  • BB Processor BB Processor
  • ATX may include a first DAC (i.e., DAC1), a mixer and a squarer (x ⁇ 2) connected in sequence, wherein the first DAC provides an analog signal to the mixer receiving end in the ATX, and the other receiving end of the mixer can be connected to the local oscillator source in the receiving path to mix the analog echo signal output by the first DAC using the second LO signal provided by the local oscillator source, and then output an analog intermediate frequency signal to the input end of the squarer, and the squarer connects the processed analog intermediate frequency signal to the input end of the TIA to calibrate the circuits and components such as the TIA, HPF, and ADC in the receiving path.
  • DAC1 i.e., DAC1
  • a mixer i.e., a mixer and a squarer (x ⁇ 2) connected in sequence
  • the first DAC provides an analog signal to the mixer receiving end in the ATX
  • the other receiving end of the mixer can be connected to the local oscillator source in the receiving path to mix the analog
  • the ATX can also output an analog echo signal through a DAC directly to the receiving end of the squarer, and the squarer sends the processed analog echo signal to the input end of the LNA to calibrate the LNA, mixer, TIA, HPF, ADC and other circuits and components in the receiving path.
  • a second DAC may be provided for an ADC including a VGA, an SDM unit and a decimation filter in a receiving path, and the output of the second DAC may be connected to any node between the mixer and the TIA, between the TIA and the HPF, between the HPF and the VGA, and between the VGA and the ADC in the receiving path to calibrate the corresponding circuits and components.
  • the first DAC may be a multi-bit (e.g., 10-bit) DAC
  • the second DAC may be a 1-bit DAC.
  • the ATX when an ATX is set near a receiving path, the ATX can be set by utilizing the gaps between the receiving paths, and two or more receiving paths can share one ATX.
  • the ARX when an ARX is set near a transmitting path, the ARX can be set by utilizing the gaps between the transmitting paths, and two or more transmitting paths can share one ARX.
  • the above-mentioned ATX and/or ARX can be operated intermittently.
  • the ATX can transmit a calibration tone signal during the intervals between the working of the receiving path (such as between frames or chirps) to effectively calibrate and compensate the receiver in real time.
  • the ARX can also generate a calibration tone signal using a tone generator during the intervals between the working of the receiver (such as between frames or chirps), and effectively calibrate the ARX first.
  • the transmitter is then effectively calibrated using the calibrated ARX to ensure that calibration and compensation operations can be performed in a preset manner.
  • auxiliary circuits are added to integrated circuits such as chips (chip or die) to assist in calibrating the main path circuits, so as to effectively improve the performance of analog circuits and modules.
  • integrated circuits such as chips (chip or die)
  • at least part of the RF circuits and devices can be calibrated in real time on the fly, thereby effectively improving the calibration performance, thereby effectively improving the RF performance while reducing the difficulty of RF implementation.
  • the mixer Since the mixer is a very important and key device for frequency conversion in the transceiver link, it is widely used in radio devices such as communications and radars, such as a single-sideband mixer with good suppression effect on image signals, and an IQ mixer in the transceiver link in the embodiment of the present disclosure.
  • the IQ mixer (such as a single-sideband mixer) includes two branches, an I branch and a Q branch, with a phase difference of 90° to transmit signals.
  • the physical distance between the I branch and the Q branch, and between the input branch and the output branch before mixing are short.
  • electromagnetic wave signals may be leaked between branches, between input and output ports, and between the IQ matching network and the mixer output through methods such as magnetic coupling, substrate coupling, and electrical coupling, which may cause signal leakage problems, thereby causing the image suppression ratio and local oscillator leakage of the mixer to deteriorate seriously.
  • the present disclosure provides a new mixer structure, that is, by improving the passive network of the mixer output, the isolation between the two IQ branches is effectively improved while the leakage of the local oscillator signal is effectively reduced.
  • FIG23 is a schematic diagram of a circuit module of an IQ Mixer in an embodiment of the present disclosure.
  • an IQ mixer may include an I-branch mixing unit, a Q-branch mixing unit, and a transformer unit, wherein the I-branch mixing unit may be configured to output an I-channel signal, the Q-branch mixing unit may be configured to output a Q-channel signal, and the transformer unit may be configured to magnetically couple the I-channel signal and the Q-channel signal to synthesize an IQ mixing output signal, and transmit the IQ mixing output signal to a subsequent circuit (Next Block).
  • the transformer unit may be arranged between the I-branch mixing unit and the Q-branch mixing unit, so as to increase the physical distance between the I-branch mixing unit and the Q-branch mixing unit relative to a short-circuited IQ mixer, thereby effectively reducing the coupling between the I-branch mixing unit and the Q-branch mixing unit, thereby achieving the purpose of improving the IQ mixer rejection ratio, and also facilitating the layout design of the board.
  • the short-circuited IQ mixer is formed by short-circuiting the outputs of the two branches of the mixer IQ respectively, and then matching them to the subsequent circuit through a matching network (Matching Networks), that is, performing signal synthesis by electrical coupling.
  • a branch inductor can be connected in series between the output ends of the I branch mixing unit and the output ends of the Q branch mixing unit respectively, and then a magnetic coupling inductor is set between the two branch inductors to form a three-turn transformer structure, that is, the output of the I branch mixing unit and the output of the Q branch mixing unit are magnetically coupled and synthesized by magnetic coupling, thereby obtaining the above-mentioned IQ mixing output signal; wherein the common modes of the I branch mixing unit and the Q branch mixing unit are not directly connected, so they can be adjusted separately, making the application scenario of the device more flexible; then, the two ends of the above-mentioned magnetic coupling inductor are used
  • the LO signal and the signal to be mixed (such as the echo signal) enter the P and N terminals of the IQ branches of the three-turn transformer, the current direction and magnetic field direction formed on the three-turn transformer are shown in FIG24.
  • the Isb magnetic field is superimposed and the usb magnetic field is offset, and a high-quality single-sideband signal can be output. That is, based on the three-turn transformer structure, down-mixing operation or up-mixing operation of the signal to be mixed can be realized.
  • the output of the three-turn transformer is still a single-sideband signal.
  • the magnetically coupled IQ mixer provided in the embodiment of the present disclosure performs power synthesis by magnetic coupling, compared with the short-circuited IQ mixer that performs power synthesis by electrical coupling.
  • the magnetically coupled IQ mixer can be completely consistent and can be axially symmetrical, and will not introduce additional phase errors due to the inconsistent wiring lengths of the I and Q branches, thereby making the related circuits (such as high-frequency circuits such as millimeter waves) more robust in subsequent manufacturing processes and more resistant to process instability.
  • the magnetically coupled IQ mixer in the embodiment of the present disclosure as shown in FIG.
  • the transformer tap on the midline between the P and N ports can be set, so that the structure of the mixer can be axially symmetrical based on the midline, so that the common-mode path lengths of the P port and the N port are equal, that is, the symmetry of PN in the magnetically coupled IQ mixer is better than the symmetry of PN in the short-circuited IQ mixer, thereby making the leakage of the local oscillator signal smaller.
  • the magnetically coupled IQ mixer provided in the embodiments of the present disclosure may be applied to various circuits such as a transmission channel and an on-chip self-test, as an up-conversion mixer in various electromagnetic wave circuits, etc.
  • the IQ mixers in various transmission channels (or transmission links) and BIST in the embodiments of the present disclosure may adopt the magnetically coupled IQ mixer in the embodiments of the present disclosure.
  • FIG26 is a schematic diagram of the physical structure of another IQ Mixer in an embodiment of the present disclosure.
  • an embodiment of the present disclosure also provides another structure of a magnetically coupled IQ mixer, that is, there is a partial overlap in the tap feed lines of the I and Q branches.
  • the tap of the I branch can bypass the Q branch side (such as through a through hole via) and extend to the I branch side, and then connect to the common-mode bias voltage.
  • the tap of the Q branch can bypass the I branch side (such as through a through hole via) and extend to the Q branch side, and then connect to the common-mode bias voltage.
  • the above-mentioned routing method can make the path of the common-mode path more certain and less susceptible to interference from other circuits.
  • the three-turn transformer in the IQ Mixer in the embodiment of the present disclosure can be square, octagonal, regular octagonal, etc., that is, as long as it can be axially symmetrically distributed along the center between NPs in a top view or a bottom view.
  • an embodiment of the present disclosure also provides a signal transmission method, which is applied to an electromagnetic wave device having at least one signal transmission link, and the signal transmission method includes: step 2701, determining the phase of the RF transmission signal of each signal transmission link; step 2702, determining the initial phase of the baseband signal in each of the signal transmission links according to the phase of the RF transmission signal; step 2703, generating an initial baseband signal according to the determined initial phase; step 2704, compensating the initial baseband signal using pre-acquired compensation information to obtain the baseband signal; step 2705, performing a phase shift operation on the transmitting end local oscillator signal based on the baseband signal to obtain the RF transmission signal.
  • the signal transmission method of the embodiment of the present disclosure generates a digital baseband signal in the digital domain by using a digital phase shifter architecture, which has better orthogonality and lower sidelobes.
  • the phase shift phase can be generated very accurately, so that the phase modulation accuracy is higher, thereby realizing a vehicle-mounted radar system with a high-precision digital phase shifting function, reducing the isolation requirements between antennas, and having the advantages of small link loss, low cost, and no need for offline calibration. It can also support more flexible wave transmission schemes, such as high-performance Doppler division multiplexing and frequency division multiplexing, and can support frequency response compensation in the digital domain.
  • an embodiment of the present disclosure also provides a signal transmission method, which is applied to an electromagnetic wave device having at least one signal transmission link, and the signal transmission method includes: step 2801, obtaining observation information of the main signal transmission path under the current compensation coefficient, wherein the main signal transmission path is used to generate a radio frequency transmission signal after performing a compensation operation on the generated signal according to the compensation coefficient, so as to achieve target detection and/or communication.
  • Step 2802 when the current observation information meets the iteration condition, the current compensation coefficient is used as the compensation coefficient used for the compensation operation of the signal transmission link; otherwise, the current compensation coefficient is iterated until the obtained observation information meets the iteration condition.
  • Step 2803 using the compensation coefficient to compensate the baseband signal.
  • Step 2804 performing a phase shift operation on the transmitting end local oscillator signal based on the compensated baseband signal to obtain a radio frequency transmission signal.
  • the signal transmission method of the embodiment of the present disclosure, the signal calibration link, the transmission link and method, the transceiver link, the integrated circuit of the embodiment of the present disclosure, the compensation information generated by the signal calibration link can compensate the generated signal, solve the problems of orthogonal imbalance, LO leakage and harmonic distortion, thereby effectively improving the signal quality in the main path of signal transmission.
  • the compensation coefficient required for the main path of signal transmission is determined by iterative method, which can effectively indicate the accuracy of the function compensation coefficient.
  • the embodiment of the present disclosure also provides a signal transmission method, which is applied to an electromagnetic wave device having at least one signal transmission link.
  • the signal transmission method includes: step 2901, determining the initial observation information O(0), the first observation information O(1) and the second observation information O(2) corresponding to the main signal transmission path under the conditions of different values of the initial compensation coefficient h(0), the first compensation coefficient h(1) and the second compensation coefficient h(2), wherein the main signal transmission path is used to generate a radio frequency transmission signal after compensating the generated signal according to the compensation coefficient, so as to achieve target detection and/or communication.
  • Step 2902 using the initial observation information O(0), the first observation information O(1) and the second observation information O(2), Determine a third compensation coefficient h(3).
  • Step 2903 Compensate the baseband signal using the compensation coefficient.
  • Step 2904 Perform a phase shift operation on the transmitting end local oscillator signal based on the compensated baseband signal to obtain a radio frequency transmission signal.
  • the signal transmission method of the embodiment of the present disclosure, the signal calibration link, the transmission link and method, the transceiver link, the integrated circuit of the embodiment of the present disclosure, the compensation information generated by the signal calibration link can compensate the generated signal, solve the problems of orthogonal imbalance, LO leakage and harmonic distortion, thereby effectively improving the signal quality in the main path of signal transmission. Observation information is obtained through multiple calibration operations to determine the compensation coefficient required for the main path of signal transmission, which can effectively indicate the accuracy of the function compensation coefficient.
  • the transmitting end digital baseband signal is a single tone signal, and the transmitting end local oscillator signal is a swept frequency signal; or, the transmitting end digital baseband signal is a swept frequency signal, and the transmitting end local oscillator signal is a single tone signal.
  • the frequency bandwidth of the swept frequency signal is above 2 GHz.
  • the embodiment of the present disclosure also provides an integrated circuit, which may include a radio frequency module, an analog signal processing module and a digital signal processing module connected in sequence; the radio frequency module is used to generate a radio frequency transmission signal and receive a radio frequency reception signal; the analog signal processing module is used to down-convert the radio frequency reception signal to obtain an intermediate frequency signal; the digital signal processing module is used to perform analog-to-digital conversion on the intermediate frequency signal to obtain a digital signal; and the radio frequency module includes any of the above-mentioned signal transmission links, the above-mentioned signal transceiver links, any of the above-mentioned signal calibration links, any of the above-mentioned signal compensation links, any of the above-mentioned signal calibration systems, and/or the above-mentioned IQ mixers; and/or the digital signal processing module performs compensation in the digital domain based on the above-mentioned feeder unequal length compensation method; wherein the radio frequency reception signal is an echo signal formed by the radio frequency transmission signal being emitted and
  • the integrated circuit may be an AiP (Antenna-In-Package) chip structure, an AoP (Antenna-On-Package) chip structure, an AoC (Antenna-On-Chip) chip structure or a RoP (Radiator on Package, i.e., a radiating structure (Radiator) is set on the package (Package), and a waveguide structure is formed around the Radiator with a sphere, the RF signal is transitioned to the waveguide structure through the radiating structure, and then converted to an external antenna by the waveguide structure) structure.
  • AiP Antenna-In-Package
  • AoP Antenna-On-Package
  • AoC Antenna-On-Chip
  • an electromagnetic wave sensor may include an antenna, and an integrated circuit as described above.
  • the integrated circuit is electrically connected to the antenna for receiving and transmitting electromagnetic wave signals.
  • the electromagnetic wave sensor may include: a carrier, an integrated circuit and an antenna as described in any of the above embodiments, and the integrated circuit may be arranged on the carrier; the antenna may be arranged on the carrier, or integrated with the integrated circuit as a device arranged on the carrier (that is, the antenna may be an antenna provided in an AiP, AoP or AoC structure at this time); wherein the integrated circuit is connected to the antenna (that is, the sensor chip or integrated circuit is not integrated with an antenna at this time, such as a conventional SoC, etc.), for receiving and transmitting electromagnetic wave signals.
  • the carrier may be a printed circuit board PCB, and the first transmission line may be a PCB trace.
  • An embodiment of the present disclosure provides a device, which may include: a device body; and an electromagnetic wave sensor as described above, which is arranged on the device body; wherein the electromagnetic wave sensor is used for target detection and/or communication to provide reference information for the operation of the device body.
  • the present disclosure also provides an electronic device, which can be expressed in the form of a general-purpose computing device.
  • the components of the electronic device may include, but are not limited to: at least one processing unit, at least one storage unit, a bus connecting different system components (including a storage unit and a processing unit), a display unit, etc.
  • the storage unit stores a program code, and the program code can be executed by the processing unit so that the processing unit executes the method described in this specification according to various exemplary embodiments of the present disclosure.
  • the storage unit may include a readable medium in the form of a volatile storage unit, such as a random access memory unit (RAM) and/or a cache memory unit, and may further include a read-only memory unit (ROM).
  • RAM random access memory unit
  • ROM read-only memory unit
  • the storage unit may also include a program/utility having a set (at least one) of program modules, such program modules including but not limited to: an operating system, one or more application programs, other program modules, and program data, each of which or some combination may include an implementation of a network environment.
  • program modules including but not limited to: an operating system, one or more application programs, other program modules, and program data, each of which or some combination may include an implementation of a network environment.
  • the bus may represent one or more of several types of bus structures, including a memory unit bus or memory unit controller, a peripheral bus, an accelerated graphics port, a processing unit, or a local bus using any of a variety of bus architectures.
  • the electronic device may also communicate with one or more external devices (e.g., keyboards, pointing devices, Bluetooth devices, etc.), may communicate with one or more devices that enable a user to interact with the electronic device, and/or may communicate with any device that enables the electronic device to communicate with one or more other computing devices (e.g., routers, modems, etc.). This communication may be performed through an input/output (I/O) interface.
  • the electronic device may also communicate with one or more networks (e.g., local area networks (LANs), wide area networks (WANs), and/or public networks, such as the Internet) through a network adapter.
  • the network adapter may communicate with other modules of the electronic device through a bus.
  • the electronic device in the embodiments of the present disclosure may also include: a device body; and an electromagnetic wave sensor as described in any of the above embodiments arranged on the device body; wherein the electromagnetic wave sensor can be used to realize functions such as target detection and/or wireless communication.
  • the electromagnetic wave sensor can be arranged outside the device body, or arranged inside the device body, and in other optional embodiments of the present disclosure, the electromagnetic wave sensor can also be arranged partially inside the device body and partially outside the device body.
  • the present disclosure embodiment does not limit this, and it can be determined according to the specific situation.
  • the device body can be a component or product used in fields such as smart cities, smart homes, transportation, smart homes, consumer electronics, security monitoring, industrial automation, in-cabin detection (such as smart cockpits), medical devices, and health care.
  • the device body can be intelligent transportation equipment (such as cars, bicycles, motorcycles, ships, subways, trains, etc.), security equipment (such as cameras), liquid level/flow rate detection equipment, smart wearable devices (such as bracelets, glasses, etc.), smart home devices (such as sweeping robots, door locks, televisions, air conditioners, smart lights, etc.), various communication devices (such as mobile phones, tablet computers, etc.), and gates, smart traffic lights, smart signs, traffic cameras, and various industrialized robotic arms (or robots), etc. It can also be various instruments for detecting life characteristic parameters and various devices equipped with the instruments, such as life characteristic detection in car cabins, indoor personnel monitoring, smart medical equipment, consumer electronic equipment, etc.
  • the embodiment of the present disclosure also provides a non-transitory computer-readable storage medium having computer-readable instructions stored thereon.
  • the processor executes the feeder unequal length compensation method as described above.
  • the technical solution according to the implementation methods of the present disclosure can be embodied in the form of a software product, which can be stored in a non-volatile storage medium (which can be a CD-ROM, a USB flash drive, a mobile hard disk, etc.) or on a network, including several instructions to enable a computing device (which can be a personal computer, a server, or a network device, etc.) to execute the above method according to the implementation methods of the present disclosure.
  • a non-volatile storage medium which can be a CD-ROM, a USB flash drive, a mobile hard disk, etc.
  • a computing device which can be a personal computer, a server, or a network device, etc.
  • the software product may use any combination of one or more readable media.
  • the readable medium may be a readable signal medium or a readable storage medium.
  • the readable storage medium may be, for example, but not limited to, a system, device or device of electricity, magnetism, light, electromagnetic, infrared, or semiconductor, or any combination thereof. More specific examples (non-exhaustive list) of readable storage media include: an electrical connection with one or more wires, a portable disk, a hard disk, a random access memory (RAM), a read-only memory (ROM), an erasable programmable read-only memory (EPROM or flash memory), an optical fiber, a portable compact disk read-only memory (CD-ROM), an optical storage device, a magnetic storage device, or any suitable combination thereof.
  • Computer readable storage media may include data signals propagated in baseband or as part of a carrier wave, which carry readable program code. Such propagated data signals may take many forms, including but not limited to electromagnetic signals, optical signals, or any suitable combination of the above.
  • the readable storage medium may also be any readable medium other than a readable storage medium.
  • the readable medium can send, propagate or transmit a program for use by or in conjunction with an instruction execution system, device or device.
  • the program code contained on the readable storage medium can be transmitted using any suitable medium, including but not limited to wireless, wired, optical cable, RF, etc., or any suitable combination of the above.
  • Program code for performing the operations of the present disclosure may be written in any combination of one or more programming languages, including object-oriented programming languages such as Java, C++, and the like, as well as conventional procedural programming languages such as "C" or similar programming languages.
  • the program code may be executed entirely on the user computing device, partially on the user device, as a separate software package, partially on the user computing device and partially on a remote computing device, or entirely on a remote computing device or server.
  • the remote computing device may be connected to the user computing device through any type of network, including a local area network (LAN) or a wide area network (WAN), or may be connected to an external computing device (e.g., via the Internet using an Internet service provider).
  • LAN local area network
  • WAN wide area network
  • Internet service provider e.g., via the Internet using an Internet service provider
  • the computer-readable medium carries one or more programs. When the one or more programs are executed by a device, the computer-readable medium implements the aforementioned functions.
  • modules can be distributed in the device according to the description of the embodiment, or can be changed accordingly and only used in one or more devices different from the embodiment.
  • the modules of the above embodiments can be combined into one module, or further divided into multiple sub-modules.
  • a computer program including a computer program or an instruction, which, when executed by a processor, can execute the method described above.
  • the above-mentioned integrated circuit can be a millimeter wave radar chip.
  • the type of digital function module in the integrated circuit can be determined according to actual needs.
  • a data processing module can be used for functions such as distance-dimensional Doppler transform, velocity-dimensional Doppler transform, constant false alarm detection, direction of arrival detection, point cloud processing, etc., to obtain information such as the distance, horizontal angle, pitch angle, speed, height, micro-Doppler motion characteristics, shape, size, surface roughness and dielectric properties of the target.
  • radio devices can achieve functions such as target detection and/or communication by transmitting and receiving radio signals to provide the device body with detection target information and/or communication information, thereby assisting or even controlling the operation of the device body.
  • the radio device such as millimeter-wave radar
  • the radio device can assist the ADAS system to realize application scenarios such as adaptive cruise control, automatic braking assistance (i.e., AEB), blind spot detection warning (i.e., BSD), assisted lane change warning (i.e., LCA), reversing assistance warning (i.e., RCTA), parking assistance, rear vehicle warning, collision avoidance, pedestrian detection, etc.
  • AEB automatic braking assistance
  • BSD blind spot detection warning
  • assisted lane change warning i.e., LCA
  • reversing assistance warning i.e., RCTA

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  • Computer Networks & Wireless Communication (AREA)
  • Physics & Mathematics (AREA)
  • Radar, Positioning & Navigation (AREA)
  • Remote Sensing (AREA)
  • General Physics & Mathematics (AREA)
  • Signal Processing (AREA)
  • Electromagnetism (AREA)
  • Transmitters (AREA)
  • Radar Systems Or Details Thereof (AREA)
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Abstract

本公开涉及电磁波传感器技术领域，具体涉及一种信号发射、校准、补偿及收发链路，集成电路、电磁波传感器及设备，所述发射链路包括模拟信号源和数字移相器，所述模拟信号源可被配置为用于提供初始模拟信号，所述数字移相器可被配置为用于在数字域产生移相信号，并基于该移相信号对所述初始模拟信号进行移相，以将所述初始模拟信号进行预设的移相操作，以有效的提升调相精度和调相准确度，同时还能避免对发射链路中的移相器等链路及器件进行离线校准的操作，进而降低了工程实现的复杂度及难度。另外，还能有效减小移相架构的发射链路面积和损耗，提升系统的稳定性，降低通道耦合度。

Description

信号发射、校准、补偿及收发链路，集成电路、电磁波传感器及设备
本申请要求于2023年6月14日提交中国专利局、申请号为202310702586.5、发明名称为“信号发射、校准、补偿及收发链路，IQ混频器、集成电路、传感器及设备”的中国专利申请的优先权以及于2024年5月14日提交中国专利局、申请号为202410599441.1、发明名称为“信号发射链路及方法、收发链路、集成电路及电磁波器件”的中国专利申请的优先权，其内容应理解为通过引用的方式并入本公开中。
技术领域
本公开实施例涉及但不限于电磁波传感器技术领域，具体涉及一种信号发射、校准、补偿及收发链路，集成电路、电磁波传感器及设备。
背景技术
由于采用模拟移相器架构的信号发射链路，其存在调相精度(Resolution)、调相准确度(Accuracy)等均较低的问题，进而使得其无法满足高精度及准确性要求的各种需求。
发明内容
为了解决上述技术问题，本公开实施例提供了一种信号发射、校准、补偿及收发链路，IQ混频器、集成电路、电磁波传感器及设备等，基于数字移相器架构形成信号收发链路，以及对应的校准及补偿链路等技术，以有效的提升调相精度(Resolution)和调相准确度(Accuracy)，同时还能避免对发射链路中的移相器等链路及器件进行离线(off-line)校准的操作，进而降低了工程实现的复杂度及难度。另外，还能有效减小移相架构的发射链路面积和损耗，提升系统的稳定性，降低通道耦合度。
本公开实施例提供了一种信号发射链路，应用于电磁波传感器中，所述发射链路包括模拟信号源和数字移相器，所述模拟信号源可被配置为用于提供初始模拟信号，所述数字移相器可被配置为用于提供在数字域产生移相信号，并基于该移相信号对所述初始模拟信号进行移相，以将所述初始模拟信号进行预设的移相操作。
示例性的，信号发射链路还包括发射天线，所述发射天线可被配置用于将移相后的初始模拟信号向预设空间区域辐射。
示例性的，所述数字移相器包括数字移相信号源、数模转换器和混频器，所述数字移相器可被配置用于产生数字移相信号，所述数模转换器可被配置用于将接收的所述数字移相信号转换为模拟移相信号，所述混频器可被配置用于利用所接收的所述模拟移相信号对所接收的所述初始模拟信号进行混频操作，以将所述初始模拟信号进行预设的移相操作。
示例性的，所述数字移相信号源包括直接数字频率合成器，所述数模转换器为IQ数模转换器，所述混频器为IQ混频器。
示例性的，所述数字移相信号为单音信号，所述初始模拟信号为扫频信号；或者，所述数字移相信号为扫频信号，所述初始模拟信号为单音信号。
示例性的，所述信号发射链路发射的是调频连续波信号。
本公开实施例还提供了一种信号发射链路，包括集成于同一集成电路中的信号发射主通路和信号校准链路；其中：所述信号校准链路被配置为用于对所述信号发射主通路进行校准以获取补偿信息；以及所述信号发射主通路被配置为用于根据所述补偿信息进行补偿操作后生成射频发射信号，以用于实现目标探测和/或通信。
示例性的，所述补偿信息包括谐波失真补偿参数、本振泄漏补偿参数和正交失衡补偿参数中的至少一个。
示例性的，所述信号发射主通路包括第一信号源和移相器；其中，所述第一信号源被配置为生成第一模拟信号；以及所述移相器被配置为对所述第一模拟信号进行频率搬移和/或移相，以形成射频发射信号。
示例性的，在所述移相器为非正交架构时，所述移相器包括第二信号源和发端混频器，其中所述第二信号源被配置为用于生成第二模拟信号，所述发端混频器被配置为对所述第一模拟信号和所述第二模拟信号进行混频处理，以形成所述射频发射信号；
在所述移相器为正交架构时，其中所述移相器包括：第二信号源、数模转换模块和发端混频器；其中所述第二信号源被配置为生成第一数字信号；所述数模转换模块被配置为将所述第一数字信号转换为第二模拟信号；以及所述发端混频器被配置为基于所述第二模拟信号对所述第一模拟信号进行频率搬移和/或移相，以形成所述射频发射信号。
示例性的，所述发射主通路还包括补偿电路，其中所述补偿电路的信号输入端与所述第二信号源相连，信号输入端与所述移相器相连，所述补偿电路用于将补偿信号和第二信号源输出的信号进行合并处理后再输出。
示例性的，在所述信号发射主通路输出的信号为两路正交信号时，所述补偿电路所使用的补偿信号为两路正交信号；在所述信号发射主通路输出的信号不是两路正交信号时，所述补偿电路所使用的补偿信号为一路与第二信号源输出的信号类型相同的信号，其中信号类型为数字信号或模拟信号。
示例性的，所述补偿电路包括补偿信号产生器和加法器，其中：所述补偿信号产生器，被配置为生成所述补偿信号；所述加法器，与所述补偿信号产生器和所述第二信号源相连，用于将所述第二信号源输出的信号与所述补偿信号产生器输出的补偿信号进行信号叠加操作。
示例性的，在所述信号发射主通路输出的信号为两路正交信号时，所述补偿信号产生器包括谐波补偿信号单元、正交失衡补偿信号单元和本振泄露补偿信号单元中的至少一个；在所述信号发射主通路输出的信号不是两路正交信号时，所述补偿电路包括谐波补偿信号单元和本振泄露补偿信号单元中的至少一个；其中，所述谐波补偿信号单元产生的补偿信号用于消除所述信号发射主通路中主频信号的谐波信号；所述本振泄露补偿信号单元的补偿信号用于补偿所述信号发射主通路中发端本振信号产生的泄露信号；所述正交失衡补偿信号单元产生的补偿信号用于补偿所述信号发射主通路中主频信号的镜像信号。
示例性的，所述谐波补偿信号单元产生的补偿信号与所述信号发射主通路中谐波信号的频率相同、振幅相同、相位相反。
示例性的，所述谐波补偿信号单元包括n次方模块或n倍频信号发生器；n次方模块，其中所述n次方模块的信号输入端与所述第二信号源的信号输出端相连，用于利用第二信号源输出的信号，生成频率为所述第二信号源输出的信号的频率的n倍的信号，得到所述补偿信号；n倍频信号发生器，用于产生频率为所述第二信号源输出的信号的频率的n倍的信号，得到所述补偿信号，其中n的取值为正整数。示例性的，n的取值为奇数。示例性的，n的取值为3。
示例性的，所述本振泄露补偿信号单元产生的补偿信号是基于移相器所使用的第一模拟信号对应的泄漏信号生成的。
示例性的，所述本振泄露补偿信号单元产生的补偿信号与所述泄露信号的频率相同、振幅相同但相位相反。
示例性的，所述正交失衡补偿电路所使用的补偿信号在所述信号发射主通路对应的镜像信号，与所述信号发射主通路产生的期望信号对应的镜像信号之间频率相同、振幅相同、相位相反。
示例性的，所述正交失衡补偿电路所使用的补偿信号是根据第二信号源输出的信号以及所述第二信号源输出的信号的频率反转的复共轭信号确定的。
示例性的，所述正交失衡补偿电路产生的补偿信号的获取方式包括：获取预设的预补偿系数与所述复共轭信号的乘积，得到所述复共轭信号对应的调整信号；计算所述第二信号源输出的信号与所述调整信号之间的差值，得到所述正交失衡补偿电路产生的补偿信号。
示例性的，所述预补偿系数是基于期望信号的振幅与所述期望信号对应的镜像信号的振幅的比值确定的。
示例性的，所述信号校准链路包括所述信号发射主通路对应的信号接收链路，其中所述信号接收链路用于对FMCW射频发射信号对应的回波信号进行接收处理；其中：所述信号接收链路，被配置为从所述信号发射主通路获取采样信号，并根据所述采样信号得到补偿信号的配置信息；其中，所述信号发射主通路使用的发端本振信号和所述信号接收链路中使用的收端本振信号的频率不同。
示例性的，所述信号校准链路还包括：频率调整电路，被配置为调整所述采样信号和收端本振信号中至少一个的频率。
示例性的，所述频率调整电路的信号输入端连接于所述信号发射主通路和发射天线之间，所述频率调整电路的信号输出端连接于所述信号接收链路和接收天线之间，其中所述频率调整电路被配置为调整所述信号发射主通路输出的射频发射信号的频率；对应的，所述信号接收链路，被配置为从所述频率调整电路获取采样信号，并根据所述采样信号得到补偿信号的配置信息；或者，所述频率调整电路，连接于所述信号接收链路中的收端混频器与收端本振器之间，被配置为调整收端本振信号的频率，其中所述收端本振器用于产生收端本振信号，所述收端混频器用于利用收端本振信号对接收的信号进行解调处理；所述信号接收链路，连接于所述信号发射主通路和发射天线之间，被配置为从所述信号发射主通路获取采样信号，并利用频率调整电路输出的信号对所述采样信号进行处理，得到补偿信号的配置信息。
示例性的，在所述信号发射主通路传输的射频发射信号为两路正交信号时，所述信号校准链路为支持处理包括两路正交信号的回波信号的信号接收链路。
示例性的，在所述信号发射主通路传输的射频发射信号为两路正交信号时，所述信号校准链路设置有正交处理电路以及两路信号接收链路，其中每条信号接收链路均不支持处理正交的回波信号；其中所述正交处理电路，被配置为对接收的正交信号进行射频处理，得到两路信号，并将两路信号分别发送到两路信号接收链路。
示例性的，所述信号校准链路，所述信号校准链路的第一输入端连接于所述信号发射主通路中发端混频器内电压电流转换器与电流开关之间，第二输入端所述信号发射主通路与发射天线之间，信号输出端与所述信号发射主通路中的补偿电路相连，用于从所述第一输入端和所述第二输入端中的至少一个获取所述信号发射主通路中的信号，并根据获取的信号确定补偿信息。
示例性的，所述信号校准链路包括校准解调器、多路选择器和校准模块；其中：所述校准解调器，被配置为从所述第二输入端获取信号发射主通路中的信号，并进行解调处理；所述多路选择器具有两个信号输入端和一个信号输入端，其中一个信号输入端连接于发端混频器内电压电流转换器与电流开关之间，另一个信号输入端连接于校准解调器的信号输入端，用于输出两个信号输入端中一个对应的信号；校准模块，被配置为根据所述多路选择器输出的信号确定所述补偿信息。
示例性的，在所述信号发射主通路传输的射频发射信号为两路正交信号时，所述信号校准链路传输的采集信号为两路正交信号。
示例性的，所述信号校准链路传输的采集信号为两路正交信号，所述信号校准链路还包括：模数转换器，连接在信号采集电路和校准模块之间，被配置为对所述信号采集电路输出的采集信号进行模数转换处理。
本公开实施例还提供了一种信号收发链路，包括如本公开任一实施例所述的信号发射链路，以及信号接收链路；所述信号接收链路包括收端混频器、模数转换器和数字信号处理模块；其中，所述收端混频器可被配置为用于基于所接收的收端本振信号对所接收的回波信号进行下降频以得到模拟中频信号，所述模数转换器可被配置为用于对所接收的所述中频信号进行模数转换以得到数字中频信号，所述数字信号处理模块可被配置为用于对所数字中频信号进行处理以得到目标参数；所述回波信号为所述信号发射链路所发射信号被目标物反射和/或散射而形成的信号。
示例性的，所述收端混频器为实数混频器，所述模数转换器为实数模数转换器；或者，所述收端混频器为正交混频器，所述模数转换器为正交模数转换器。
示例性的，所述收端本振信号为扫频信号；或者，所述收端本振信号为单音信号。
本公开实施例还提供了一种信号校准链路，包括如本公开任一实施例所述的信号收发链路；所述信号接收链路的接收天线连接端口连接至所述信号发射链路的发射天线连接端口，所述信号接收链路可被配置为用于对所述信号发射链路进行校准。
示例性的所述信号接收链路的本振信号与所述信号发射链路的本振信号之间具有预设的差频。
示例性的，所述信号校准链路还包括BIST模块，所述BIST模块设置在本振信号源与所述收端混频器之间；其中，所述BIST模板可被配置为基于预设的频偏信号对接收的本振信号进行混频，以使得所述收端混频器接收到的本振信号与所述信号发射链路的本振信号之间具有预设的差频。
本公开实施例还提供了一种信号校准链路，包括如本公开任一实施例所述的信号收发链路，以及BIST模块；所述信号接收链路的接收天线连接端口通过所述BIST模块连接至所述信号发射链路的发射天线连接端口，所述信号接收链路可被配置为用于对所述信号发射链路进行校准。
本公开实施例还提供了一种信号校准链路，包括两条信号接收链路、BIST模块、辅助电路单元和如本公开任一实施例所述的信号发射链路，以及；任一条所述信号接收链路均包括实数混频器、实数模数转换器和数字信号处理模块；所述实数混频器可被配置为用于基于所接收的本振信号对所接收的回波信号进行下降频以得到模拟中频信号，所述实数模数转换器可被配置为用于对所接收的所述中频信号进行模数转换以得到数字中频信号，所述数字信号处理模块可被配置为用于对所数字中频信号进行处理以得到目标参数；所述回波信号为所述信号发射链路所发射信号被目标物反射和/或散射而形成的信号；两条所述信号接收链路的接收天线连接端口依次通过所述辅助电路单元和所述BIST模块分别连接至所述信号发射链路的发射天线连接端口，所述信号接收链路可被配置为用于对所述信号发射链路的中频部分进行校准。
本公开实施例还提供了一种信号发射主通路的信号校准链路，其中所述信号发射主通路用于根据补偿系数对产生的信号进行补偿操作后生成射频发射信号，以用于实现目标探测和/或通信，其中：所述信号校准链路，被配置为用于获取当前补偿系数下信号发射主通路当前的观测信息；以及，在当前的观测信息满足迭代条件时，将当前的补偿系数作为所述信号发射链路补偿操作使用的补偿系数；否则，对当前的补偿系数进行迭代直到所得到的观测信息满足所述迭代条件为止。
示例性的，所述补偿系数包括谐波失真补偿参数、本振泄漏补偿参数和正交失衡补偿参数中的至少一个。
示例性的，所述信号发射主通路和所述信号校准链路集成于同一集成电路中。
示例性的，所述集成电路为毫米波雷达芯片，和/或，所述射频发射信号为FMCW信号。
示例性的，当前的补偿系数依次包括初始补偿系数h(0)、第一补偿系数h(1)、…、第k-1补偿系数h(k-1)以及第k补偿系数h(k)，其中第k补偿系数h(k)是根据第k-1补偿系数h(k-1)和第k-1观测信息O(k-1)确定的；其中，k为大于或等于2的整数。
示例性的，在初始观测信息O(0)和第一观测信息O(1)之间的差异值大于预设的差异门限时，根据第k-1补偿系数h(k-1)和第k-1观测信息O(k-1)确定第k补偿系数h(k)；在初始观测信息O(0)和第一观测信息O(1)之间的差异值小于预设的差异门限时，根据预设的相位调整量对初始观测信息O(0)进行调整，得到调整后的第一补偿系数h(1)，再根据调整后的第一补偿系数h(1)重新得到新的第一观测信息O(1)，以新的第一观测信息O(1)为基础，确定第k补偿系数h(k)；其中，初始观测信息O(0)与新的第一观测信息O(1)之间的差异值大于所述差异门限。
示例性的，所述根据第k-1补偿系数h(k-1)和第k-1观测信息O(k-1)确定第k补偿系数h(k)，包括：在初始观测信息O(0)的绝对值与第一观测信息O(1)的绝对值之间的差值大于预设的差值门限时，以第一补偿系数h(1)更新为初始观测量O(0)为基础，迭代计算第k-1补偿系数h(k-1)和k-1观测信息O(k-1)之间的总和，确定第k补偿系数h(k)；在第一观测信息O(1)的绝对值与初始观测信息O(0)的绝对值之间的差值大于预设的差值门限时，以第一补偿系数h(1)更新为初始观测量O(0)的相反数为基础，迭代计算第k-1补偿系数h(k-1)和k-1观测信息O(k-1)之间的差值，确定第k补偿系数h(k)。
示例性的，所述以新的第一观测信息O(1)为基础，确定第k补偿系数h(k)，包括：在初始观测信息O(0)的绝对值与新的第一观测信息O(1)的绝对值之间的差值大于所述差值门限时，迭代计算第k-1观测信息O(k-1)与经所述相位调整量处理第k-1观测信息O(k-1)后得到的调整后的第k-1观测信息O(k-1)之间的总和，确定第k补偿系数h(k)；在新的第一观测信息O(1)的绝对值与初始观测信息O(0)的绝对值之间的差值大于所述差值门限时，更新第一补偿系数h(1)为调整后的第一补偿系数h(1)的相反数，迭代计算第k-1补偿系数h(k-1)与经所述相位调整量处理第k-1观测信息O(k-1)后得到的调整后的第k-1观测信息O(k-1)之间的差值，确定第k补偿系数h(k)。
示例性的，所述第一补偿系数h(1)是根据初始观测信息O(0)确定的。
示例性的，所述初始补偿系数数h(0)的数值为0。
示例性的，所述信号发射主通路包括基带模块和射频模块，其中所述基带模块用于产生基带信号；所述射频模块用于对利用发端本振信号对所述基带信号进行频率搬移和/或移相，以形成射频发射信号；其中，所述观测信息是从所述射频模块获取的。
示例性的，所述射频模块包括发端混频器和功率放大器；其中所述发端混频器利用发端本振信号对所述基带信号进行频率搬移和/或移相；所述功率放大器用于对发端混频器输出的信号进行功率放大处理；其中，所述观测信息从发端混频器的信号输出端、所述功率放大器的信号输出端以及所述发端混频器中电压电流转换器的信号输出端中至少一个获取。
本公开实施例还提供了一种信号发射主通路的信号校准链路，其中所述信号发射主通路用于根据补偿系数对产生的信号进行补偿操作后生成射频发射信号，以用于实现目标探测和/或通信，其中：所述信号校准链路，被配置为用于确定在数值不同的初始补偿系数h(0)、第一补偿系数h(1)以及第二补偿系数h(2)的条件下信号发射主通路对应的初始观测信息O(0)、第一观测信息O(1)以及第二观测信息O(2)；利用初始观测信息O(0)、第一观测信息O(1)和第二观测信息O(2)，确定第三补偿系数h(3)，作为所述信号发射链路补偿操作使用的补偿系数。
示例性的，所述补偿系数包括谐波失真补偿参数、本振泄漏补偿参数和正交失衡补偿参数中的至少一个。
示例性的，所述信号发射主通路和所述信号校准链路集成于同一集成电路中。
示例性的，所述集成电路为毫米波雷达芯片，和/或，所述射频发射信号为FMCW信号。
示例性的，所述利用初始观测信息O(0)、第一观测信息O(1)和第二观测信息O(2)，确定第三补偿系数h(3)，包括：以初始观测信息O(0)分别对第一观测信息O(1)和第二观测信息O(2)进行归一化处理；以第三观测信息O(3)为0为条件，利用归一化处理的结果，确定第三补偿系数h(3)。
示例性的，所述以第三观测信息O(3)为0为条件，利用归一化处理的结果，确定第三补偿系数h(3)，包括：以第三观测信息O(3)为0为条件，根据第一比值d1和第二比值d2确定第一补偿系数h(1)对应的第一系数x1和第二补偿系数h(2)对应的第二系数x2；其中第一比值d1为第一差值与初始观测信息O(0)之间的比值，所述第一差值为初始观测信息O(0)与第一观测信息O(1)之间的差值；第二比值d2为第二差值与初始观测信息O(0)之间的比值，所述第二差值为初始观测信息O(0)与第二观测信息O(2)之间的差值；计算第一系数x1与第一补偿系数h(1)之间的乘积，得到第一乘法结果；以及，计算第二系数x2与第二补偿系数h(2)之间的乘积，得到第二乘法结果；根据所述第一乘法结果和所述第二乘法结果确定第三补偿系数h(3)。
示例性的，所述根据第一比值d1和第二比值d2确定第一补偿系数h(1)对应的第一系数x1和第二补偿系数h(2)对应的第二系数x2，包括：构建观测方程，确定所述观测方程中的第一系数x1和第二系数x2；其中所述观测方程的计算表达式为2*2矩阵的逆矩阵与1行2列的第一矩阵之间的乘积，得到1行2列的第二矩阵；其中，2*2矩阵中第一行记录有第一比值d1的实数部分的数值和虚数部分的数值，第二行记录有第二比值d2的实数部分的数值和虚数部分的数值；所述第一矩阵中第一行的取值为1，第二行的取值为0；所述第二矩阵的第一行为第一系数x1的取值，第二行为第二系数x2的取值。
示例性的，所述根据所述第一乘法结果和所述第二乘法结果确定第三补偿系数h(3)，包括：计算所述第一乘法结果和所述第二乘法结果的总和，得到所述第三补偿系数h(3)。
示例性的，所述第一补偿系数h(1)和所述第二补偿系数h(2)是根据初始观测信息O(0)确定的。示例性的，所述初始补偿系数数h(0)的数值为0。
示例性的，所述信号发射主通路包括基带模块和射频模块，其中所述基带模块用于产生基带信号；所述射频模块用于对利用发端本振信号对所述基带信号进行频率搬移和/或移相，以形成射频发射信号；其中，所述观测信息是从所述射频模块获取的。
示例性的，所述射频模块包括发端混频器和功率放大器；其中所述发端混频器利用发端本振信号对所述基带信号进行频率搬移和/或移相；所述功率放大器用于对发端混频器输出的信号进行功率放大处理；其中，所述观测信息从发端混频器的信号输出端、所述功率放大器的信号输出端以及所述发端混频器中电压电流转换器的信号输出端中至少一个获取。
本公开实施例还提供了一种信号补偿链路，包括如本公开任一实施例所述的信号发射链路，以及补偿单元，所述补偿单元可被配置为用于对所述信号发射链路的IQ失配、IQ失衡、信号泄漏、谐波失真缺陷中的至少一个进行补偿。
示例性的，所述补偿单元可被配置为用于基于如本公开任一实施例所述的信号校准链路所得到校准数据对所述信号发射链路进行补偿。
示例性的，所述信号校准链路中的所述信号接收链路作为辅助校准模块集成于待校准的发射链路附近区域，以在所述发射链路工作的间隙进行实时校准操作。
本公开实施例还提供了一种馈线不等长补偿方法，应用于具有至少两个信号链路的电磁波传感器的天线阵列中，所述方法包括：将所述至少两个信号链路中馈线最短的作为基准链路，获取剩余各发射链路相对于该基准链路的时延差；以及基于所述时延差在数字域对所述天线阵列进行馈线不等长的补偿。
示例性的，所述馈线不等长为不同的发射链路之间或者不同的接收链路之间，本振源LO到各个发射通道中的混频器之间存在的馈线长度不等长，和/或各个发射通道的PA到其发射天线之间的馈线长度不等长。
本公开实施例还提供了一种信号校准系统，应用于电磁波传感器中，所述信号校准系统包括信号传输链路和辅助链路，所述辅助链路临近所述信号传输链路集成于所述电磁波传感器中；其中，所述辅助链路可被配置为用于对所述信号传输链路进行实时校准。
示例性的，所述信号传输链路包括信号发射链路和/或信号接收链路；所述辅助链路对应所述信号传输链路包括辅助接收链路和/或辅助信号发射链路；其中，所述辅助接收链路可被配置为用于对所述发射链路进行校准，所述辅助发射链路可被配置为用于对所述接收链路进行校准。
示例性的，所述信号接收链路包括射频部分和基带部分，所述辅助发射链路对应所述信号接收链路包括射频辅助发射单元和基带辅助发射单元；以及所述射频辅助发射单元可被配置为用于对所述射频部分进行校准，所述基带辅助发射单元可被配置为用于对所述基带部分进行校准。
示例性的，所述射频辅助发射单元包括IQ器件；以及其中，在利用所述辅助发射链路对信号接收链路进行校准时，先利用所述基带辅助发射单元对所述基带部分进行校准后，再利用校准后的所述基带部分对所述射频辅助发射单元进行校准，然后再利用校准后的所述射频辅助发射单元对所述射频部分进行校准。
示例性的，所述辅助接收链路包括辅助接收单元和校准接收单元，所述辅助接收单元包括IQ器件；其中，所述校准接收单元可被配置为用于对所述辅助接收单元进行校准， 校准后的所述辅助接收单元可被配置为用于对所述发射链路进行校准。
本公开实施例还提供了一种IQ混频器，所述IQ混频器包括I支路混频单元、Q支路混频单元和变压器单元，其中I支路混频单元可被配置用于输出I路信号，Q支路混频单元可被配置用于输出Q路信号，变压器单元则可被配置用于对I路信号和Q路信号进行磁耦合以合成IQ混频输出信号。
示例性的，所述变压器单元为三圈变压器结构，所述三圈变压器结构包括两个支路电感和一个磁耦合电感；其中，所述I支路混频单元的输出端之间、Q支路混频单元的输出端之间分别串联一个支路电感，所述磁耦合电感设置在两个支路电感之间。
示例性的，所述IQ混频器应用于本公开任一实施例所述的信号发射链路中的数字移相器中，和/或，应用于本公开任一实施例所述的信号校准链路中，和/或，应用于本公开任一实施例所述的信号校准系统中。
本公开实施例还提供了一种集成电路，包括依次连接的射频模块、模拟信号处理模块和数字信号处理模块；所述射频模块用于产生射频发射信号和接收射频接收信号；所述模拟信号处理模块用于对所述射频接收信号进行降频处理以得到中频信号；所述数字信号处理模块用于对所述中频信号进行模数转换以得到数字信号；以及所述射频模块包括本公开任一实施例所述的信号发射链路、本公开任一实施例所述的信号收发链路、本公开任一实施例所述的信号校准链路、本公开任一实施例所述的信号补偿链路、本公开任一实施例所述的信号校准系统，和/或本公开任一实施例所述的的IQ混频器；和/或，所述数字信号处理模块基于本公开任一实施例所述的馈线不等长补偿方法在数字域进行补偿。
示例性的，所述集成电路还包括数据处理模块，所述数据处理模块用于对所述数字信号进行处理以实现目标探测和/或无线通信。示例性的，所述集成电路为毫米波或UWB等芯片。
示例性的，所述射频接收信号为所述射频发射信号被目标发射和/或散射所形成的回波信号，所述集成电路为传感器芯片。
本公开实施例还提供了一种电磁波传感器，包括：承载体；本公开任一实施例所述的所述的集成电路，设置在所处承载体上；天线，设置在所述承载体上，或者所述天线与所述集成电路集成为一体器件设置在所述承载体上；其中，所述集成电路与所述天线连接，用于发射所述射频发射信号和/或接收所述射频接收信号。
本公开实施例还提供了一种设备，包括：设备本体；以及设置于所述设备本体上的如本公开任一实施例所述的电磁波传感器；其中，所述电磁波传感器用于目标检测和/或通信，以向所述设备本体的运行提供参考信息。
本公开实施例还提供了一种非瞬时性计算机可读存储介质，其上存储有计算机可读指令，当所述指令被处理器执行时，使得所述处理器执行本公开任一实施例所述的馈线不等长补偿方法。
附图概述
通过以下参照附图对本公开实施例的描述，本公开的上述以及其他目的、特征和优点将更为清楚。
图1A为模拟移相器架构的信号发射链路简易示意图；
图1B为图1A所示信号发射链路中模拟移相器的简易示意图；
图2A为本公开实施例中数字移相器架构的信号发射链路简易示意图；
图2B为本公开实施例中数字移相器架构的另一种信号发射链路简易示意图；
图2C为一种采用锯齿波调制的FMCW发射信号和回波信号的波形示意图；
图3A为正交不平衡时，发射机正交调制的信号模型示意图；
图3B为真实混频器产生的信号示意图；
图3C为三阶谐波失真的示意图；
图4A为本公开实施例提供一种信号发射链路的结构示意图；
图4B为本公开实施例提供的正交失衡补偿的示意图；
图4C为本公开实施例提供的谐波失真补偿的第一示意图；
图4D为本公开实施例提供的谐波失真补偿的第二示意图；
图5A为本公开实施例中一种信号发射链路中数字移相器架构的示意图；
图5B为本公开实施例中一种包含有具体补偿单元的发射链路的示意图；
图6A为本公开实施例提供的一种信号收发链路的结构示意图；
图6B为本公开示例性实施例提供的另一种信号收发链路的结构示意图；
图7A为本公开实施例中一种包含TX IQ Mod、BIST IQ Mod和RX Real Mixer的收发链路的示意图；
图7B为本公开实施例中一种包含TX IQ Mod、RX IQ De-Mod和LO Freq Diff的收发链路的示意图；
图8为本公开实施例中一种基于图7B所示结构结合BIST的收发链路的示意图；
图9为本公开实施例中一种包含TX IQ Mod、BIST IQ Mod和RX IQ De-Mod的收发链路的示意图；
图10为本公开实施例中一种包含辅助电路和BIST IQ Mod的收发链路的示意图；
图11为本公开实施例中另一种包含辅助电路和BIST IQ Mod的收发链路的示意图；
图12A为本公开实施例中一种混频器的结构示意图；
图12B为本公开实施例中一种发射机中补偿单元的结构示意图；
图13A为本公开实施例中一种基于三次方模块的数字预补偿HD3架构的示意图；
图13B为本公开实施例中一种基于倍频波形发生器模块的数字预补偿HD3架构的示意图；
图13C为本公开实施例中一种基于数字移相器架构发射链路的校准补偿示意图；
图14A为本公开实施例提供的一种信号发射方法的流程示意图；
图14B为本公开实施例提供的确定补偿系数方式的原理说明示意图；
图15为本公开实施例中一种具有至少两个发射通道的发射链路的示意图；
图16为本公开实施例中数字LO信号生成器的结构示意图；
图17为本公开实施例中包含馈线不等长补偿模块的发射链路的结构示意图；
图18为本公开实施例中利用辅助电路对收发链路进行校准及补偿的示意图；
图19为本公开实施例中利用辅助发射电路对接收链路进行校准及补偿的示意图；
图20为本公开实施例中利用辅助接收电路对发射链路进行校准及补偿的示意图；
图21为本公开实施例中一种辅助电路的结构示意图；
图22为本公开实施例中另一种辅助电路的结构示意图；
图23为本公开实施例中IQ Mixer的电路模块示意图；
图24为本公开实施例中一种IQ Mixer的结构示意图；
图25为对应图24中所示结构的示意图；
图26为本公开实施例中另一种IQ Mixer的物理结构示意图；
图27为本公开实施例提供的另一种信号发射方法的流程示意图；
图28为本公开实施例提供的又一种信号发射方法的流程示意图；
图29为本公开实施例提供的又一种信号发射方法的流程示意图。
详述
为了便于理解本公开，下面将参照相关附图对本公开进行更全面的描述。附图中给出了本公开的较佳实施例。但是，本公开可以通过不同的形式来实现，并不限于本文所描述的实施例。相反的，提供这些实施例的目的是使对本公开内容的理解更加透彻全面。
除非另有定义，本文所使用的所有的技术和科学术语与属于本公开的技术领域的技术人员通常理解的含义相同。在本公开的说明书中所使用的术语只是为了描述具体的实施例的目的，不是旨在于限制本公开。
下面，参照附图对本公开技术方案进行详细说明。
图1A为模拟移相器架构的信号发射链路简易示意图，图1B为图1A所示信号发射链路中模拟移相器的简易示意图。
如图1A所示，传感器在发射信号时，针对一个发射链路而言，通过诸如由锁相环(PLL)所构成的信号产生器11来生成LO信号(如77GHz频段的扫频信号)，例如可以是FMCW信号；模拟移相器(Analog PS)12针对所接收的LO信号进行移相操作后，经发射天线13辐射至预定的空间区域内，以进行目标检测及测量等操作。
可选的，图1A所示的发射链路结构中，对应的模拟移相器架构可如图1B所示，其具体的移相原理可如下式所示：
其中
即通过对所接收的LO信号(即LOIN)分别进行0°和90°移相后得到两路信号(即LOI 和LOQ)，并经功率放大器(PA)放大后拟合为一路输出信号LOOUT，即该输出信号LOOUT相较于LOIN输入信号移相θ，其中该移相θ的值是由两路信号LOI的幅度A和LOQ的幅度B所决定。
另外，上述的模拟移相器架构还可通过延迟线(Delay Line Unit)的方式实现，即利用信号的窄带假设(Narrow-band assumption)，通过时延的方式进行移相，其原理如下式所示：
其中，τ为延迟线所延迟时间。
由于上述的模拟移相器的调相精度(Resolution)、调相准确度(Accuracy)均较低，从而无法满足当前传感器的需求。同时，虽然可通过校准来提升其调相精度和调相准确度，但需要对模拟架构的移相器进行离线(off-line)的校准，进而会使得工程实现及产品量产的难度及复杂度大幅提升，当工作处于不同状态(温度)时，离线校准的结果未必准确。同时，模拟移相器还存在面积大、损耗大、以及稳定性和通道耦合等比较严重的问题，在多天线移相器共同作用时，移相器的性能会互相影响，使得系统的性能进一步恶化。
基于此，本公开发明人创造性的提出了一种数字移相器架构的信号发射链路，以有效的提升调相精度和调相准确度，同时还能避免对发射链路中的移相器等链路及器件进行非线上校准的操作，进而降低了工程实现的复杂度及难度。另外，还能有效减小移相架构的发射链路面积和损耗，提升系统的稳定性，降低通道耦合度。
图2A为本公开实施例中数字移相器架构的信号发射链路简易示意图，图2B为本公开实施例中数字移相器架构的另一种信号发射链路简易示意图，图2C为一种采用锯齿波调制的FMCW发射信号和回波信号的波形示意图。
本公开实施例提供了一种信号发射链路，可应用于电磁波传感器中，发射链路可包括模拟信号源和数字移相器，该模拟信号源可被配置为用于提供初始模拟信号(如LO信号)，数字移相器则可被配置为用于在数字域产生移相信号，数字移相器还可基于所产生的移相信号对初始模拟信号进行移相，以将初始模拟信号进行预设的移相操作。
如图2A所示，在一些可选的实施例中，一种数字移相器架构的信号发射链路，可包括模拟信号源21、数字移相器(Digital PS)22和发射天线23等，即模拟信号源21可被配置为用于提供LO信号，数字移相器22则可被配置为用于对所接收的LO信号进行预设移相操作，以使得移相后的LO信号经发射天线23辐射到预设空间区域中。其中，模拟信号源21也可为包含有锁相环PLL的架构，可提供电磁波(例如激光、微波等)信号。其中，模拟信号源21、数字移相器22和发射天线23可集成为一体的器件，也可为相互分立的元器件；例如，模拟信号源21和数字移相器22可集成于一封装体中形成诸如SoC芯片(chip)等，而发射天线23则可通过芯片的外设端口连接，形成在诸如PCB板等载体上。同时，一些可选的实施例中，发射天线23还可集成在芯片的封装上形成AiP或者AoP，具有封装天线的芯片结构。
如图2A所示，本公开实施例中的数字移相器22可包括混频器(Mixer)221、数模转换器(即DAC)222和移相信号源(如数字基带信号源Baseband)223等，即移相信号源223可被配置为用于提供数字移相信号；数模转换器222则可被配置为用于对所接收的数字移相信号进行模数转换，以将数字移相信号转换为模拟的移相信号；混频器221则可被配置为用于将所接收的模拟的移相信号与所接收的来自模拟信号源21的电磁波信号进行混频操作，以达到利用数字移相信号对上述的电磁波信号进行设定的移相操作。可选的，当上述的信号发射链路提供扫频信号时，例如提供FMCW激光信号或FMCW微波信号时，可基于模拟信号源21提供扫频电磁波信号，和/或基于移相信号源223提供扫频数字移相信号，以使得经过混频器221混频后，输出扫频连续波信号。
在一些可选的实施例中，基于图2A所示的结构，模拟信号源21可被配置为用于提供微波中的厘米波频段或毫米波频段(如3.1GHz、24GHz、60GHz、77GHz等频段)FMCW信号，移相信号源223可被配置为用于提供MHz(例如3MHz～5MHz，如3MHz、4MHz、5MHz等)级别的数字移相信号，即数模转换器222对MHz级别的数字移相信号进行数模转换后得到对应频率范围的模拟移相信号，混频器221可被配置为基于所接收的固定频段的模拟移相信号对所接收的毫米波频段的FMCW信号进行上混频或者下混频操作，以实现对FMCW信号进行预设移相的操作。
在一些可选的实施例中，3.1GHz频段的厘米波信号可包括3.1GHz～10.6GHz，例如3.1GHz、、5GHz、5GHz、6GHz、8GHz、10.6GHz等；77GHz频段的毫米波信号可包括76GHz～81GHz的信号，例如76GHz～77GHz、77GHz～79GHz、79GHz～81GHz等扫频信号，或者76GHz、77GHz、78GHz、79GHz、80GHz、81GHz等固定频段信号。
基于图2A所示的结构，由于移相信号源223提供的是数字信号，为了进一步适配信号特性，混频器221可设置为IQ Mixer，数模转换器222为IQ DAC。同时，移相信号源223可被配置为用于提供用于移相的数字基带信号源(DDFS)和/或作为波形控制器(Waveform Control)提供对应的源信号。
如图2B所示，本公开实施例提供了一种信号发射链路，该信号发射链路应用于雷达系统中，该信号发射链路包括：发端基带(Baseband)数字模块201(包括前述的移相信号源223)、数模转换(Ditial-to-Analog Converter，DAC)模块202、发端本振器203和发端正交调制器204，数模转换模块202包括两个相同的数模转换器，其中：发端基带数字模块201，被配置为生成两路正交的发端数字基带信号(即数字移相信号)，并将两路正交的发端数字基带信号分别送入一个数模转换模块202中；数模转换模块202，被配置为将两路正交的发端数字基带信号转换为两路发端模拟基带信号(即模拟移相信号)；发端本振器203，被配置为提供发端本振信号TX_LO(即初始模拟信号)；发端正交调制器204，被配置为基于两路发端模拟基带信号，对发端本振信号TX_LO进行频率搬移的同时进行移相操作，以形成预定移相后的FMCW射频发射信号。
本公开实施例中，发端基带数字模块201提供的发端数字基带信号中包括预设的相位信息；数模转换模块202对所接收的发端数字基带信号进行数模转换，以将发端数字基带信号转换为发端模拟基带信号(如不改变相位信息)；发端正交调制器204将所接收的发端模拟基带信号与发端本振器203生成的发端本振信号TX_LO进行混频操作，以达到基于发端模拟基带信号对发端本振信号进行频率搬移的同时进行预设的移相操作，以形成预定移相后的FMCW射频发射信号。
本公开实施例的信号发射链路，通过使用发端基带数字模块201、数模转换模块202和发端正交调制器204组成了数字移相器架构，由于该架构的基带信号是在数字域产生的，有更好的正交性和更低的旁瓣，因此其移相相位可以非常准确产生，使得调相精度更高，从而实现了带有高精度的数字移相功能的车载雷达系统，降低了天线之间隔离度的需求，同时具有链路损耗小、成本低、无需离线校准的优点，并且可以支持更为灵活的发波方案，比如高性能的多普勒分复用和频分复用等，且能够支持在数字域进行频率响应补偿。
本公开实施例中，由于发端基带数字模块201提供的是数字信号，为了进一步适配信号特性，发端调制器设置为正交调制器(IQ Modulator)，数模转换模块202设置为正交数模转换器(IQ DAC)。
本公开实施例中，发端本振器203可以为包含有锁相环(Phase-Locked Loop，PLL)的架构，可提供电磁波(例如激光、微波等)信号。
在一些示例性实施方式中，该信号发射链路还包括：功率放大器(Power Amplifier，PA)205，其中：功率放大器205，被配置为对移相后的射频信号进行功率放大并将放大后的信号输出至发射天线。
在一些示例性实施方式中，该信号发射链路还包括：发射天线206，其中：发射天线206，被配置为将放大后的信号向预设空间区域辐射。
本公开实施例中，经功率放大器205放大后的信号可通过封装为一体或外置的发射天线206辐射至预设的空间区域中。即，发端本振器203、数字移相器和发射天线206等可集成为一体的器件，也可为相互分立的元器件；例如，发端本振器203和数字移相器可集成于一封装体中形成诸如SoC芯片(chip)等，而发射天线206则可通过芯片的外设端口连接，形成在诸如PCB板等载体上。同时，一些可选的实施例中，发射天线206还可集成在芯片的封装上形成AiP或者AoP，具有封装天线的芯片结构。
在一些示例性实施方式中，扫频信号的频带宽度在2GHz以上。例如，调频连续波雷达系统的发射天线发出的发射信号的电磁波为高频调频连续波，调频连续波雷达系统的接收天线接收的回波信号是目标物反射/散射回来的电磁波。图2C示出了一种示例性的FMCW发射信号和回波信号的波形示意图。如图2C所示，发射信号和回波信号的频率随时间规律变化。调频连续波一般为锯齿形、三角形等，本公开以锯齿形为例进行说明，每个调频周期T内的电磁波称为一个啁啾(Chirp)，每个Chirp的信号的频率随时间线性增加。本公开实施例中，一个啁啾的带宽范围B大于或等于2GHz。
在一些示例性实施方式中，发端数字基带信号为单音信号，发端本振信号为扫频信号。本公开实施例中，发端本振器203可被配置为用于提供微波中的厘米波频段或毫米波频段(如3.1GHz、24GHz、60GHz、77GHz等频段)的FMCW信号，发端基带数字模块201可被配置为用于提供MHz(例如3MHz～5MHz，如3MHz、4MHz、5MHz等)级别的单音发端数字基带信号，即数模转换模块202对MHz级别的单音发端数字基带信号进行数模转换后得到对应频率范围的单音发端模拟基带信号，发端正交调制器204可被配置为基于所接收的单音发端模拟基带信号对所接收的毫米波频段的FMCW信号进行上混频或者下混频操作，以实现对FMCW信号进行预设移相的操作。
示例性的，3.1GHz频段的FMCW信号可包括3.1GHz～10.6GHz之间的扫频信号，如7.163-8.812GHz；77GHz频段的FMCW信号可包括76GHz～81GHz的扫频信号，或者，76GHz～77GHz、77GHz～79GHz、79GHz～81GHz等扫频信号。
在另一些示例性实施方式中，发端数字基带信号可为扫频信号，发端本振信号为单音信号。
本公开实施例中，发端本振器203可被配置为用于提供微波中的厘米波频段或毫米波频段(如3.1GHz、24GHz、60GHz、77GHz等频段)的单音发端本振信号，发端基带数字模块201可被配置为用于提供MHz(例如3MHz～5MHz，如3MHz、4MHz、5MHz等)级别的发端数字基带FMCW信号，即数模转换模块202对MHz级别的发端数字基带FMCW信号进行数模转换后得到对应频率范围的发端模拟基带FMCW信号，发端正交调制器204可被配置为基于所接收的发端模拟基带FMCW信号对所接收的厘米波频段或毫米波频段的单音发端本振信号进行上混频或者下混频操作，以实现对单音发端本振信号进行预设移相和扫频的操作。
示例性的，3.1GHz频段的单音发端本振信号可以为3.1GHz、5GHz、6GHz、8GHz、10.6GHz等固定频段的单音模拟信号；77GHz频段的单音发端本振信号可以为76GHz、77GHz、78GHz、79GHz、80GHz、81GHz等固定频段的单音模拟信号。
在一些示例性实施方式中，该信号发射链路还包括：低通滤波器(Low Pass Filter，LPF)207，设置在数模转换模块202和发端正交调制器204之间，被配置为对数模转换模块202输出的发端模拟基带信号进行低通滤波并输出至发端正交调制器204。
如图2B所示，发端基带数字模块201产生两路正交的数字基带(Baseband)信号，即I路数字基带信号和Q路数字基带信号，并将产生的数字基带信号送入数模转换模块202(包含两个完全相同的DAC，即IQ DAC)中，得到两路模拟基带信号。然后将两路模拟基带信号输入低通滤波器207，滤除带外噪声信号，并通过发端正交调制器204进行正交调制，得到调制后的射频信号，再通过功率放大器205以及发射天线206，将调制后的射频信号辐射出去。
在一些示例性实施方式中，信号发射链路还可以包括直接数字频率合成器(Direct Digital Frequency Synthesizer，DDFS)(图2B中未示出)，设置在发端基带数字模块201和数模转换模块202之间，直接数字频率合成器可被配置为基于所接收的源信号实现CDM(Code-Division Multiplexing，码分复用)、DDM(DopplerDivision Multiplexing，多普勒分割复用)、TDM(Time-Division Multiplexing，时分复用)、SDM(Space Division Multiplexing，空分复用)、CSD(Circuit Switch Data，电路交换数据)、Digital IF(Digital Intermediate Frequency，数字中频)等多种信号波形及发波方式中的至少一种，以实现信号发射形式及发射波形的灵活配置。
图3A为正交不平衡时，发射机正交调制的信号模型示意图。如图3A所示，DSP为数字信号处理器，DAC为数模转换器，LPF为低通滤波器，LO为本振信号源，其中，AI(f)为I通道LPF输出信号的频域表达式，EI(f)为I通道基带频率响应,A′I(f)为I通道基带输出信号的频域表达式；SI(f)为I通道调制器输出信号的频域表达式，GI(f)为I通道射频频率响应，S′I(f)为I通道射频输出信号的频域表达式；其中，αI为I通道混频器的幅度增益，为I通道混频器的相位响应；AQ(f)为Q通道LPF输出信号的频域表达式，EQ(f)为Q通道基带频率响应,A′Q(f)为Q通道基带输出信号的频域表达式；SQ(f)为Q通道调制器输出信号的频域表达式，GQ(f)为Q通道射频频率响应，S′Q(f)为Q通道射频输出信号的频域表达式；其中，αQ为Q通道混频器的幅度增益，为Q通道混频器的相位响应；.URF(f)为发射的调制信号的频域表达式。本公开实施例中，射频包括从混频器到射频信号所经过的所有电路，基带包括从DAC到混频器之前所经过的所有电路。
根据图3A所示的信号模型，正交解调器的输出URF(f)可以表示为：
2URF(f)＝A(f-fm)V1(f-fm)+A*(-f-fm)V1 *(-f-fm)+A*(-f+fm)V2(f-fm)+A(f+fm)V2 *(-f-fm)
其中，A(f-fm)V1(f-fm)为在频率fm附近的期望信号；
A*(-f-fm)V1 *(-f-fm)为在频率-fm附近的期望信号；
A*(-f+fm)V2(f-fm)为在频率fm附近的镜像信号；
A(f+fm)V2 *(-f-fm)为在频率-fm附近的镜像信号；

用表示URF(f)信号的复基带形式，即其中，在f<0时，u(f)＝0；在f≥0时，u(f)＝1。
因此，正交调整器的基带等效模型可为：其中，A(f)V1(f)为在频率fm附近的期望信号；A*(-f)V2(f)为在频率-fm附近的镜像信号。用相对镜像比(relative image ratio，RIR)衡量正交混频器抑制不需要的镜像频率分量的能力，因此RIR的定义是期望信号和镜像信号分量的功率之比，因此可以写成上式说明，I和Q通道的级联增益和相位响应需要相等(每个通道的级联增益和相位响应都应该被考虑)，否则通道是不平衡的，并且镜像分量不会被完全抑制。
图3B为真实混频器产生的信号示意图。如图3B所示，FIN表示混频器的输入信号，FOUT表示混频器的输出信号，FLO表示混频器使用的本振信号，其中，混频器的输入信号为：频率为FBB的基带信号，混频器使用的本振信号为：频率为FLO的高频信号，混频器的输出信号为：频率为FLO±FBB的信号。
理想的混频器将产生两个输入的乘积的输出。在频率方面，输出应该是FLO+FBB和FLO-FBB，不含其他项。如果任一输入不在驱动状态下，则不会有输出。然而，在FBB和FLO，真实混频器还将产生一些能量。在FBB处产生的能量可以忽略不计，因为它远离所需的输出，并且将被混频器输出之后的RF器件滤除。无论FBB处产生的能量如何，FLO处产生的能量都可能是一个问题。它非常接近或在所需的输出信号内，并且很难或无法通过滤波去除，因为滤波也会滤除所需的信号。LO还有其他途径可以泄漏到系统输出端，例如通过电源或跨越硅本身。无论本振如何泄漏，其泄漏都可被称为LO泄漏(LO Leakage，LOL)。
虽然模拟(射频)电路可以用小信号表示的线性模型来近似，但器件的非线性往往会产生一些无法预测的有趣且重要的现象。无记忆系统中非线性的输入和输出特性可以近似为：
y(t)＝α1x(t)+α2x2(t)+α3x3(t)+...≈α1x(t)+α2x2(t)+α3x3(t)
其中许多器件的非线性效应是由三阶项引起，如果且 那么

其中：
如果系统具有奇对称性，也就是说，系统是完全微分的，那么偶次谐波项可以抵消掉，剩下的则是奇次谐波项，这也被称之为谐波失真(Harmonic distortion)。
图3C为三阶谐波失真的示意图。如图3C所示，基带处理器产生的基带信号为频率为BB的基带信号，经IQ调制器处理后可得到期望信号Wanted、镜像信号IMG以及三阶谐波HD3，经功率放大器PA输出的信号包括期望信号Wanted、镜像信号IMG以及三阶谐波HD3。
HD3的主要来源是基带模拟器件的三阶非线性。如上述推导可知，基带BB信号的3阶谐波分量HD3位于-3BB处。经过混频器后，基带的HD3上变频为LO-3BB，即C-IM3。经过PA放大后，调制产生的信号很难在高频进行区分。
基于上述说明可知，对于数字正交调制的信号发射链路而言，需要面对IQ Imbalance、harmonic distortion、以及LO Leakage等问题。理想的单音基带信号经过正交上混频器后，会在高频产生多个非理想不同频的单音信号。因此，本公开从系统架构上，利用在发射端增加数字预补偿模块，去解决数字正交调制雷达TX中由于模拟器件非理想特性引入的问题。本公开在设计系统架构时，不仅包含预补偿模块，还包含校准环路，实现实时地校准以及补偿。
图4A为本公开实施例提供一种信号发射链路的结构示意图。如图4A所示，信号发射链路包括集成于同一集成电路中的信号发射主通路和信号校准链路；其中：所述信号校准链路被配置为用于对所述信号发射主通路进行校准以获取补偿信息；以及所述信号发射主通路被配置为用于根据所述补偿信息进行补偿操作后生成射频发射信号，以用于实现目标探测和/或通信。
由于信号校准链路集成于包括有所述发射主通路的集成电路中，使得信号校准链路能实时对发射主通路进行校准操作，且校准链路的校准操作可以不受发射主通路的运行环境的变化而变化，使得发射主通路能够得到更加精确的校准信息，从而提高发射主通路的信号处理性能。
进一步的，所述补偿信息包括谐波失真补偿参数、本振泄漏补偿参数和正交失衡补偿参数中的至少一个。由于该信号校准链路产生的补偿信息能够解决正交失衡、LO泄漏以及谐波失真问题，从而有效提升数字移相器架构的发射主通路中信号质量。
在一种实现方式中，所述集成电路为毫米波雷达芯片或超宽带(Ultra Wide Band，UWB)芯片等，和/或，所述射频发射信号为FMCW信号。
所述信号发射主通路可包括第一信号源和移相器；其中，所述第一信号源被配置为生成第一模拟信号；以及所述移相器被配置为对所述第一模拟信号进行频率搬移和/或移相，以形成射频发射信号。
在所述移相器为非正交架构时，所述移相器包括第二信号源和发端混频器，其中所述第二信号源被配置为用于生成第二模拟信号，所述发端混频器被配置为对所述第一模拟信号和所述第二模拟信号进行混频处理，以形成所述射频发射信号；在所述移相器为正交架构时，其中所述移相器包括：第二信号源、数模转换模块和发端混频器；其中所述第二信号源被配置为生成第一数字信号；所述数模转换模块被配置为将所述第一数字信号转换为第二模拟信号；以及所述发端混频器被配置为基于所述第二模拟信号对所述第一模拟信号进行频率搬移和/或移相，以形成所述射频发射信号。
其中，所述发射主通路还包括补偿电路，其中所述补偿电路的信号输入端与所述第二信号源相连，信号输入端与所述移相器相连，所述补偿电路用于将补偿信号和第二信号源输出的信号进行合并处理后再输出。
在所述信号发射主通路输出的信号为两路正交信号时，所述补偿电路所使用的补偿信号为两路正交信号；在所述信号发射主通路输出的信号不是两路正交信号时，所述补偿电路所使用的补偿信号为一路与第二信号源输出的信号类型相同的信号，其中信号类型为数字信号或模拟信号。
其中，所述补偿电路包括补偿信号产生器和加法器，所述补偿信号产生器，可被配置为生成所述补偿信号；所述加法器，与所述补偿信号产生器和所述第二信号源相连，用于将所述第二信号源输出的信号与所述补偿信号产生器输出的补偿信号进行信号叠加操作。
具体的，在所述信号发射主通路输出的信号为两路正交信号时，所述补偿信号产生器包括谐波补偿信号单元、正交失衡补偿信号单元和本振泄露补偿信号单元中的至少一个；在所述信号发射主通路输出的信号不是两路正交信号时，所述补偿电路包括谐波补偿信号单元和本振泄露补偿信号单元中的至少一个；其中，所述谐波补偿信号单元产生的补偿信号用于消除所述信号发射主通路中主频信号的谐波信号；所述本振泄露补偿信号单元的补偿信号用于补偿所述信号发射主通路中发端本振信号产生的泄露信号；所述正交失衡补偿信号单元产生的补偿信号用于补偿所述信号发射主通路中主频信号的镜像信号。
所述正交失衡补偿电路所使用的补偿信号在所述信号发射主通路对应的镜像信号，与所述信号发射主通路产生的期望信号对应的镜像信号之间频率相同、振幅相同、相位相反。
在一个示例性实施例中，所述正交失衡补偿电路所使用的补偿信号是根据第二信号源输出的信号以及所述第二信号源输出的信号的频率反转的复共轭信号确定的。其中，所述正交失衡补偿电路产生的补偿信号的获取方式包括：获取预设的预补偿系数与所述复共轭信号的乘积，得到所述复共轭信号对应的调整信号；计算所述第二信号源输出的信号与所述调整信号之间的差值，得到所述正交失衡补偿电路产生的补偿信号。所述预补偿系数是基于期望信号的振幅与所述期望信号对应的镜像信号的振幅的比值确定的。
图4B为本公开实施例提供的正交失衡补偿的示意图。如图4B所示，为了实现TX正交调制器的补偿，即抑制不希望的镜像信号的存在，可以采用预补偿方法。令hIQ为预补偿系数，作用于待传输的基带信号的复共轭信号A*(-f)。预补偿基带信号的频域表达式可以写成
Apc(f)＝A(f)-hIQ·A*(-f)→apc(n)＝α(n)-hIQ·a*(n)
当这个预补偿的信号现在用作输入正交调制器的信号时，调制器输出的频域形式可以表示为：
从上式可看出，只需要保证V2(-ft)-hIQV1(ft)＝0，就能够通过预补偿的方式，压制镜像分量。因此，预补偿系数可以表示为：
其中，预补偿系数表示最优预补偿滤波器的频率响应，该滤波器将在调制器的数字域中用于在传输前对I和Q信号进行预失真(预补偿)。可见，只需要估计V2(-ft)与V1(ft)之间的比值即可。
在一个示例性实施例中，所述本振泄露补偿信号单元产生的补偿信号是基于移相器所使用的第一模拟信号对应的泄漏信号生成的。其中，所述本振泄露补偿信号单元产生的补偿信号与所述泄露信号的频率相同、振幅相同但相位相反。
具体的，消除LOL(LO Leakage Correction,LOC)是通过产生一个与LOL振幅相等但相位相反的信号来实现的，从而抵消它。在得到LOL的确切幅度和相位后，可以通过对发射机的输入施加直流偏移来产生抵消信号。正交混频器结构使其能够很好地产生抵消信号。由于混频器中存在LO的正交信号，因此可以在LO的频率产生任何相位和幅度的信号。可以在复平面进行几何解释。第一行为时域IQ两路合成的结果，第二行为从IQ两路信号频谱角度进行几何解释。对发射信号施加直流偏置后，混频器的输出端将包含所需的发射信号以及所需的LOL抵消信号。特意生成的抵消信号LOC将与无用的LOL抵消，仅留下所需的发射信号。
理想情况下TxLOC＝hDC，I×LOI-hDC，Q×LOQ＝-TxLOL因此LO泄露的补偿方式可以表示为：apc(n)＝a(n)-hDC
在实际中，可以将LOL按照下面的方式进行建模：
理想的IQ分别为αIcos(wmt)和αQsin(wmt)；实际的IQ分别为αIcos(wmt+φI)和αQsin(wmt+φQ)；LO泄漏为因此，需要通过如下方式去对消LO Leakage：
因此，可以通过给I和Q路信号加不同的直流偏置，来对校LOL。
在一个示例性实施例中，所述谐波补偿信号单元产生的补偿信号与所述信号发射主通路中谐波信号的频率相同、振幅相同、相位相反。
具体的，所述谐波补偿信号单元包括n次方模块或n倍频信号发生器；其中：n次方模块，其中所述n次方模块的信号输入端与所述第二信号源的信号输出端相连，用于利用第二信号源输出的信号，生成频率为所述第二信号源输出的信号的频率的n倍的信号，得到所述补偿信号；n倍频信号发生器，用于产生频率为所述第二信号源输出的信号的频率的n倍的信号，得到所述补偿信号，其中n的取值为正整数。其中，n的取值为奇数。进一步的，n的取值为3。
图4C和图4D为本公开实施例提供的谐波失真补偿的示意图。如图4C和图4D所示，与消除LOL的原理相似，对于三阶谐波失真存在两种补偿方式，分别是基于n次方模块的补偿架构和基于n倍频波形发生器的补偿架构。
apc(n)＝a(n)-hHD3·(α*(n))3
apc(n)＝a(n)-hHD3·a3(n)
图5A为基于图2A所示信号发射链路中数字移相器架构的示意图；图5B为本公开实施例中一种包含有具体补偿单元的发射链路的示意图。如图5A所示，数字移相器架构的信号发射链路(TX digital phase shifter architecture)可包括数字基带信号源(Baseband)、直接数字频率合成器(Direct Digital Frequency Synthesizer，简称DDFS)、IQ数模转换器(Digital to Analog Convertor，简称DAC)、低通滤波器(Low-Pass Filter，简称LPF)、IQ调制器(IQ modulator/IQ Mixer)、功率放大器(Power Amplifier，PA)等，即基带信号源被配置为用于提供数字移相源信号，直接数字频率合成器可被配置为基于所接收的源信号实现CDM(Code-Division Multiplexing，码分复用)、DDM(Doppler Division Multiplexing，多普勒分割复用)、TDM(Time-Division Multiplexing，时分复用)、SDM(Space Division Multiplexing，空分复用)、CSD(Circuit Switch Data，电路交换数据)、Digital IF(Digital Intermediate Frequency，数字中频)等多种信号波形及发波方式中的至少一种，以实现信号发射形式及发射波形的灵活配置。其中，经功率放大器放大后的信号可通过封装为一体或外置的发射天线辐射至预设的空间区域中。
针对本公开实施例中的数字移相器架构的信号发射链路，由于数字移相器架构被配置为可在数字域产生基带信号序列，并可通过DAC产生模拟基带信号，然后经过正交混频器将发射信号调制到高频，即因为该架构的基带信号是在数字域产生的，有更好的正交性和更低的旁瓣，因此其移相相位可以非常准确产生，使得调相精度更高。
在一些可选的实施例中，针对数字移相器架构的信号发射链路，当采用RF LO扫频实现所发射信号为FMCW信号时，针对所可能出现的因IQ失配(Mismatch)而引起的TX IQ失衡(Imbalance)、信号泄漏(如TX LO Leakage)谐波失真(harmonic distortion,简称HD)等问题，还可在该信号发射链路增设补偿单元。如图5A所示，可通过在TX DDFS与IQ DAC之间设置补偿单元(TX compensation)，以对数字移相器架构的信号发射链路进行校准(calibration)及补偿(compensation)等操作，以到达解决上述问题中的至少之一的操作。其中，由基带的三阶非线性所引起的HD可简称为HD3。
在一些可选的实施例中，如图5A至图5B所示，TX补偿单元(TX compensation)可包括LO泄露补偿单元(TX LO leakage compensation)、IQ失衡补偿单元(TX IQ Imbalance compensation)和HD3补偿单元(TX HD3 compensation)中的至少一个，其中，LO泄露补偿单元可被配置为用于针对信号泄漏(leakage)而进行的补偿操作，IQ失衡补偿单元可被配置为用于针对IQ失衡而进行的补偿操作，HD3补偿单元可被配置为用于针对上述的HD3而进行的补偿操作。其中，IQ失衡补偿单元被配置为可用于对IQ调制器失衡(TX IQ Modulator imbalance)、IQ通道失衡(IQ channel imbalance)中的至少一个进行补偿操作。另外，当TX补偿单元包括LO泄露补偿单元、IQ失衡补偿单元和HD3补偿单元中的至少两个时，可以依据实际需求及信号特性等同步进行补偿(如并联方式)，也可依次进行补偿(如串联方式)，如图5B所示可以先进行IQ失衡补偿、然后再进行LO泄露补偿，最后进行HD3补偿等操作。
在一些可选的实施例中，针对数字移相器架构的信号发射链路，还可包括有针对DAC的误差校正模块(TX DAC Board Error Correction)和针对高斯白噪声的AWGN(additive white gaussian noise)模块等，图中未示出，具体的可根据实际需求而增设或删减。其中，本公开实施例中提及的IQ中I可表示为In-Phase(即同相)的简称，Q表示为Quadrature(即正交)的简称，RF可表示为Radio Frequency(即射频)的简称。
本公开实施例还提供了一种信号收发链路，包括信号发射链路和信号接收链路，如图6A或图6B所示，信号发射链路可以包括：发端基带数字模块201、数模转换模块202、发端本振器203和发端正交调制器204，其中：发端基带数字模块201，被配置为生成两路正交的发端数字基带信号，并将生成的发端数字基带信号送入数模转换模块202中；数模转换模块202，被配置为将发端数字基带信号转换为发端模拟基带信号；发端本振器203，被配置为提供发端本振信号TX_LO；发端正交调制器204，被配置为基于发端模拟基带信号对发端本振信号TX_LO进行移相操作，得到移相后的射频信号。
信号接收链路可以包括收端本振器302、收端混频器303、模数转换器(Analog-to-Digital Converter，ADC)304和收端基带数字模块305；其中，收端本振器302，被配置为提供收端本振信号；收端混频器303，被配置为基于收端本振信号对所接收的回波信号进行混频操作，得到收端模拟基带信号；模数转换器304，被配置为将收端模拟基带信号转换为收端数字基带信号；收端基带数字模块305，被配置为对收端数字基带信号进行处理，以实现目标检测和/或无线通信，例如，得到目标的诸如距离、速度、角度、高度及微运动特性等参数信息。
本公开实施例中，通过发端基带数字模块201产生的两路理想的I路数字基带信号和Q路数字基带信号，经由数模转换模块202后，可以得到非常理想的复信号，且该复信号的相位可以通过发端基带数字模块201精确控制。通过如图6A或如图6B所示的信号收发链路中的接收机结构，能够有效获取信号发射链路的射频信号的相位信息，从而可以实现多天线的相位调制。
在一些示例性实施方式中，信号发射链路还可以包括：功率放大器205，其中，功率放大器205，被配置为对移相后的射频信号进行功率放大并将放大后的信号输出至发射天线。
在一些示例性实施方式中，信号发射链路还可以包括：发射天线206，其中，发射天线206，被配置为将放大后的信号向预设空间区域辐射。
在一些示例性实施方式中，信号接收链路还可以包括接收天线301，其中，接收天线301，被配置为接收回波信号，回波信号为信号发射链路所发射的信号被目标物反射和/或散射而形成的信号。
在一些示例性实施方式中，收端本振信号可以为扫频信号，或者，收端本振信号可以为单音信号。
本公开实施例中，在上述信号收发链路中，针对信号发射链路中的发端正交调制器204所接收的TX-LO信号与信号接收链路中的收端混频器303所接收的RX-LO信号的频率可相同。例如，假设发端基带数字模块201输出的信号为x MHz的正弦波(sine wave)，则TX-LO信号和RX-LO信号则可同为z GHz的正弦波(sine wave)，其中，x、z均为正数，一般可在0～1000之间。
下面通过图6A所示的信号收发链路对本公开的原理进行介绍。如前所述，对于FMCW雷达系统而言，信号发射链路可以有两种发波方案：1)发端本振信号扫频，发端数字基带信号单音；2)发端本振信号单音，发端数字基带信号扫频。假设发端本振信号TX_LO、发端数字基带信号、调制后的发射信号分别用TLO(t)、BB(t)和TX(t)来表示，并用下标I和q表示I路信号和Q路信号，用上标a表示其复信号形式，则在两种发波方案下，信号发射链路各阶段的信号可以表示如下：
1)发端本振信号扫频，发端数字基带信号单音：



2)发端本振信号单音，发端数字基带信号扫频：



其中，为FMCW信号的扫频斜率，fbb为发端数字基带信号的起始频率，ftlo为发端本振信号的起始频率，为发端数字基带信号的初始相位，为发端本振信号的初始相位。
假设图6A所示的信号接收链路中的收端本振信号RX_LO用RLO(t)来表示，本实施例中，假设收端本振信号RX_LO为扫频信号，则RLO(t)可以表示为：
其中，frlo为收端本振信号的起始频率，Φ0为收端本振信号的初始相位，那么，信号发射链路发射的射频信号，经过目标反射/散射后产生回波信号，经过接收天线301、收端混频器303和收端低通滤波器307后可得：
其中，τ代表信号发射链路发射的射频信号经过目标反射/散射后返回信号接收链路的时延。
上式说明：上述两种扫频方式，均可以通过控制发端数字基带信号的初始相位来进行精确的相位调制，从而实现高精度高准确度的移相，避免了在高频进行直接移相。
在一些可选的实施例中，接收天线301可通过芯片的外设端口连接，形成在诸如PCB板等载体上。同时，在另一些可选的实施例中，接收天线也可集成在芯片的封装上形成AiP或者AoP，即具有封装天线的芯片结构。
在一些示例性实施方式中，信号接收链路还可以包括低噪声放大器(Low Noise Amplifier，LNA)306，设置在接收天线301和收端混频器303之间，对接收天线301接收的回波信号进行低噪声放大后，再送入收端混频器303。
在一些示例性实施方式中，信号接收链路还可以包括串联连接的低通滤波器(Low Pass Filter，LPF)307和高通滤波器(High Pass Filter，HPF)308，设置在收端混频器303和模数转换器304之间，低通滤波器307和高通滤波器308组成带通滤波器，用于滤除带外噪声。
在一些示例性实施方式中，如图6A所示，在信号接收链路中，收端混频器303可以为实数混频器，模数转换器304可以为实数模数转换器。
本公开实施例中，虽然信号发射链路采用数字移相架构，但是，信号接收链路可包括正交接收架构或非正交接收架构的接收机，因此，能够有效的兼容各种架构的接收链路的传感器，有效降低整个收发链路系统的开发成本。
在另一些示例性实施方式中，如图6B所示，在信号接收链路中，收端混频器303可以为正交混频器，模数转换器304可以为正交模数转换器。
为了匹配数字移相器架构的信号发射链路，本公开实施例将信号接收链路中的收端混频器303调整为IQ解调器(IQ Demodulator)，同时将模数转换器304调整为IQ ADC，经接收天线接收的回波信号依次经上述的低噪声放大器306、收端混频器303、低通滤波器307、高通滤波器308及模数转换器304处理后，转换为IQ数字基带信号，后续的收端基带数字模块305对该IQ数字基带信号处理可得到目标的诸如距离、速度、角度、高度及微运动特性(即微多普勒)等参数信息。
当收端混频器303为正交混频器且收端本振信号为扫频信号时，收端本振信号RX_LO可以表示为：
在另一些示例性实施方式中，信号接收链路的收端本振信号除了为式(9)或式(11)所示的扫频信号外，还可以为式(12)或式(13)所示的单音信号。
收端混频器303为实数混频器：RLO(t)＝cos(2πfrlot+Φ0)(12)。
收端混频器303为正交混频器：
当收端本振信号为单音信号时，只需要在收端基带数字模块305中增加数字域的信号处理流程(包括数字域混频操作)，同样可以得到需要的数字基带信号。
综上所述，本公开实施例根据不同的发射方案和接收方案的搭配组合(例如，发端采用数字基带信号单音、本振信号扫频还是采用数字基带信号扫频、本振信号单音；收端采用实数混频器、实数模数转换器还是采用正交混频器、正交模数转换器；收端采用单音本振信号还是采用扫频本振信号)，可以拓展出多种系统级别的技术方案。
图7A为本公开实施例中一种包含TX IQ调制(Mod)、RX实数解调(Real De-Mod)的收发链路的示意图，图7B为本公开实施例中一种包含TX IQ Mod、RX IQ De-Mod的收发链路的示意图，图8为本公开实施例中一种基于图7B所示结构结合BIST的收发链路的示意图，图9为本公开实施例中一种包含TX IQ Mod、BIST IQ Mod和RX IQ De-Mod的收发链路的示意图。
下面基于本公开实施例中所记载的发射链路结构，针对其所构成的收发链路进行说明：
如图7A所示，一种收发链路，可包括发射链路和接收链路等。发射链路可包括依次连接的数字基带信号源(Baseband)、直接数字频率合成器(TX DDFS)、IQ数模转换器(IQ DAC)、低通滤波器(LPF)、IQ调制器(IQ Modulator)、功率放大器(power amplifier，PA)等，同时经功率放大器放大后的信号经发射天线向预设空间区域进行辐射。接收链路则可包括依次连接的低噪声放大器(low noise amplifier，简称LNA)、实数混频器(Real Mixer)、跨阻放大器(Trans-Impedance Amplifier，简称TIA)、低通滤波器(LPF)、高通滤波器(High-Pass Filter，简称HPF)、实数数模转换器(Real ADC)等，即经接收天线接收的回波信号依次经上述的LNA、Real Mixer、TIA、LPF、HPF及Real ADC处理后，转换为实数数字基带信号，后续的数字信号处理模块对该实数数字基带信号处理可得到目标的诸如距离、速度、角度、高度及微运动特性等参数信息。其中，上述收发链路中，针对发射链路中的IQ调制器所接收的TX-LO信号与接收链路中的Real Mixer所接收的RX-LO信号的频率可相同。例如，如图7A所示，若是Baseband输出的信号为x MHz的正弦波(sine wave)，则TX-LO信号和RX-LO信号则可同为z GHz的正弦波(sine wave)。
在图7A所示的实施例中，其发射链路采用的数字移相架构，而接收链路则可采用模拟架构的元器件，即不用采用IQ的元器件的，所以能够有效的兼容模拟架构的接收链路的传感器，有效降低整个收发链路系统的开发成本。
可选的，在本公开的实施例中，接收链路可包含有接收天线，即该接收天线可通过芯片的外设端口连接，形成在诸如PCB板等载体上。同时，一些可选的实施例中，接收天线也可集成在芯片的封装上形成AiP或者AoP，即具有封装天线的芯片结构。
在一些可选的实施例中，为了匹配数字移相器架构的发射链路，可针对接收链路对应进行相关的调整，如图7B所示的收发链路中，可包含基于图7A中类似的发射链路架构和接收链路(为了避免赘述，相同的部分具体在此便不予阐述)，即将图7A中接收链路中的Real Mixer调整为IQ解调器(IQ Demodulator)，同时将Real ADC调整为IQ ADC，即此时接收链路则可包括依次连接的低噪声放大器LNA)、IQ解调器(IQ Demodulator)、跨阻放大器(TIA)、低通滤波器(LPF)、高通滤波器(HPF)、IQ数模转换器(IQ ADC)等，即经接收天线接收的回波信号依次经上述的LNA、IQ Demodulator、TIA、LPF、HPF及IQ ADC处理后，转换为IQ数字基带信号，后续的数字信号处理模块对该IQ数字基带信号处理可得到目标的诸如距离、速度、角度、高度及微运动特性(即微多普勒)等参数信息。
同时，基于图7B所示的收发链路进行自校准时，只要将发射链路的信号输出端口与接收链路的信号输入端口直接通过传输线连接，即发射链路通过该传输线直接将发射信号发送至接收链路中，以在不经过发射天线和接收天线情况下，实现收和/或发链路的自校准操作。其中，此时发射链路中的IQ调制器所接收的TX-LO信号与接收链路中的IQ Demodulator所接收的RX-LO信号之间具有一定的频偏。例如，如图7B所示，若是Baseband输出的信号为x MHz的正弦波(sine wave)，则TX-LO信号可为z GHz的正弦波(sine wave)，此时RX-LO信号则可为(z GHz-yMHz)的正弦波(sine wave)，即此时的频偏为y MHz。
在一些可选的实施例中，如图7B所示，可通过增加接收链路(如图中所示的接收机(Receiver，RX))对发射链路(如图中所示的发射机(Transmitter，TX))进行校准，并通过发射链路中的TX IQ失衡补偿单元基于校准的数据进行补偿操作。同时，也可通过复用实际用于信号发收的接收链路(如图中所示的接收机(Receiver，RX))对发射链路(如图中所示的发射机(Transmitter，TX))进行校准，通过发射链路和/或接收链路中的TX IQ失衡补偿单元基于校准的数据进行补偿操作。在其他的实施例中也可进行类似的实施，为了阐述简便，后续便不予赘述。
在一些可选的实施例中，基于图7B所示结构的基础上，为了实现收发链路精准校准，可在图7B所示的接收链路的IQ Demodulator的RX-LO端口设置内部自测试模块(Built-in Self-Test，简称BIST)模块，即如图8所示的，基于图7B所示收发链路结构的基础上，在接收链路的IQ Demodulator的RX-LO端口设置IQ BIST架构，以实现在接收链路的IQ  Demodulator的RX-LO端口输入具有预设频偏的LO信号。例如，通过相位角度转换器和IQ调制器(IQ Modulator)等所构成IQ BIST，利用所接收的诸如TX-LO信号经相位角度转换器通过IQ调制器，以基于该IQ调制器的另一路输入信号BIST-LO的频偏信号形成频偏后的信号输入至IQ Demodulator的RX-LO端口。例如，若TX-LO的信号为z GHz的正弦波，而BIST-LO的信号为y MHz的正弦波，则得到输入至IQ Demodulator的RX-LO端口的频偏后的信号为(z GHz-yMHz)。需要说明的是，不同的实施例之间，x、y、z均为示意值，具体的数值可相同或不相同。
在一些可选的实施例中，基于图8中所示收发链路结构的IQ BIST架构，还可通过复用收发链路中的接收链路针对数字移相器架构的发射链路进行校准；其中，在其他实施例中，涉及到利用接收链路对发射链路的校准操作，以及利用发射链路对接收链路的校准操作，均可通过复用实际进行信号收发的链路中对应的接收链路或者发射链路实现上述的校准操作，也可通过增设对应的校准接收链路或者校准发射链路来实现对实际进行信号收发的链路中对应的发射链路或者接收链路的校准操作。
可选的，该IQ BIST可包括相位角度转换器和IQ调制器(IQ Modulator)，相位角度转换器用于实现对数字架构的发射链路中I路和Q路的分别校准，而IQ调制器的另一路输入信号BIST-LO则可为y MHz的正弦波(sine wave)，用于模拟发射信号被目标反射所形成回波信号相关的特性，其中图8中的x、y、z均为正数，且x≠y≠z，一般可在0～1000之间。
可选的，在图8所示的收发链路中，还可在发射链路中(例如在的TX DDFS与IQ DAC之间)设置TX IQ失衡补偿单元(TX IQ Imbalance Compensation)，和/或在接收链路中(例如在IQ ADC之后)设置TX IQ失衡补偿单元(TX IQ Imbalance Compensation)，即可基于上述自校准操作所得到的校准参数(或系数)对所发射和/或接收的信号进行补充，以解决诸如IQ不平衡等问题。
在一些可选的实施例中，基于图8所示的结构，如图9所示，可将上述的IQ BIST模块设置在发射链路的信号输出端口与接收链路的信号输入端口之间，即发射链路通过该IQ BIST模块直接将发射信号发送至接收链路中，以在不经过发射天线和接收天线情况下，实现收链路和/或发链路的自校准操作。
需要说明的是，在图7A、图8至图9所示的发射链路结构中，其仅示例出IQ失衡补偿单元(TX IQ Imbalance compensation)，在实际的应用中，也可基于实际需求，于发射链路中增加LO泄露补偿单元(TX LO leakage compensation)和HD3补偿单元(TX HD3compensation)等，以构成包含有LO泄露补偿单元(TX LO leakage compensation)、IQ失衡补偿单元(TX IQ Imbalance compensation)和/或HD3补偿单元(TX HD3 compensation)等单元的补偿单元(TX compensation)。
图10为本公开实施例中一种包含辅助电路和BIST IQ Mod的收发链路的示意图，图11为本公开实施例中另一种包含辅助电路和BIST IQ Mod的收发链路的示意图。
如图10所示，一种收发链路，结合图5B和图9所示的结构及相关描述，该收发链路可包括发射链路、接收链路和校准链路。其中发射链路可包括依次连接的TX数字基带信号源(TX Baseband)、直接数字频率合成器(TX DDFS)、补偿单元(Compensation)、IQ数模转换器(IQ DAC)、低通滤波器(LPF)、IQ调制器(IQ Modulator)和功率放大器(power amplifier，PA)等，同时经功率放大器放大后的信号经发射天线向预设空间区域进行辐射。接收链路则可包括依次连接的低噪声放大器(low noise amplifier，简称LNA)、实数混频器(Real Mixer)、跨阻放大器(Trans-Impedance Amplifier，简称TIA)、高通滤波器(High-Pass Filter，简称HPF)、可变自动增益放大器(Variable Gain Amplifier，简称VGA)、实数数模转换器(Real ADC)和用于TX RF校准(Calibration)的RX Baseband等，即经接收天线接收的回波信号依次经上述的LNA、Real Mixer、TIA、HPF、VGA及Real ADC等处理后，转换为实数数字基带信号，后续的数字信号处理模块对该实数数字基带信号处理可得到目标的诸如距离、速度、角度、高度及微运动特性等参数信息。其中，针对发射链路，通过设置在TX DDFS与IQ DAC之间的补偿单元(TX compensation)可包括LO泄露补偿单元(TX LO leakage compensation)、IQ失衡补偿单元(TX IQ Imbalance compensation)和/或HD3补偿单元(TX HD3 compensation)等单元，用以实现针对数字移相器架构的发射链路中的LO leakage、IQ Imbalance及HD3等对应的补偿操作。
在一些可选的实施例中，在发射链路与接收链路之间还可设置有校准模块，该校准模块可被配置用于复用接收链路对上述的数字移相器架构的发射链路进行校准等操作。同时，补偿模块(即前述实施例中的各种补偿单元)则可基于该校准模块的校准操作所获取的参数或系数，以在发射链路端就可以实现对发射信号的补偿操作。而在其他的实施例中，还可同时或单独的在接收链路中设置对应的接收补偿模块，即此时接收补偿模块可基于上述校准操作所获取的参数或系数等，以在接收链路端实现对回波信号的补偿。
如图10所示，上述的校准模块可包括BIST单元和辅助电路单元(auxiliary circuit)等， 即发射链路的输出端口通过BIST单元和辅助电路单元连接至接收链路中Real Mixer与Real ADC之间的任一节点上，如发射链路中的IQ Modulator基于x MHz的数字移相基带信号和z GHz的LO信号，生成(z GHz±x MHz)的射频信号后，经输出端口输出至BIST单元，该BIST单元对于接收的射频信号进行y MHz的频偏操作后得到(z GHz±x MHz±y MHz)的模拟回波信号，再利用辅助单元中的IQ De-Modulator进行降频后，以得到预设的中频信号(z GHz±x MHz±y MHz-z GHz＝±x MHz±y MHz)后，该中频信号输入至接收链路中预设节点，以实现对发射链路中的校准操作。
可选的，辅助电路单元可为正交解调器电路，该辅助电路单元的输出端可连接至接收链路中的TIA与HPF之间的节点、HPF与VGA之间的节点、VGA与Real ADC之间的节点中的任一节点上。另外，为了最大程度的复用接收链路的结构，一路发射链路的输出端口通过BIST单元和辅助电路单元后，I、Q两条支路还可分别连接至不同的发射链路上，即如图10所示，即通过复用两路接收链路来对一路发射链路进行校准。其中，在完成针对发射链路的校准后，可利用上述的补偿单元(TX compensation)中的LO泄露补偿单元(TX LO leakage compensation)、IQ失衡补偿单元(TX IQ Imbalance compensation)和/或HD3补偿单元(TX HD3 compensation)等补偿单元，用以基于校准所得到的参数实现针对数字移相器架构的发射链路中的LO leakage、IQ Imbalance及HD3等问题对应的补偿操作。
在一些的可选的实施例中，上述的BIST单元可包括依次连接的相位角度转换器和IQ调制器(IQ Modulator)，辅助电路单元可包括依次连接的LNA、IQ De-Modulator和TIA，即相位角度转换器接收从发射链路输出的射频信号，而IQ Modulator的一个输入端连接相位角度转换器的输出端，另一个输入端则接收y MHz的BIST-LO信号，用以生成预设的回波信号。LNA则将接收的回波信号经放大后发送至IQ De-Modulator一个输入端，IQ De-Modulator另一个输入端则用于接收z GHz的RX-LO信号后，该IQ De-Modulator的两个输出支路(即I支路和Q支路)分别经TIA后连接至各自对应的接收链路中相应的节点，以将生成的预设的中频信号输出至两个接收链路，进而在实现校准操作的同时，更加高效的复用接收链路设计。
需要说明的是，针对本公开实施例中的校准操作，若是发射链路发射的是扫频信号，在实际的校准操作时，可采用TX LO信号为单tone信号进行逐点校准；同时，也可以采用TX LO信号为扫频信号进行大带宽的校准操作，甚至可采用扫频带宽校准一次实现针对全频段的扫频信号的校准操作。
基于图10所示的结构，为了进一步的将数字移相器架构的发射链路中，诸如HD3、LO Leakage、IQ Imbalance等压制预设的程度，可通过级联至少两个BIST单元的方式来实现；如图11所示，通过两个串联的BIST单元，可将因上述缺陷而导致的噪声压制在-50dB的程度，从而有效的降低相关链路模拟器件的开发设计的难度。
在一些可选的实施例中，基于本公开实施例所记载的数字移相器架构的发射链路的基础上，在针对IQ Imbalance进行校准及补偿操作时，可在时域(Time-Domain)基于频谱分析得到IQ Imbalance的补偿系数，也可在频域(Frequency-Domain)基于频谱谱峰比值得到IQ Imbalance的补偿系数。
在一些可选的实施例中，为了进一步提升IQ Imbalance补偿系数的精准性，还可通过迭代校准及补偿的方式来逼近理想的补偿系数，也可通过多观测校准及补偿的方式来得到理想的补偿系数。
例如，针对迭代校准及补偿的方式，可基于前后两次校准补偿的补偿系数之间的大小关系，或者该两次校准补偿的补偿系数之间的差值是否满足预设迭代条件，来确定是否停止迭代操作，并以停止迭代操作时所得到的补偿系数作为当前场景中，最终的补偿系数进行后续的操作。针对多观测校准及补偿的方式，则可通过多次(例如三次)校准补偿操作后，分别对各次操作得到的测量数据进行FFT(快速傅里叶变换)并获取相应的幅度和相位信息后，可将测量做差并归一化处理得到相关的数据，并构建出观测矩阵；后续基于对该观测矩阵求逆得到的数据反向求解出对应的补偿系数。
在一些可选的实施例中，基于上述得到IQ Imbalance补偿系数类似的思想，也可采用诸如迭代校准及补偿的方式，或者多次观测校准及补偿的方式来得到LO leakage和/或HD3的补偿系数。
在上述带有补偿单元的发射机各示例中提及了关于解决发射机中谐波失真的问题。经研究发现，一些谐波失真的原因可能来自于发射机中的混频器等包含非线性特点的器件。
以图12A为例，图12A为本公开实施例中混频器的结构示意图。如图12A所示，混频器包括电压电流转换器(V/I Converter)、电流开关(Current Switch)和电流电压转换器(I/V Converter)。其中，所述电压电流转换器将接收的电压信号转换为电流信号；所述电流开关与所述电压电流转换器和所述第二信号发生器相连，用于利用本振信号对电压电流转换器输出的电流信号进行处理；所述电流电压转换器与所述电流开关相连，用于对所述电流开关输出的电流信号转换为电压信号。
在上述结构中，由于电压电流转换器设置有晶体管放大器，基于晶体管放大器的非线性特性以及基带信号的频率较低的特性，使得电压电流转换器输出的电流信号存在基带信号对应的谐波信号。例如，由基带的三阶非线性所引起的HD可简称为HD3。类似地，五阶非线性所引起的谐波称为HD5。在电流开关处理电压电流转换器输出的电流信号时，谐波的频率经过上变频处理后变为射频频段。由于抑制射频频段的谐波信号的操作复杂度较高，硬件成本较高。如果对射频频段的谐波信号不进行去除，会影响雷达收发的信号质量，进而影响雷达测量的准确度。
所述补偿单元用于将所生成的抵消信号输入至信号发射链路，以抵消所述射频信号中的谐波信号。其中，所述补偿单元与所述第一信号发生器相互独立。
为此，补偿单元可包括抵消信号发生器。利用补偿单元输出的抵消信号能够抑制射频信号中的谐波信号，减少射频信号中谐波分量，从而提高发射机输出的射频信号的信号质量。
在本公开实施例中，针对信号发射链路中的谐波信号，补偿单元利用反馈或根据发射波的特点，将所生成的抵消信号输入至信号发射链路，以抵消所述信号发射链路所输出的射频信号中的谐波信号。其中该抵消信号具有与射频发射电路中所传输的谐波信号具有相位相反、幅值相近等特点，以达到抑制谐波信号的目的。
在一些示例中，补偿单元根据第一信号发生器所产生的基带信号的相位、频率、或幅值，甚至与LO信号合并的路径长度等参数，产生包含抵消作用的补偿信号。
例如，图5A中示出了一种将补偿单元接入信号发射链路中的发射机示例。在图5A所示结构中，补偿单元为TX补偿单元。TX补偿单元中包含可根据发射波的特点生成抵消信号发生器(未予图示)。所述抵消信号发生器可举例为如图5B中所示的TX HD3补偿单元。
其中，基带处理器(图5A中简称为基带框)控制TX DDFS生成的正交数字的基带信号，TX补偿单元根据所述正交数字信号的参数，生成正交补偿信号，将正交补偿信号和正交数字信号合并并发送给IQ DAC以转换为模拟的基带信号。通过LPF滤波处理后，交由混频器(即，图5A中的IQ调制器)进行混频处理，得到基于TX LO信号和模拟基带信号混频的射频信号，PA对混频信号进行放大处理后通过发射天线输出。其中，该补偿信号抵消了射频发射电路中至少部分的谐波信号，如HD3谐波信号。因此，所发射的射频信号中的杂波将大大减少。射频信号可为FMCW信号。
在另一些示例中，补偿单元根据经射频发射电路反馈而得到的谐波信息，来生成补偿信号。参见图12B，图12B为图5A所示发射机中补偿单元的结构示意图。如图12B所示，所述补偿单元包括采集电路和抵消信号发生器。
其中，所述采集电路耦接所述射频发射电路，用于采集所述射频发射电路中的信号，得到采集信号。所述采集信号(或称采样信号)能反映谐波信号中的波形信息(又称谐波参数)，如主频信号的相位、谐波信号的相位，谐波信号的频率，主频信号的频率，谐波信号的功率，主频信号的功率等。
需要说明的是，所述采集信号所反映的谐波参数与采集电路所能采集的信号所携带的信息相关。例如，采集电路为一种功率采集电路，则相应的采集信号包含主频的功率。又如，采集电路利用接收机的至少部分电路，则采集信号反映主频信号的相位、谐波信号的相位、谐波信号的频率、主频信号的频率、谐波信号的功率、主频信号的功率等。
上述至少一种谐波参数可藉由模拟电路来提取。例如，通过耦合器和功率检测器来输出主频信号的功率。或利用雷达芯片中的数字电路在频域计算的优势，来提取谐波参数。例如，通过耦接射频发射电路，来获取与耦接处传输的信号相同的信号作为采集信号。该采集信号中携带主频信号和谐波信号。采集信号经ADC转为数字信号，并交由数字电路在频域计算以得到更多谐波参数。
在一种实现方式中，所述采集电路的输入端与所述混频器的输出端或信号检测端相连。该种方式可检测到因电压电流转换器所产生的谐波信号，且具有简化的采集电路。
在一个示例性实施例中，所述信号校准链路，所述信号校准链路的第一输入端连接于所述信号发射主通路中发端混频器内电压电流转换器与电流开关之间，第二输入端所述信号发射主通路与发射天线之间，信号输出端与所述信号发射主通路中的补偿电路相连，用于从所述第一输入端和所述第二输入端中的至少一个获取所述信号发射主通路中的信号，并根据获取的信号确定补偿信息。例如图13A、13B所示结构中的一种连接方式，该采集电路的输入端连接在电压电流转换器和电流开关之间的检测端，并与ADC耦接。
参见图13C所示，所述信号校准链路包括校准解调器(IQ Demodulator)、多路选择器(MUX)和校准模块(TX Calbration)；其中：所述校准解调器，被配置为从所述第二输入端获取信号发射主通路中的信号，并进行解调处理；所述多路选择器具有两个信号输入端和一个信号输入端，其中一个信号输入端连接于发端混频器内电压电流转换器与电流开关之间，另一个信号输入端连接于校准解调器的信号输出端，用于输出两个信号输入端中一个对应的信号；校准模块，被配置为根据所述多路选择器输出的信号确定所述补偿信息。
在另一种方式中，所述采集电路的输入端连接所述射频发射电路的射频输出端或射频检测端相连。其中，所述射频输出端举例为射频发射电路的输出端。射频检测端举例为射频发射电路中至少一级PA的输入端或输出端。该种方式可采集到射频发射电路中的更准确的谐波参数，但具有较复杂的电路结构。
在一些包含BIST模块的芯片中，采集电路可藉由BIST模块中的部分或全部电路获取采集信号。例如，参见图13B所示，采集电路的输入端耦接射频输出端，其依次包括下变频器、滤波器等，并连接IQ ADC，以输出数字的采集信号。其中，下变频器、滤波器等可复用BIST模块或接收机。
所采集的采集信号输入至抵消信号发生器。其中，所述抵消信号发生器为补偿单元中的至少一种电路。该抵消信号发生器与所述第一信号发生器相连，使得射频发射电路接收的信号同时包括基带信号和抵消信号。
例如，抵消信号发生器包括前述提及的抵消信号发生器，以及提取谐波信息的数字电路。其中，提取谐波信息的数字电路可独立配置，或至少部分的共用于雷达芯片中数字电路。
其中，提取谐波信息的数字电路举例利用雷达芯片中用于处理差频基带信号的数字电路来提取谐波频率、主频频率、主频功率等谐波信息，并提供给抵消信号发生器。抵消信号发生器根据所接收的参数生成抵消信号。
又如，提取谐波信息的数字电路提取采集信号中的主频幅度，并根据预设的主频幅度和谐波幅度之间的差异，计算出谐波幅度。抵消信号发生器根据所计算出的谐波幅度和预配置的其他谐波参数，生成谐波信号补偿信号。其中，该预配置的各谐波参数可根据雷达芯片待发射的主频信号的扫频范围、相位等计算得到。
所述抵消信号发生器中的抵消信号发生器可与第一信号发生器独立配置，或至少部分的共用。例如，抵消信号发生器所产生的抵消信号输入第一信号发生器，使得第一信号发生器所输出的基带信号中包含有所述抵消信号。其中，抵消信号发生器可包括三次谐波发生器和五次谐波发生器。
又如，所述补偿单元中还包括加法器，耦接抵消信号发生器和第一信号发生器，以合并第一信号发生器所产生的基带信号以及所述抵消信号发生器所产生的抵消信号。本实施例中，抵消信号包括三次谐波发生器所产生的抵消三次谐波的抵消信号Signal_HD3，和五次谐波发生器所产生的抵消五次谐波的抵消信号Signal_HD5。抵消信号Signal_HD3和Signal_HD5以及第一信号发生器所产生的基带信号通过加法器合并，并输出至射频发射电路。
综上可见，本公开提供的采用反馈方式向射频发射电路中预输入抵消信号的发射机各电路示例，可在不同环境下，保证芯片所发射的射频信号中含有足够低的谐波信号。
为了在芯片使用过程中能按照芯片的实际运行环境有效地对谐波信号进行抑制，例如考虑环境温度对半导体器件的影响等，本公开还提供一种利用反馈机制对上述发射机中的谐波信号进行信号抵消方法，包括：步骤10、对信号发射链路中的信号进行采集操作，得到采集信号；其中，所述信号发射链路用于产生雷达探测用的射频信号，其中所述射频信号中包含谐波信号。步骤20、检测所述采集信号，生成用于抵消所述谐波信号的抵消信号，并输出至所述信号发射链路。
本公开实施例提供的方法，对信号发射链路中的信号进行采集操作，得到采集信号，并利用采集信号生成抵消信号，并输出至信号发射链路，从而利用抵消信号抑制射频信号中的谐波信号，减少射频信号中谐波分量，从而提高发射机输出的射频信号的信号质量，进而提高接收机对射频信号的接收性能。
下面结合附图7B至图13C，举例发射机及其工作过程：
例如，图7B中示出了一种利用反馈机制提取发射机中的谐波信息，使得补偿单元产生相应的抵消信号的示例。在图7B所示结构中，以补偿单元包括TX HD3补偿单元为例。TX HD3补偿单元根据接收信号的波形特点生成补偿信号。其中，该反馈机制可在雷达芯片的校准模式下执行，以防止削弱雷达芯片正常探测时的信号发射功率。
发射机中的基带处理器(如图7B中的基带框)控制TX DDFS生成的正交数字的基带信号，经TX补偿单元产生的正交数字的抵消信号合并后发送给IQ DAC，以转换为模拟的基带信号。该模拟的基带信号中混合有用于抵消发射链路中谐波信号的模拟的抵消信号。该模拟的基带信号通过LPF滤波处理后进入第一混频器(即，图7B中的IQ调制器)。第一混频器利用TX LO对接收的滤波信号进行混频处理，得到射频信号。该射频信号经耦合通过接收机输出至补偿单元中的TX HD3校准电路(如图7B中的TX HD3校准框)。其中，所述TX HD3校准电路可视为一种提取谐波信息的数字电路。
在接收机中，LNA对发射机输出的信号进行放大处理后，输出给第二混频器(即，图7B中的IQ解调器)，得到解调信号，将解调信号发送给跨阻放大器进行放大处理，再依次经过LPF和HPF后，再经过IQ ADC进行模数转换操作后发送给TX HD3校准电路。该TX HD3校准电路从反馈的信号中提取发射机中的谐波信息，并通过上层控制器转换为产生抵消信号所需的参数，并提供给TX HD3补偿单元。其中，TX HD3校准电路所获取的谐波信息举例包括以下一种或多种参数：谐波信号(或主频信号)的初始相位、起始频率、截止频率、频率变化时长、中心频率等。TX HD3补偿单元或上层控制器根据谐波信息确定补偿单元中用于生成抵消信号的参数，如初始相位、抵消信号的频率，时延等。
需要说明的是，上述各示例中涉及三次谐波和/或五次谐波的抵消操作，可根据发射机的需求而定。
图13A为本公开实施例中一种基于三次方模块的数字预补偿HD3架构的示意图，图13B为本公开实施例中一种基于倍频波形发生器模块的数字预补偿HD3架构的示意图。
在一些可选的实施例中，基于本公开实施例所记载的数字移相器架构的发射链路的基础上，在针对HD3进行校准及补偿操作时，基于有源混频器中HD3的产生主要源头为V/IConverter的非线性中三次谐波，所以可通过如图13A所示的基于三次方模块的补偿架构，或者如图13B所示的基于三倍频波形发生器的补偿结构来实现。
图13C为本公开实施例中一种基于数字移相器架构发射链路的校准补偿示意图。如图13C所示，基于本公开实施例中针对IQ Imbalance、LO leakage和HD3校准补偿操作的相关技术内容：针对IQ Imbalance的补偿操作，可通过补偿BB(基带)信号的共轭信号来反向抵消镜像分量的方式实现，且该补偿操作方式不受IQ Imbalance的校准方式的影响。针对LO Leakage的补偿则可通过调节IQ两路的DC分量(即直流偏置)来实现，同样该LO Leakage的校准方式对其补偿方案不产生影响。针对HD3，由于正交混频器V/I Converter的3阶谐波失真是产生HD3的主要来源，而谐波失真会受到直流偏置的影响，因此当发射链路的LO Leakage和HD3均需要校准时，则需要在进行LO Leakage的校准后再进行HD3的校准，以确保HD3校准的精准性能。
另外，基于数字三次方模块的数字预补偿架构和基于倍频波形发生器模块的数字预补偿架构的HD3的补偿方式，会直接影响后续的校准方案和后续的补偿流程。具体的：
在一个可选的实施例中，针对基于数字三次方模块的数字预补偿架构，可先校准并补偿LO Leakage后，在稳定的DC偏置下，校准出HD3问题的根本来源，即HD3的补偿系数；随后校准IQ Imbalance，并继续补偿IQ Imbalance后，分别对IQ两路依据IQ Imbalance预补偿的结果，去补偿三次谐波失真。
在一个可选的实施例中，针对基于倍频波形发生器模块的数字预补偿架构，可在校准并补偿LO Leakage后，校准得到HD3的补偿系数，并在稳定的DC偏置下校准IQ Imbalance，以及补偿IQ Imbalance；后续通过补偿后的结果，分别计算出IQ两路信号的实际波形，HD3的补偿系数，反算出需要预补偿的3倍频和5倍频的波形信息。
在另一个可选的实施例中，针对基于倍频波形发生器模块的数字预补偿架构，可先校准并补偿LO Leakage，然后通过多次(如三次)观测以同时校准出HD3和IQ Imbalance的预补偿系数，之后再通过再次观测(如两次)来校准出HD3镜像位置处的预补偿系数；最后通过HD3和HD3镜像位置处的预补偿系数反算出需要预补偿的3倍频和5倍频系数。需要说明的是，本公开实施例中的观测用以表示测试以及不同测试结果比较分析等操作。
参见图13C所示结构可知，该采集电路具有两条采集支路，且两条采集支路能够动态切换；其中，一条采集支路的输入端连接在电压电流转换器和电流开关之间；另一条采集支路的输入端连接在功率放大器的输出端，且该采集支路设置有IQ解调器；此外，图13C所示结构中，该采集电路还设置有多路选择器(Multiplexers)，其中该多路选择器的输入端分别连接两个采集支路的输出端，输出端用于输出所述采集信号。
在图13A至图13C所示结构中，该采集信号可以为模拟信号，即，该采集电路的输出端与IQ ADC相连；或者，该采集信号可以为数字信号，则该采集电路至少包括IQ ADC。
需要说明的是，当信号发射链路发送的射频信号不是正交信号时，则图13A至图13C中采集电路可以利用非正交元器件来完成。例如，可以利用单端的下变频混频器代替IQ解频器，可以使用单端的模数转换器代替IQ ADC。
如图14A所示，本公开实施例还提供了一种信号发射方法，应用于具有至少一个信号发射链路的电磁波器件中，该信号发射方法包括：步骤1401、确定每个信号发射链路的射频发射信号的相位；步骤1402、根据射频发射信号的相位确定每个信号发射链路中的发端数字基带信号的初始相位；步骤1403、根据确定的初始相位，生成发端数字基带信号；步骤1404、将发端数字基带信号转换为发端模拟基带信号，基于发端模拟基带信号对发端本振信号进行移相操作。
本公开实施例的信号发射方法，通过使用数字移相器架构在数字域产生数字基带信号，有更好的正交性和更低的旁瓣，其移相相位可以非常准确产生，使得调相精度更高，从而实现了带有高精度的数字移相功能的车载雷达系统，降低了天线之间隔离度的需求，同时具有链路损耗小、成本低、无需离线校准的优点，并且可以支持更为灵活的发波方案，比如高性能的多普勒分复用和频分复用等，且能够支持在数字域进行频率响应补偿。
在一些示例性实施方式中，发端数字基带信号为单音信号，发端本振信号为扫频信号；或者，发端数字基带信号为扫频信号，发端本振信号为单音信号。
在一些示例性实施方式中，扫频信号的频带宽度在2GHz以上。
在一些可选的实施例中，为了进一步提升IQ Imbalance补偿系数的精准性，还可通过迭代校准及补偿的方式来逼近理想的补偿系数，也可通过多观测校准及补偿的方式来得到实际工作状态下使用的补偿系数。其中，补偿系数的确定需要额外考虑链路特性对补偿系数的计算准确度的影响。
以正交失衡为例进行说明，实际工作状态下的补偿系数h为理想状态下的补偿系数H为由于二者的表达式存在明显差别，使得实际工作状态下的补偿系数与理想状态下的补偿系数存在差异。其中：
由于ft和载频相比，属于窄带信号，因此可以近似认为
需要保证W1(Δf-ft)≈W1(Δf+ft)，或者能够通过校准的方式较好地估计出这两个系数的比值；
V1(ft)和V1(-ft)的差异
*EI(ft)(EQ(ft))与EI(-ft)(EQ(-ft))幅度相同，但相位相反；
*与由于ft＝5MHz和载频76-81GHz相比，属于窄带信号，因此可以近似认为相等；
*和不受影响。
如果未考虑了链路的频响特性，而直接利用观测的进行补偿，是不能对消镜像分量，因此在实际的系统中，必须考虑到链路的频率响应对校准操作的影响。
同理可知，谐波失真和LO泄露，在信号发射链路实际工作状态下和理想状态下的补偿系数也存在差异。
图14B为本公开实施例提供的确定补偿系数方式的原理说明示意图。下面对图14B中每个坐标系中的内容进行逐个说明：图14B中(a)表示在信号发射链路在理想状态下补偿系数H的相位和幅值；图14B中(b)表示在信号发射链路在理想状态下补偿系数H的相位和幅值，在实际工作状态下补偿系数h的相位和幅值，以及，实际工作状态下补偿系数h与理想状态下补偿系数H的差值D；图14B中(c)表示图14B中(b)中三个向量不受幅值的变化而变化；因此，只需在相位上控制补偿系数h无限接近补偿系数H即可，具体参见图14B(d)。图14B中(e)表示区域1表示补偿系数h已接近补偿系数H，区域2表示补偿系数h未接近补偿系数H；图14B中(f)对图14B中(e)的区域进行优化，包括缩小区域1的大小，从而增大区域2，并对增大后的区域2进行进一步划分，得到的区域3和新的区域2，其中区域3补偿系数h未接近补偿系数H，新的区域2表示与补偿系数h的相位的相反值的补偿系数已接近补偿系数H；例如，图14B(g)和图14B(h)中的补偿系数h均不在区域1，但是图14B(h)中的补偿系数h较图14B(g)中的补偿系数h更接近补偿系数H，可通过图14B(g)和图14B(h)中差值D的长度明确获知。
基于上述分析，本公开实施例提出迭代校准及补偿的方式。
具体的，可基于前后两次校准补偿的补偿系数之间的大小关系，或者该两次校准补偿的补偿系数之间的差值是否满足预设迭代条件，来确定是否停止迭代操作，并以停止迭代操作时所得到的补偿系数作为当前场景中，最终的补偿系数进行后续的操作。
在一些可选的实施例中，基于上述得到IQ Imbalance补偿系数类似的思想，也可采用诸如迭代校准及补偿的方式，或者多次观测校准及补偿的方式来得到LO leakage和/或HD3的补偿系数。
具体的，本公开实施例提供一种信号发射主通路的信号校准链路，其中所述信号发射主通路用于根据补偿系数对产生的信号进行补偿操作后生成射频发射信号，以用于实现目标探测和/或通信，其中：所述信号校准链路，被配置为用于获取当前补偿系数下信号发射主通路当前的观测信息；以及，在当前的观测信息满足迭代条件时，将当前的补偿系数作为所述信号发射链路补偿操作使用的补偿系数；否则，对当前的补偿系数进行迭代直到所得到的观测信息满足所述迭代条件为止。
下面以实施例对上述信号校准链路进行说明：
在一些可选的实施例中，一种信号校准的方法可包括以下步骤：步骤A1：以初始补偿系数h(0)为0，获取信号发射链路的信号状态，得到初始观测信息O(0)。步骤A2：利用初始观测信息O(0)确定的第一补偿系数h(1)对该信号发射链路进行第一次补偿，获取信号发射链路的信号状态，得到第一观测信息O(1)。步骤A3：如果初始观测信息O(0)和第一观测信息O(1)之间的差异值大于预设的差异门限，则根据第k-1补偿系数h(k-1)和第k-1观测信息O(k-1)确定第k补偿系数h(k)，k为大于或等于2的整数。如果初始观测信息O(0)的绝对值与第一观测信息O(1)的绝对值之间的差值大于预设的差值门限，则以第一补偿系数h(1)更新为初始观测量O(0)为基础，迭代计算第k-1补偿系数h(k-1)和k-1观测信息O(k-1)之间的总和，确定第k补偿系数h(k)。如果第一观测信息O(1)的绝对值与初始观测信息O(0)的绝对值之间的差值大于预设的差值门限，以第一补偿系数h(1)更新为初始观测量O(0)的相反数为基础，迭代计算第k-1补偿系数h(k-1)和k-1观测信息O(k-1)之间的差值，确定第k补偿系数h(k)。步骤A4：如果初始观测信息O(0)和第一观测信息O(1)之间的差异值小于预设的差异门限，则根据预设的相位调整量对初始观测信息O(0)进行调整，得到调整后的第一补偿系数h(1)，再根据调整后的第一补偿系数h(1)重新得到新的第一观测信息O(1)，以新的第一观测信息O(1)为基础，确定第k补偿系数h(k)；其中，初始观测信息O(0)与新的第一观测信息O(1)之间的差异值大于所述差异门限。
可选的，如果初始观测信息O(0)的绝对值与新的第一观测信息O(1)的绝对值之间的差值大于所述差值门限，则迭代计算第k-1观测信息O(k-1)与经所述相位调整量处理第k-1观测信息O(k-1)后得到的调整后的第k-1观测信息O(k-1)之间的总和，确定第k补偿系数h(k)。
可选的，如果新的第一观测信息O(1)的绝对值与初始观测信息O(0)的绝对值之间的差值大于所述差值门限，更新第一补偿系数h(1)为调整后的第一补偿系数h(1)的相反数，迭代计算第k-1补偿系数h(k-1)与经所述相位调整量处理第k-1观测信息O(k-1)后得到的调整后的第k-1观测信息O(k-1)之间的差值，确定第k补偿系数h(k)。其中，在步骤A3和步骤A4中每种迭代操作过程中，判断每个补偿系数对应的观测信息是否小于预设的迭代门限；如果观测信息小于所述迭代门限，停止迭代操作，将小于所述迭代门限的观测信息所使用的补偿系数作为补偿系数。
在另一些可选的实施例中，以正交失衡的补偿操作为例进行说明，其中以IQ的下标对上述实施例提及的符号进行区分，具体的：步骤B1：当hIQ(0)＝0时，估计得到OIQ(0)。步骤B2：当hIQ(1)＝OIQ(0)时，得到OIQ(1)。步骤B3：比较OIQ(0)与OIQ(1)的大小关系，决定后续迭代关系；如果|OIQ(0)|＞|OIQ(1)|+ε，其中ε表示门限，则后续迭代表达式如下：hIQ(1)＝OIQ(0)；hIQ(k)＝hIQ(k-1)+OIQ(k-1)，其中k＝2,3，…….。如果|OIQ(0)|＜|OIQ(1)|-ε，其中ε表示门限，则后续迭代表达式如下：hIQ(1)＝-OIQ(0)；hIQ(k)＝hIQ(k-1)-OIQ(k-1)，其中k＝2,3，…….。步骤B4：如果不满足步骤B3中的关系，则令重新得到OIQ(1)：如果|OIQ(0)|＞|OIQ(1)|+ε，其中ε表示门限，则后续迭代表达式如下： 其中k＝2,3，…….。如果|OIQ(0)|＜|OIQ(1)|-ε，其中ε表示门限，则后续迭代表达式如下： 其中k＝2,3，…….。步骤B5：当OIQ(k)小于某个门限(其中门限的取值小于或等于-50db)，则停止迭代，此时
在另一些可选的实施例中，以LO泄漏的补偿操作为例进行说明,其中以DC的下标对前述实施例中提及的符号进行区分，具体的：步骤C1：当hDC,I(0)＝hDC,Q(0)＝0时，估计得到ODC(0)；其中，hDC,I表示I路信号的初始补偿系数，hDC,Q表示Q路信号的初始补偿系数。步骤C2：当hDC,I(1)＝Re{ODC(0)}、hDC,Q(1)＝Im{ODCQ(0)}时，得到ODC(1)；其中，Re{ODC(0)}表示ODC(0)的实数部分；Im{ODCQ(0)}表示对ODC(0)的虚数部分。步骤C3：比较OIQ(0)与OIQ(1)的大小关系，决定后续迭代关系；如果|OIQ(0)|＞|OIQ(1)|+ε，其中ε表示门限，则后续迭代表达式如下：hDC,I(1)＝Re{ODC(0)}；hDC,Q(1)＝Im{ODCQ(0)}；hDC,I(k)＝hDC,I(k-1)+Re{ODC(k-1)}，其中k＝2,3，…….；hDC,Q(k)＝hDC,Q(k-1)+Im{ODC(k-1)}，其中k＝2,3，…….；如果|OIQ(0)|＜|OIQ(1)|+ε，其中ε表示门限，则后续迭代表达式如下：hDC,I(1)＝-Re{ODC(0)}；hDC,Q(1)＝-Im{ODCQ(0)}；hDC,I(k)＝hDC,I(k-1)-Re{ODC(k-1)}，其中k＝2,3，…….；hDC,Q(k)＝hDC,Q(k-1)-Im{ODC(k-1)}，其中k＝2,3，…….。步骤C4：如果不满足步骤C3中的关系，则令重新得到ODC(1)：如果|OIQ(0)|＞|OIQ(1)|+ε，其中ε表示门限，则后续迭代表达式如下： 其中k＝2,3，…….；其中k＝2,3，…….；如果|OIQ(0)|＜|OIQ(1)|+ε，其中ε表示门限，则后续迭代表达式如下： 其中k＝2,3，…….；其中k＝2,3，…….。步骤C5：当ODC(k)小于某个门限(其中门限的取值小于或等于-50db)，则停止迭代，此时
在另一些可选的实施例中，以谐波失真的补偿操作为例进行说明，其中以HD3的下标对前述实施例中提及的符号进行区分。
基带(BB)信号的初相位是x度，那么后续的3BB信号的初相需要是3x度，且O(n)代表3BB信号所对应频点的幅度相位值÷BB信号所对应频点的幅度相位值。
具体的，步骤D1：当hHD3(0)＝0时，估计得到OHD3(0)。步骤D2：当hHD3(1)＝OHD3(0)时，得到OHD3(1)。步骤D3：比较OHD3(0)与OHD3(1)的大小关系，决定后续迭代关系；如果|OHD3(0)|＞|OHD3(1)|+ε，其中ε表示门限，则后续迭代表达式如下：hHD3(1)＝OHD3(0)；hHD3(k)＝hHD3(k-1)+OHD3(k-1)，其中k＝2,3，…….；如果|OHD3(0)|＜|OHD3(1)|-ε，其中ε表示门限，则后续迭代表达式如下：hHD3(1)＝-OHD3(0)；hHD3(k)＝hHD3(k-1)-OHD3(k-1)，其中k＝2,3，…….；步骤D4：如果不满足步骤D3中的关系，则令重新得到OIQ(1)：如果|OHD3(0)|＞|OHD3(1)|+ε，其中ε表示门限，则后续迭代表达式如下：其中k＝2,3，…….；如果|OHD3(0)|＜|OHD3(1)|-ε，其中ε表示门限，则后续迭代表达式如下： 其中k＝2,3，…….。步骤D5：当OHD3(k)小于某个门限(其中门限的取值小于或等于-50db)，则停止迭代，此时
此外，基于上述分析，本公开实施例还提出多观测校准及补偿的方式。
具体的，可通过多次(例如三次)校准补偿操作后，分别对各次操作得到的测量数据进行FFT(快速傅里叶变换)并获取相应的幅度和相位信息后，可将测量做差并归一化处理得到相关的数据，并构建出观测矩阵；后续基于对该观测矩阵求逆的得到数据反向求解出对应的补偿系数。
在一些可选的实施例中，基于上述得到IQ Imbalance补偿系数类似的思想，也可采用诸如迭代校准及补偿的方式，或者多次观测校准及补偿的方式来得到LO leakage和/或HD3的补偿系数。
具体的，一种信号发射主通路的信号校准链路，其中所述信号发射主通路用于根据补偿系数对产生的信号进行补偿操作后生成射频发射信号，以用于实现目标探测和/或通信，其中：所述信号校准链路，被配置为用于确定在数值不同的初始补偿系数h(0)、第一补偿系数h(1)以及第二补偿系数h(2)的条件下信号发射主通路对应的初始观测信息O(0)、第一观测信息O(1)以及第二观测信息O(2)；利用初始观测信息O(0)、第一观测信息O(1)和第二观测信息O(2)，确定第三补偿系数h(3)，作为所述信号发射链路补偿操作使用的补偿系数。
在另一个可选的实施例中，针对正交失衡补偿的方法可包括：步骤E1：以初始补偿系数h(0)为0，获取信号发射链路的信号状态，得到初始观测信息O(0)。步骤E2：利用初始观测信息O(0)确定的第一补偿系数h(1)对该信号发射链路进行第一次补偿，获取信号发射链路的信号状态，得到第一观测信息O(1)；以及，利用初始观测信息O(0)确定的第二补偿系数h(2)对该信号发射链路进行第二次补偿，获取信号发射链路的信号状态，得到第二观测信息O(2)；其中第二补偿系数h(2)和第一补偿系数h(1)的数值不同，第二观测信息O(2)和第一观测信息O(1)的数值不同。步骤E3：获取第一差值与初始观测信息O(0)之间的比值，得到第一比值d1；以及，获取第二差值与初始观测信息O(0)之间的比值，得到第二比值d2；其中，所述第一差值为初始观测信息O(0)与第一观测信息O(1)之间的差值，所述第二差值为初始观测信息O(0)与第二观测信息O(2)之间的差值。步骤E4：以第三观测信息O(3)为0为条件，根据第一比值d1和第二比值d2确定第一补偿系数h(1)对应的第一系数x1和第二补偿系数h(2)对应的第二系数x2；具体的，构建2*2矩阵，其中2*2矩阵中第一行记录有第一比值的实数部分的数值和虚数部分的数值，第二行记录有第二比值的实数部分的数值和虚数部分的数值；计算2*2矩阵的逆矩阵与1行2列的第一矩阵之间的乘积，得到1行2列的第二矩阵，其中所述第一矩阵中第一行的取值为1，第二行的取值为0，其中第二矩阵的第一行为第一系数x1的取值，第二行为第二系数x2的取值。步骤E5：计算第一系数x1与第一补偿系数h(1)之间的乘积，得到第一乘法结果；以及，计算第二系数x2与第二补偿系数h(2)之间的乘积，得到第二乘法结果；将第一乘法结果和第二乘法结果作为第三补偿系数h(3)。
例如，步骤1：当hIQ(0)＝0时，估计得到步骤2：当hIQ(1)＝b＝|OIQ(0)|时，步骤3：当时，步骤4：将量测作差并归一化处理得到：

步骤5：补偿的本质：
在另一个可选的实施例中，针对LO泄露的方法可包括：步骤F1：以I路信号和Q路信号的初始补偿系数均为0，获取信号发射链路的信号状态，得到初始观测信息O(0)。步骤F2：以I路信号的第一补偿系数为第一数值，Q路信号的第一补偿系数为0，对该信号发射链路进行第一次补偿，获取信号发射链路的信号状态，得到第一观测信息O(1)；以及，以I路信号的第二补偿系数为0，Q路信号的第二补偿系数为第二数值，对该信号发射链路进行第二次补偿，获取信号发射链路的信号状态，得到第二观测信息O(2)。步骤F3：获取第一差值与初始观测信息O(0)之间的比值，得到同相比值CI；以及，获取第二差值与初始观测信息O(0)之间的比值，得到正交比值CQ；其中，所述第一差值为初始观测信息O(0)与第一观测信息O(1)之间的差值，所述第二差值为初始观测信息O(0)与第二观测信息O(2)之间的差值。步骤F4：以第三观测信息O(3)为0为条件，根据同相比值CI和正交比值CQ确定同相信号的第三补偿系数h(3)和正交信号的第三补偿系数h(3)；
具体的，构建2*2矩阵，其中2*2矩阵中第一行记录有同相比值CI和正交比值CQ的实数部分的数值，第二行记录有同相比值CI和正交比值CQ的虚数部分的数值；
计算2*2矩阵的逆矩阵与1行2列的第一矩阵之间的乘积，得到1行2列的第二矩阵，其中所述第一矩阵中第一行的取值为初始观测信息O(0)的实数部分的数值，第二行的取值为初始观测信息O(0)的虚数部分的数值，其中第二矩阵的第一行为同相比值CI的取值，第二行为正交信号CQ的取值。
例如，一种针对LO leakage的多观测校准-补偿算法具体步骤(需要对每一次量测数据进行FFT变换，并获取Δf处的幅度和相位信息)：步骤1：当hDC,I＝hDC,Q＝0时，估计得到ODC(0)＝TxLOL·TF；其中，hDC,I表示I路信号的理想补偿系数，hDC,Q表示Q路信号的理想补偿系数。步骤2：当hDC,I＝DCI、hDC,Q＝0时，ODC(1)＝[TxLOL+DCI·αIcos(wmt+φI)]·TF。步骤3：当hDC,I＝0hDC,Q＝DCQ时，ODC(2)＝[TxLOL-DCQ·αQcos(wmt+φQ)]·TF。步骤4：将量测作差并归一化处理得到：
步骤5：补偿的本质：
在另一个可选的实施例中，针对HD3补偿的方法，若基带(BB)信号的初相位是x度，那么后续的3BB信号的初相需要是3x度，且O(n)代表3BB信号所对应频点的幅度相位值除以BB信号所对应频点的幅度相位值。具体的：步骤G1：以初始补偿系数h(0)为0，获取信号发射链路的信号状态，得到初始观测信息O(0)。步骤G2：利用初始观测信息O(0)确定的第一补偿系数h(1)对该信号发射链路进行第一次补偿，获取信号发射链路的信号状态，得到第一观测信息O(1)；以及，利用初始观测信息O(0)确定的第二补偿系数h(2)对该信号发射链路进行第二次补偿，获取信号发射链路的信号状态，得到第二观测信息O(2)；其中第二补偿系数h(2)和第一补偿系数h(1)的数值不同，第二观测信息O(2)和第一观测信息O(1)的数值不同。步骤G3：获取第一差值与初始观测信息O(0)之间的比值，得到第一比值d1；以及，获取第二差值与初始观测信息O(0)之间的比值，得到第二比值d2；其中，所述第一差值为初始观测信息O(0)与第一观测信息O(1)之间的差值，所述第二差值为初始观测信息O(0)与第二观测信息O(2)之间的差值。步骤G4：以第三观测信息O(3)为0为条件，根据第一比值d1和第二比值d2确定第一补偿系数h(1)对应的第一系数x1和第二补偿系数h(2)对应的第二系数x2。具体的，构建2*2矩阵，其中2*2矩阵中第一行记录有第一比值的实数部分的数值和虚数部分的数值，第二行记录有第二比值的实数部分的数值和虚数部分的数值；计算2*2矩阵的逆矩阵与1行2列的第一矩阵之间的乘积，得到1行2列的第二矩阵，其中所述第一矩阵中第一行的取值为1，第二行的取值为0，其中第二矩阵的第一行为第一系数x1的取值，第二行为第二系数x2的取值。步骤G5：计算第一系数x1与第一补偿系数h(1)之间的乘积，得到第一乘法结果；以及， 计算第二系数x2与第二补偿系数h(2)之间的乘积，得到第二乘法结果；将第一乘法结果和第二乘法结果作为第三补偿系数h(3)。
例如，步骤1：当hHD3(0)＝0时，估计得到
步骤2：当hHD3(1)＝b＝|OHD3(0)|时，
步骤3：当时，
步骤4：将量测作差并归一化处理得到：

步骤5：补偿的本质
基于本公开实施例中所记载的数字移相器架构的发射链路和/或接收链路的架构，除了应用于电磁波传感器中进行发射和/或接收信号外，还可针对不同发射/接收链路之间天线馈线不等长、TXLO(即LO与各发射通道中混频器之间)的馈线不等长的问题，以及将本公开实施例中所记载的数字移相器架构的发射链路和/或接收链路的架构作为辅助链路针对模拟移相器架构的接收链路和/或发射链路进行校准等操作。
需要说明的是，本公开中的电磁波包括无线电波和光波，无线电波包括短波、中波、长波和微波等，微波包括厘米波(即3GHz～30GHz的电磁波，如3.1GHz～10.6GHz的电磁波、24GHz频段的电磁波等)和毫米波(即30GHz～300GHz的电磁波，如60GHz频段的电磁波、77GHz频段(如77GHz～81GHz等)的电磁波)，光波则可包括紫外线、可见光、红外线、和激光等，其中激光的电磁波频段为(3.846～7.895)*10^5GHz，即激光包含于紫外线与可见光的部分频段内。
图15为本公开实施例中一种具有至少两个发射通道的发射链路的示意图，图16为本公开实施例中数字LO信号生成器的结构示意图，图17为本公开实施例中包含馈线不等长补偿模块的发射链路的结构示意图。
如图15所示，针对采用至少两个发射通道的发射链路，由于各个发射通道(或发射信道)会分别与同一个本振(Local Osillator，简称LO)源连接，从而使得本振信号到每个发射通道中的混频器(Mixer)之间的馈线长度(简称为TXLO)不相同，进而会导致不同的发射通道之间会产生相对的时延问题。另外，由于天线阵列中采用不等长发送天线馈线可以实现更好的链路预算，更加简单的天线设计和走线，更低的天线间耦合，更好的目标解角性能，更小的模组和成本等，所以采用不等长发送天线馈线的天线设计越来越被广泛的应用，尤其是针对近距离测量、相对封闭的空间区域(如室内或舱内灯)等应用场景，或者采用封装天线芯片(AiP)应用的各个场景中，如扫频较宽的应用场景。但是，由于不同发射通道之间发送天线馈线的长度不同，其会使得射频信号经过不同的发送天线产生了相对的时延问题。
针对上述在收发链路中存在因不同发射通道之间或接收通道之间馈线长度不同而导致的信号相对时延问题，本公开实施例基于上述数字移相架构提供了一种针对上述因馈线不等长而导致的相对时延的补偿方案，以有效提升发收信号的的质量。
具体的，如图15所示，数字移相器架构的发射天线阵列包括四个发射通道，各个发射通道共用波形控制器模块(waveform control)和本振源LO，针对任一发射通道均包括依次连接的发射直接数字频率合成器TX DDFS、IQ数模转换器DAC、模拟低通滤波器Analog LPF、IQ混频器Mixer、功率放大器PA及发射天线等，即波形控制器模块分别与各个发射通道中TX DDFS的连接，同时本振源LO分别连接至各个发射通道中的Mixer的一个输入端，以对各个发射通道中LPF输出的移相信号进行上混频后形成射频信号以经各发射天线进行发射。针对上述的本振源LO到各个发射通道中的Mixer之间的馈线长度不等长的问题，和/或各个发射通道的PA到其发射天线之间的馈线长度不等长的问题，均可在发射直接数字频率合成器TX DDFS进行预置补偿，以降低因上述馈线不等长所导致的发射信号的时延问题。
例如，针对本振源LO到各个发射通道中的Mixer之间的馈线长度不等长的问题，可以本振源LO到其中一个发射通道中的Mixer最短的馈线作为基准，然后获取其余各发射通道中Mixer到该本振源LO相较于上述基准的长度差，因为发射信号传播的速度c相对固定，所以通过将上述各个发射通道的长度差除以速度c即可得到对应的时延值，后续在发射信号时，直接在TX DDFS中进行对应的补偿即可各个发射通道相较于基准的时延。同时，针对各个发射通道的PA到其发射天线之间的馈线长度不等长的问题也可采用类似的方式，来获取并补偿各个发射通道之间馈线差异而导致的时延问题。
在针对发射FMCW波的发射链路进行具体进行补偿时，可通过利用数字LO生成器，基于因馈线长度相较于基准而导致的时延μ、扫频周期T、扫频带宽β、扫频起始频率fc等参数，并适配后续DAC的时钟信号来对上述时延进行补偿。如图16中所示，在DDFS中利用数字LO生成器Digital LO generator进行补偿，即其一路输入△θ＝μ*β/T，另一路输入θ0＝fcμ-μ2*β/2T；其中，μ为各发射通道对应馈线相较于基准馈线的时延差；同时，如图16-17所示，该数字LO生成器作为信号补偿器的输出LO-I(即cos(ωn+ψ))作为作为DAC中I路的输入，数字LO生成器的输出LO-Q(即sin(ωn+ψ))作为作为DAC中Q路的输入。
需要说明的是，图16至图17示意的是一个发射通道的结构示意图，而图15中则示出了具有四个发射通道的发射天线阵列，而在实际应用发射天线阵列只要包含有至少两个(如两个、三个或五个等)发射通道即可，也可在接收信号和/或信号处理过程中进行补偿操作；同时针对具有一个发射通道和/或一个接收通道的结构，也可利用类似的思想，直接在TX DFFS中或者在接收信号和/或信号处理过程中进行补偿即可。
针对将本公开实施例中所记载的数字移相器架构的发射链路和/或接收链路的架构的思路作为辅助链路针对模拟移相器架构的接收链路和/或发射链路、数字移相器架构的接收链路和/或发射链路进行校准等操作的相关技术方案，可参考图18至图21所示。
图18为本公开实施例中利用辅助电路对收发链路进行校准及补偿的示意图，图19为本公开实施例中利用辅助发射电路对接收链路进行校准及补偿的示意图。
具体的，信号收发链路中，一般可包括天线、射频模拟器件、基带模拟器件和基带数字处理器等，射频模拟器件可包括锁相环(PLL)、功率放大器(PA)、移相器(PS)、低噪声放大器(LNA)、混频器本振(LO)和功率检测器(PD)等，而基带模拟器件可包括低通滤波器(LPF)、高通滤波器(HPF)、模数转换器(ADC)等。
其中，模拟器件或电路中会存在多种多样的非理想状况，而且会随着温度的变化而变化，例如：发射链路(TX)中的射频模拟器件及电路、接收链路(RX)中的基带模拟器件及电路、接收链路(RX)中的射频模拟器件及电路等多处的频率响应均可能存在不平衡的缺陷，PA输出端口和LNA输入端口均可能存在的PD精度问题，接收链路(RX)中还可能存在本振泄漏(LO leakage)等问题。同时，当模拟器件工作在不同的频率时，其幅频响应也会不同，等同于在接收信号上加入了一个预料外的窗函数，进而会对后续的目标距离和速度估计等操作造成不利的影响，尤其是在多目标的场景中，甚至会导致检测出假目标或遗漏掉真实目标。另外，针对至少两个链路的情况，还会存在不同的接收链路之间的频率响应不同，进而会额外的给接收链路引入不同的相位和幅度等噪声信息，尤其是引入的相位会直接影响后续对目标的回波方向的估计，进而降低了目标角度估计的精度。
在一些的实施例中，可通过外部设备测量模拟器件及电路的误差，然后对其进行一定的补偿操作，如采用Bench校准或ATE校准等操作，来针对接收链路的模拟器件及电路性能进行校准。
本公开实施例还提供了基于辅助链路方式的发射链路和/接收链路校准的方案，以在实现精准校准的同时，实现实时地校准模拟器件及电路，同时无需外设设备参与，有效降低因环境改变而导致射频器件参数指标变动的影响。
如图18所示，针对电磁波信号的发收链路可通过设置辅助校准电路(Auxiliary)的方式对其进行实时的校准，该辅助校准电路(即辅助校准链路)可集成于电磁波传感器中，进而在实现对其进行实时精确校准的同时，同时无需外设设备参与，就能有效降低因环境改变而导致射频器件参数指标变动的影响。
例如，针对发射通路，就可在传感器中临近该发射通路的位置处设置辅助接收通路(Auxiliary Reveiver，简称ARX)，以实现对该发射通路的实时校准操作；类似的，针对接收通路，也可在传感器中临近该接收通路(即信号接收链路)的位置处设置辅助发射通路(Auxiliary Transmitter，简称ATX)，以实现对该接收通路的实时校准操作。另外，还可将辅助通路设置在两条待校准通路之间，这样还能复用一路辅助通路针对至少两个不同通路进行校准。如图18所示，在两个发射通路之间设置一条共用的ARX，而在两个接收通路之间设置一条共用的ATX，同时还能进一步的避免走线或线路过长等问题。在一个可选的实施例中，针对图18所示的四发四收的天线阵列，就可以设置两个ARX和两个ATX，即各个ARX均设置在两个发射通道之间，而各个ATX均设置在两个接收通道之间，以有效的降低走线的难度，同时还提升了校准的实时性。
如图18所示，针对接收通路，由于接收通路可划分为射频部分(即RF Rx)和基带部分(即Rx BB)，为了进一步提升校准的精度，还可针对RF Rx与Rx BB分别对应设置辅助校准通路，如针对RF Rx设置射频辅助发射单元(可简称为RF ATX)，针对Rx BB设置基带辅助发射单元(可简称为IF ATX)。另外，针对ARX通路，还可通过设置射频单音信号发射器(RF Tone Generator)来对ARX进行校准后，再利用校准后的ARX对发射通道(TX)进行校准操作。
在一些可选的实施例中，如图18所示，针对接收通路(Receiver)中的RF Rx和Rx BB，可先利用IF ATX对Rx BB进行校准后，利用校准后的Rx BB对RF ATX进行校准，最后利用校准后的RF ATX对Rx BB进行校准，从而实现对整个接收通路的校准操作。
基于图18所示的结构，结合图19所示的内容，针对接收通路(Receiver)校准方案进行说明，即该接收通道可包括依次连接的接收天线(或接收端口)、LNA、Mixer、TIA、LPF、HPF、Real ADC等器件，以及连接至LNA输入端口的PD等，并基于Mixer将可该接收通路划分为RF Rx与Rx BB，即沿信号传输的方向，将Mixer及其之前的部分定义为RF Rx，其余的部分则可定义为Rx BB；如图19所示，RF Rx可包括接收天线(或接收端口)、LNA、Real Mixer和PD等。
对应上述接收通路的Rx BB所设置的IF ATX可包括依次连接的分频器(如Freq Divider 1/N)和Real DAC(如1-bit Real DAC)，即Real DAC的输出端连接至Real Mixer的输出端。如图19所示，IF ATX可包括依次连接的分频器(如Freq Divider 1/N)和一个Bit的数模转换器(如1-bit Real DAC)，即该IF ATX可通过该分频器和数模转换器在多种环境下稳定地产生单音信号(单Tone)，且该单音信号还可被自由地配置为多种频率。
在一些可选的实施例中，为了提升校准的精度，还可在IF ATX对Rx BB进行校准前先对该IF ATX进行校准，例如针对IF ATX中的中频(IF)频率响应的校准、直流校准等。同时，在完成IF ATX的校准后，即可利用校准后的IF ATX对接收通路的Rx BB进行校准，如可通过校准后的IF ATX发射不同频率的单音信号，对Rx BB部门完成校准、补偿等操作。可选的，可基于接收通路中Real ADC的采样频率来确定IF ATX所发射单音信号的频率(如几十MHz)。
对应接收通道的RF Rx所设置的RF ATX则可包括依次连接的TX DDFS、IQ失衡补偿单元(TX IQ Comp)、加法器、IQ数模转换器(IQ DAC)、低通滤波器(LPF)、放大器(如PA)、乘法器、本地震荡器(LO，图中未示出)及平方器(x^2)，乘法器的输出端经平方器连接至接收通路中的Mixer的输出端形成第一校准支路，同时乘法器的输出端还连接至接收通路中PD与发射天线之间的链路中形成第二校准支路。其中，IQ失衡补偿单元可被配置为用于对RF ATX中IQ的不平衡的补偿，而加法器则可被配置为用于补偿RF ATX中本振泄漏(LO Leakage)，同时平方器则可被配置为用于补偿因RF ATX中IQ的不平衡带来的残留边带效应。
在一些可选的实施例中，为了提升校准的精度，还可在RF ATX对Rx ATX进行校准前先对该RF ATX进行校准，例如针对RF ATX中的本振泄漏、射频频率响应、IQ不平衡等问题的校准操作。同时，在完成RF ATX的校准后，再利用校准后的RF ATX对接收通路的RF Rx进行校准，如可通过校准后的RF ATX发射不同频率的单音信号，然后通过在接收链路的基带信号处理模块，来校准、补偿接收通路射频部分RF Rx中对LNA的输入端的功率检测器(PD)、接收通路LNA到ADC的总增益以及频率响应的辅助校准等。
在一些可选的实施例中，利用IF ATX、RF ATX对接收通路(Receiver)进行校准时，可先对IF ATX进行校准，并利用校准后的IF ATX对接收通路中的Rx BB进行校准，然后利用校准后的Rx BB通过RF ATX的第一校准支路(即通过平方器)对该RF ATX进行校准，最后再利用校准后的RF ATX中的第二校准支路实现针对接收通路中的RF Rx的校准，进而实现对整个接收通路的校准及补偿操作等。
具体的，如图19所示，可先通过1-bit Real DAC校准Rx BB及Real DAC等，然后再通过校准后的RxBB校准RF ATX中的LO泄漏及IQ不平衡，最后可利用校准后的RF ATX发射信号从接收通路中的LNA输入，用以校正RF Rx的RF LO泄漏和RF频响等。
图20为本公开实施例中利用辅助接收电路对发射链路进行校准及补偿的示意图。如图20所示，发射通路(Transmitter)可包括移相模块PS、放大器PA、功率检测器PD等，例如该发射通路可采用本公开任一实施例所阐述的数字移相架构(Digital Phase Shifter)的发射链路，具体可参见相关图示及文字描述，在此便不予赘述。由于发射通路采用的是数字移相器架构，以在实现更加精准的移相操作的同时，使得发射通道能够同时支持多天线的DDM、FDM(Frequency Division Multiplexing，频分复用)等多个模式，还可省去RF移相器(Phase Shifter)的校准操作，降低移相系统中隔离度和耦合度，减小链路损耗及生产制造成本。另外，针对所引入可能存在的TX IQ失配(Mismatch)、LO泄漏(leakage)等问题，该数字移相器架构的发射通路还可支持在数字域进行RF频率响应(Frequency Response)补偿、IQ不平衡和LO泄漏的校准操作等。
针对发射通路中存在的TX IQ失配(Mismatch)、LO泄漏(leakage)、频率响应等问题，可通过设置辅助接收通路(Auxiliary Receiver，简称ARX)来进行相关的校准及补偿操作。如图20所示，ARX可包括依次连接的混频器Mixer、TIA、LPF、HPF、IQ ADC、加法器和RF校准模块(RF Calib)，即通过将混频器的一个输入端接收ARX IQ LO信号，另一个输入端则在沿信号传输方向(即图中所示的箭头方向)连接至发射通路PD之前的节点，或者在移相器(模块)之后的任一节点。例如连接至PA的输出端(同步实现对PA的校准)、PA的输入端等，以通过该ARX对发射通路进行校准操作。其中，发射通路中的LO信号频率与ARX IQ LO信号频率之间具有设定的差频，以使得该两个信号之间存在错频，以模拟真实的收发信号回路。
在一个可选的实施例中，为了进一步提升校准精度，针对ARX(即辅助接收单元)还可设置对应的校准电路(即校准接收单元)，例如图20所示的RF单音信号生成电路(RF Tone Generator)，可包括依次连接的TX DDFS、加法器、Real DAC、LPF、放大器、乘法器和带通滤波器(Band Pass Filter，简称BPF)，其中的加法器可被配置用于对TX LO泄漏(TX LO leakage Waveform)进行校准补偿，而乘法器则可被配置用于补偿RF Tone Gen LO泄漏(leakage)，BPF可被配置用于滤除RF Tone Generator的LO leakage所产生的直流信号。即RF Tone Generator可被配置用于产生多个稳定的不同频率的单音(Tone)信号，以实现对ARX的校准操作。
在一些可选的实施例中，如图20所示，可先利用RF Tone Generator对ARX进行校准，然后利用校准后的ARX对包括诸如PA在内的发射通道(Transmitter)的校准，例如对PA输出端的PD、发射通道中的移相器、DAC到PA输出的总增益以及频率响应等器件及电路的校准。
具体的，如图20所示，可先通过利用RF Tone Generator产生多个稳定的且不同频率的单音信号以辅助校准ARX，然后基于校准后的ARX来辅助校准发射通路TX的IQ不平衡、本振泄漏、频率响应不一致等问题。
图21为本公开实施例中一种辅助电路的结构示意图，图22为本公开实施例中另一种辅助电路的结构示意图，图23为本公开实施例中IQ Mixer的电路模块示意图；图24为本公开实施例中一种IQ Mixer的结构示意图，图25为对应图24中所示结构的示意图。
在一些可选的实施例中，基于图18至图19所示的结构及相关描述，ATX还可以通过平方器结合多个bit的DAC来实现。如图21所示，接收通路包括依次连接的LNA、混频器、TIA、HPF、ADC和BB处理模块(BB Processor)，以及本振源LO，即混频器利用本振源提供的第一LO信号对LNA提供的回波信号进行下降频处理后得到中频信号(即模拟基带信号)。ATX则可包括依次连接的第一DAC(即DAC1)、混频器和平方器(x^2)，其中第一DAC提供模拟信号至ATX中的混频器接收端，该混频器的另一接收端可连接接收通路中的本振源，以利用该本振源提供的第二LO信号对第一DAC输出的模拟回波信号进行混频处理，进而输出模拟中频信号至平方器的输入端，平方器将处理后的模拟中频信号连接至TIA的输入端，以对接收通路中的TIA、HPF、ADC等电路及元器件进行校准。在另外一些可选的实施例中，ATX还可通过DAC输出模拟回波信号直接至平方器的接收端，该平方器将处理后的模拟回波信号发送至LNA的输入端，以实现对接收通路中的LNA、混频器、TIA、HPF、ADC等电路及元器件进行校准。
在一些可选的实施例中，如图22所示，基于图21所示结构的基础上，针对接收通路中包含有VGA，以及包含有SDM单元和抽取滤波单元(Decimation filter)的ADC，还可设置第二DAC，且该第二DAC的输出端可连接至接收通路中混频器与TIA之间、TIA与HPF之间、HPF与VGA之间、VGA与ADC之间中任一节点，以对对应的电路及元器件的进行校准。可选的，第一DAC可为多bit(如10-bit)DAC，第二DAC则可为1-bit DAC。
上述基于DAC与平方器构成的ATX，本领域技术人员可基于上述记载的内容进行其他合理的变化，具体的实现方式本公开并不予限制，只要其能实现本公开实施例中ATX相同或近似的功能即可。
在一些可选的实施例中，在接收通路附近设置ATX时，可如利用接收通路之间的空隙设置ATX，同时还可实现两个及以上的接收通路公共一个ATX；同样，在发射通路附近设置ARX时，可如利用发射通路之间的空隙设置ARX，同时也还可实现两个及以上的发射通路公共一个ARX。可选的，上述的ATX和/或ARX可进行间歇性工。例如，ATX可在接收通路的工作的间隙(如帧间或chirp之间)，发射校准用的单音信号，以对接收机进行实时有效地校准和补偿等操作。类似的，ARX也可在接收机工作的间隙(如帧间或chirp之间)，先利用Tone generator产生校准用的单音信号，先对ARX进行有效地校准。然后再通过校准后的ARX对发射机进行有效的校准，从而确保可按照预设的方式进行校准及补偿等操作
针对图18至图22所示的基于辅助链路方式实现发射通路和/或接收通路校准的方案，基于采用辅助链路的方式，通过在诸如芯片(chip or die)等集成电路中加入辅助电路，以辅助校准主通路电路，来有效提升模拟电路及模块的性能，同时还可对至少部分射频电路及器件能以线上(on the fly)的方式进行实时校准，从而有效提升校准的性能，进而能够有效的射频性能的同时降低射频实现的难度。
由于混频器是收发链路用于变频的一个非常重要及关键的器件，其被广泛的应用于通信、雷达等无线电的器件中，例如针对镜像信号的具有良好抑制效果的单边带混频器，以及本公开实施例中收发链路中的IQ混频器等。其中，IQ混频器(如单边带混频器)包含有相位差为90°的I支路和Q支路两个支路来传输信号。
针对版图面积相对较为紧凑的应用场景中，例如在毫米波频段等高频的传感器应用中，I支路与Q支路之间、混频前的输入支路与输出支路之间等均因物理距离较短，从而会导致电磁波信号可以通过诸如磁耦合、衬底耦合、电耦合等方式，在支路之间、输入输出端口之间、IQ匹配网络与混频器输出之间等均可能存在信号的泄漏(leakage)的问题，进而致使混频器的镜像抑制比、本振泄露等恶化严重。
在一可选的实施例中，基于混频器输出网络在确保IQ两支路功率合成，同时还满足阻抗匹配的要求，本公开提供一种新的混频器结构，即通过针对混频器输出无源网络的改进，来有效提升IQ两支路之间隔离度的同时，有效的降低本振信号的泄漏。
图23为本公开实施例中IQ Mixer的电路模块示意图。如图23所示，一种IQ混频器，可包括I支路混频单元、Q支路混频单元和变压器单元，其中I支路混频单元可被配置用于输出I路信号，Q支路混频单元可被配置用于输出Q路信号，变压器单元则可被配置用于对I路信号和Q路信号进行磁耦合以合成IQ混频输出信号，并将该IQ混频输出信号传输至后级电路(Next Block)。其中，变压器单元可设置在I支路混频单元与Q支路混频单元之间，以相对短接式的IQ混频器，增加I支路混频单元与Q支路混频单元之间的物理距离，进而有效的减小I支路混频单元与Q支路混频单元之间的耦合，达到提升IQ混频器抑制比的目的，同时还便于版图的布局设计。
短接式的IQ混频器是通过将混频器IQ两个支路的输出分别短接后，再通过一个匹配网络(Matching Networks)匹配到后级电路，即通过电耦合的方式进行信号合成，而如图23所示，本公开一个可选的实施例的IQ混频器中，I支路混频单元的输出端之间、Q支路混频单元的输出端之间则可通过分别串联一个支路电感，后续在两个支路电感之间再设置一个磁耦合电感，进而形成一个三圈变压器结构，即通过磁耦合的方式将I支路混频单元的输出与Q支路混频单元的输出进行磁耦合合成，进而得到上述的IQ混频输出信号；其中I支路混频单元和Q支路混频单元的共模不是直接相通，所以可以分开调节，使得器件的应用场景更加灵活；然后，通过上述的磁耦合电感的两端作为输出端直接传输至后级电路(Next Block)中。即该三圈变压器结构具有四个输入端和两个输出端。
基于图23所示的三圈变压器结构，若LO信号和待混频信号(如回波信号)进入三圈变压器的IQ两支路各自的P、N端后，在三圈变压器上形成的电流方向和磁场方向如图24所示，此时Isb磁场叠加，usb磁场抵消，进而就可以输出高质量的单边带信号。即基于该三圈变压器结构，可实现对待混频信号的下混频操作或者上混频操作。
一些可选的实施例中，即便LO信号与待混频信号之间存在相位差的场景中，由于信号在三圈变压器上的抵消方式一致，在LO信号和待混频信号本身不存在失调的情况下，三圈变压器输出的仍然为单边带信号。
本公开实施例中提供的磁耦合式的IQ混频器，相较于采用电耦合方式进行功率合成的短接式的IQ混频器，该磁耦合式的IQ混频器是通过磁耦合的方式进行功率合成。同时，对于I、Q两个支路，磁耦合式的IQ混频器可以完全一致，且可呈现轴对称，不会因I、Q两个支路因走线长度不一致而引入额外的相位误差，进而可使得相关的电路(如毫米波等高频电路)在后续的制造工艺更加鲁棒，且抵抗工艺不稳定性的强度更高。例如，本公开实施例中的磁耦合式的IQ混频器，如图25所示可将变压器抽头设置设置P、N端口之间的中线上，以使得混频器的结构可基于该中线呈轴对称，从而使得P端口和N端口的共模通路长度相等，即磁耦合式的IQ混频器中PN的对称性要优于短接式的IQ混频器中PN的对称性，进而使得本振信号的泄露会更小。
在一些可选的实施例中，本公开实施例提供的磁耦合式的IQ混频器，可应用于发射通道和片上自检等各种电路中，作为各个电磁波电路中的上变频混频器等。例如本公开实施例中各个发射通道(或发射链路)、BIST中的IQ混频器就可采用本公开实施例中的磁耦合式的IQ混频器。
图26为本公开实施例中另一种IQ Mixer的物理结构示意图。如图26所示，本公开实施例还提供了另一种磁耦合式的IQ混频器的结构，即I、Q支路的抽头馈线存在部分的交叠，如针对I支路而言，该I支路的抽头可从Q支路路一侧绕过(如通过通孔via)并延伸至I支路一侧后，再接入共模偏置电压，同样的，Q支路的抽头可从I支路一侧绕过(如也可通过通孔via)并延伸至Q支路一侧后，再接入共模偏置电压。上述的走线方式，可使得共模通路的路径更加确定，且更不容易受到其他电路的干扰。同时，本公开实施例中的IQ Mixer中的三圈变压器的可为方形、八边形、正八边形等，即只要其在俯视图或仰视图中能实现沿NP之间的中心呈轴对称分布即可。
如图27所示，本公开实施例还提供了一种信号发射方法，应用于具有至少一个信号发射链路的电磁波器件中，该信号发射方法包括：步骤2701、确定每个信号发射链路的射频发射信号的相位；步骤2702、根据所述射频发射信号的相位确定每个所述信号发射链路中的基带信号的初始相位；步骤2703、根据确定的初始相位，生成初始基带信号；步骤2704、利用预先获取的补偿信息对所述初始基带信号进行补偿后，得到所述基带信号；步骤2705、基于所述基带信号对发端本振信号进行移相操作，得到所述射频发射信号。
本公开实施例的信号发射方法，通过使用数字移相器架构在数字域产生数字基带信号，有更好的正交性和更低的旁瓣，其移相相位可以非常准确产生，使得调相精度更高，从而实现了带有高精度的数字移相功能的车载雷达系统，降低了天线之间隔离度的需求，同时具有链路损耗小、成本低、无需离线校准的优点，并且可以支持更为灵活的发波方案，比如高性能的多普勒分复用和频分复用等，且能够支持在数字域进行频率响应补偿。
如图28所示，本公开实施例还提供了一种信号发射方法，应用于具有至少一个信号发射链路的电磁波器件中，该信号发射方法包括：步骤2801、获取当前补偿系数下信号发射主通路的观测信息，其中所述信号发射主通路用于根据补偿系数对产生的信号进行补偿操作后生成射频发射信号，以用于实现目标探测和/或通信。步骤2802、在当前的观测信息满足迭代条件时，将当前的补偿系数作为所述信号发射链路补偿操作使用的补偿系数；否则，对当前的补偿系数进行迭代直到所得到的观测信息满足所述迭代条件为止。步骤2803、利用所述补偿系数对基带信号进行补偿。步骤2804、基于补偿后的基带信号对发端本振信号进行移相操作，得到射频发射信号。
本公开实施例的信号发射方法，本公开实施例的信号校准链路、发射链路及方法、收发链路、集成电路，信号校准链路产生的补偿信息能够对产生的信号进行补偿，解决正交失衡、LO泄漏以及谐波失真问题，从而有效提升信号发射主通路中信号质量。通过迭代方式确定信号发射主通路所需的补偿系数，可以有效提示功能补偿系数的准确性。
如图29所示，本公开实施例还提供了一种信号发射方法，应用于具有至少一个信号发射链路的电磁波器件中，该信号发射方法包括：步骤2901、确定在数值不同的初始补偿系数h(0)、第一补偿系数h(1)以及第二补偿系数h(2)的条件下信号发射主通路对应的初始观测信息O(0)、第一观测信息O(1)以及第二观测信息O(2)，其中所述信号发射主通路用于根据补偿系数对产生的信号进行补偿操作后生成射频发射信号，以用于实现目标探测和/或通信。步骤2902、利用初始观测信息O(0)、第一观测信息O(1)和第二观测信息O(2)， 确定第三补偿系数h(3)。步骤2903、利用所述补偿系数对基带信号进行补偿。步骤2904、基于补偿后的基带信号对发端本振信号进行移相操作，得到射频发射信号。
本公开实施例的信号发射方法，本公开实施例的信号校准链路、发射链路及方法、收发链路、集成电路，信号校准链路产生的补偿信息能够对产生的信号进行补偿，解决正交失衡、LO泄漏以及谐波失真问题，从而有效提升信号发射主通路中信号质量。通过多次校准操作得到观测信息，从而确定信号发射主通路所需的补偿系数，可以有效提示功能补偿系数的准确性。
在一些示例性实施方式中，发端数字基带信号为单音信号，发端本振信号为扫频信号；或者，发端数字基带信号为扫频信号，发端本振信号为单音信号。在一些示例性实施方式中，扫频信号的频带宽度在2GHz以上。
本公开实施例还提供了一种集成电路，可包括依次连接的射频模块、模拟信号处理模块和数字信号处理模块；射频模块用于产生射频发射信号和接收射频接收信号；模拟信号处理模块用于对射频接收信号进行降频处理以得到中频信号；数字信号处理模块用于对中频信号进行模数转换以得到数字信号；以及射频模块包括上述任一项的信号发射链路、上述的信号收发链路、上述任一项的信号校准链路、上述任一项的信号补偿链路、上述任一项的信号校准系统，和/或上述的IQ混频器；和/或数字信号处理模块基于上述的馈线不等长补偿方法在数字域进行补偿；其中，射频接收信号为射频发射信号被目标发射和/或散射所形成的回波信号。可选的，集成电路还可包括数据处理模块，数据处理模块用于对数字信号进行处理以实现目标探测和/或无线通信，例如该集成电路可为毫米波雷达芯片(chip or die)。
在一些可选的实施例中，所述集成电路可为AiP(Antenna-In-Package，封装内天线)芯片结构、AoP(Antenna-On-Package，封装上天线)芯片结构、AoC(Antenna-On-Chip，片上天线)芯片结构或RoP(Radiator on Package，即通过在封装(Package)上设置辐射结构(Radiator)，并在Radiator周围用球围成波导结构，RF信号通过辐射结构过渡到该波导结构中，再由波导结构转换到外接天线)结构。
根据本公开的另一些实施例，还提出一种电磁波传感器。该电磁波传感器可包括天线，以及如前所述的集成电路。其中，集成电路与天线电连接，用于收发电磁波信号。例如，该电磁波传感器可包括：承载体、如上述任一实施例所述的集成电路和天线等，所述集成电路可设置在承载体上；天线可设置在承载体上，或者与所述集成电路集成为一体器件设置在所述承载体上(即此时该天线可为AiP、AoP或AoC结构中所设置的天线)；其中，所述集成电路与天线连接(即此时传感芯片或集成电路未集成有天线，如常规的SoC等)，用于收发电磁波信号。其中，承载体可以为印刷电路板PCB，第一传输线可以为PCB走线。
本公开实施例提供了一种设备，可包括：设备本体；以及设置于设备本体上的如上述的电磁波传感器；其中，电磁波传感器用于目标检测和/或通信，以向设备本体的运行提供参考信息。
本公开实施例还提供了一种电子设备，该电子设备可以通用计算设备的形式表现。电子设备的组件可以包括但不限于：至少一个处理单元、至少一个存储单元、连接不同系统组件(包括存储单元和处理单元)的总线、显示单元等。其中，存储单元存储有程序代码，程序代码可以被处理单元执行，使得处理单元执行本说明书描述的根据本公开各种示例性实施方式的方法。存储单元可以包括易失性存储单元形式的可读介质，例如随机存取存储单元(RAM)和/或高速缓存存储单元，还可以进一步包括只读存储单元(ROM)。
存储单元还可以包括具有一组(至少一个)程序模块的程序/实用工具，这样的程序模块包括但不限于：操作系统、一个或者多个应用程序、其它程序模块以及程序数据，这些示例中的每一个或某种组合中可能包括网络环境的实现。
总线可以为表示几类总线结构中的一种或多种，包括存储单元总线或者存储单元控制器、外围总线、图形加速端口、处理单元或者使用多种总线结构中的任意总线结构的局域总线。
电子设备也可以与一个或多个外部设备(例如键盘、指向设备、蓝牙设备等)通信，还可与一个或者多个使得用户能与该电子设备交互的设备通信，和/或与使得该电子设备能与一个或多个其它计算设备进行通信的任何设备(例如路由器、调制解调器等等)通信。这种通信可以通过输入/输出(I/O)接口进行。并且，电子设备还可以通过网络适配器与一个或者多个网络(例如局域网(LAN)，广域网(WAN)和/或公共网络，例如因特网)通信。网络适配器可以通过总线与电子设备的其它模块通信。应当明白，尽管图中未示出，可以结合电子设备使用其它硬件和/或软件模块，包括但不限于：微代码、设备驱动器、冗余处理单元、外部磁盘驱动阵列、RAID系统、磁带驱动器以及数据备份存储系统等。
例如，本公开实施例中的电子设备还可包括：设备本体；以及设置于设备本体上的如上述任一实施例中所阐述的电磁波传感器；其中，该电磁波传感器可用于实现目标检测和/或无线通信等功能。
具体地，在上述实施例的基础上，在本公开的一个可选的实施例中，电磁波传感器可以设置在设备本体的外部，或者设置在设备本体的内部，而在本公开的其他可选的实施例中，电磁波传感器还可以一部分设置在设备本体的内部，一部分设置在设备本体的外部。本公开实施例对此不作限定，具体可视情况而定。
在一个可选的实施例中，上述设备本体可为应用于诸如智慧城市、智能住宅、交通、智能家居、消费电子、安防监控、工业自动化、舱内检测(如智能座舱)、医疗器械及卫生保健等领域的部件及产品。例如，该设备本体可为智能交通运输设备(如汽车、自行车、摩托车、船舶、地铁、火车等)、安防设备(如摄像头)、液位/流速检测设备、智能穿戴设备(如手环、眼镜等)、智能家居设备(如扫地机器人、门锁、电视、空调、智能灯等)、各种通信设备(如手机、平板电脑等)等，以及诸如道闸、智能交通指示灯、智能指示牌、交通摄像头及各种工业化机械臂(或机器人)等，也可为用于检测生命特征参数的各种仪器以及搭载该仪器的各种设备，例如汽车舱内生命特征检测、室内人员监控、智能医疗设备、消费电子设备等。
本公开实施例还提供了一种非瞬时性计算机可读存储介质，其上存储有计算机可读指令，当指令被处理器执行时，使得处理器执行如上述的馈线不等长补偿方法。
通过以上的实施方式的描述，本领域的技术人员易于理解，这里描述的示例实施方式可以通过软件实现，也可以通过软件结合必要的硬件的方式来实现。根据本公开实施方式的技术方案可以以软件产品的形式体现出来，该软件产品可以存储在一个非易失性存储介质(可以是CD-ROM，U盘，移动硬盘等)中或网络上，包括若干指令以使得一台计算设备(可以是个人计算机、服务器、或者网络设备等)执行根据本公开实施方式的上述方法。
软件产品可以采用一个或多个可读介质的任意组合。可读介质可以是可读信号介质或者可读存储介质。可读存储介质例如可以为但不限于电、磁、光、电磁、红外线、或半导体的系统、装置或器件，或者任意以上的组合。可读存储介质的更具体的例子(非穷举的列表)包括：具有一个或多个导线的电连接、便携式盘、硬盘、随机存取存储器(RAM)、只读存储器(ROM)、可擦式可编程只读存储器(EPROM或闪存)、光纤、便携式紧凑盘只读存储器(CD-ROM)、光存储器件、磁存储器件、或者上述的任意合适的组合。
计算机可读存储介质可以包括在基带中或者作为载波一部分传播的数据信号，其中承载了可读程序代码。这种传播的数据信号可以采用多种形式，包括但不限于电磁信号、光信号或上述的任意合适的组合。可读存储介质还可以是可读存储介质以外的任何可读介质， 该可读介质可以发送、传播或者传输用于由指令执行系统、装置或者器件使用或者与其结合使用的程序。可读存储介质上包含的程序代码可以用任何适当的介质传输，包括但不限于无线、有线、光缆、RF等等，或者上述的任意合适的组合。
可以以一种或多种程序设计语言的任意组合来编写用于执行本公开操作的程序代码，程序设计语言包括面向对象的程序设计语言-诸如Java、C++等，还包括常规的过程式程序设计语言-诸如“C”语言或类似的程序设计语言。程序代码可以完全地在用户计算设备上执行、部分地在用户设备上执行、作为一个独立的软件包执行、部分在用户计算设备上部分在远程计算设备上执行、或者完全在远程计算设备或服务器上执行。在涉及远程计算设备的情形中，远程计算设备可以通过任意种类的网络，包括局域网(LAN)或广域网(WAN)，连接到用户计算设备，或者，可以连接到外部计算设备(例如利用因特网服务提供商来通过因特网连接)。
上述计算机可读介质承载有一个或者多个程序，当上述一个或者多个程序被一个该设备执行时，使得该计算机可读介质实现前述功能。
本领域技术人员可以理解上述各模块可以按照实施例的描述分布于装置中，也可以进行相应变化唯一不同于本实施例的一个或多个装置中。上述实施例的模块可以合并为一个模块，也可以进一步拆分成多个子模块。
根据本公开的实施例，提出一种计算机程序，包括计算机程序或指令，该计算机程序或指令被处理器执行时，可以执行以上描述的方法。在一个可选的实施例中，上述集成电路可以为毫米波雷达芯片。集成电路中的数字功能模块的种类可以根据实际需求确定。例如，在毫米波雷达芯片，数据处理模块可以用于诸如距离维多普勒变换、速度维多普勒变换、恒虚警检测、波达方向检测、点云处理等，用于获取目标的距离、水平角、俯仰角、速度、高度、微多普勒运动特性、形状、尺寸、表面粗糙度及介电特性等信息。
需要说明的是，无线电器件可通过发射及接收无线电信号实现诸如目标检测和/或通信等功能，以向设备本体提供检测目标信息和/或通讯信息，进而辅助甚至控制设备本体的运行。
例如，当上述的设备本体应用于先进驾驶辅助系统(即ADAS)时，作为车载传感器的无线电器件(如毫米波雷达)则可辅助ADAS系统实现诸如自适应巡航、自动刹车辅助(即AEB)、盲点检测预警(即BSD)、辅助变道预警(即LCA)、倒车辅助预警(即RCTA)、泊车辅助、后方车辆示警、防撞、行人探测等应用场景。
以上所述实施例的各技术特征可以进行任意的组合，为使描述简洁，未对上述实施例中的各个技术特征所有可能的组合都进行描述，然而，只要这些技术特征的组合不存在矛盾，都应当认为是本说明书记载的范围。
以上所述实施例仅表达了本发明的较佳实施例及所运用技术原理，其描述较为具体和详细，但并不能因此而理解为对发明专利范围的限制。对本领域技术人员来说能够进行各种明显的变化、重新调整和替代而不会脱离本发明的保护范围。因此，虽然通过以上实施例对本发明进行了较为详细的说明，但是本发明不仅仅限于以上实施例，在不脱离本发明构思的情况下，还可以包括更多其他等效实施例，而本发明专利的保护范围由所附的权利要求范围决定。

Claims (33)

  1. 一种信号发射链路，其特征在于，应用于电磁波传感器中，所述发射链路包括模拟信号源和数字移相器，所述模拟信号源可被配置为用于提供初始模拟信号，所述数字移相器可被配置为用于提供在数字域产生移相信号，并基于该移相信号对所述初始模拟信号进行移相，以将所述初始模拟信号进行预设的移相操作。
  2. 如权利要求1所述的信号发射链路，其特征在于，还包括发射天线，所述发射天线可被配置用于将移相后的初始模拟信号向预设空间区域辐射。
  3. 如权利要求1所述的信号发射链路，其特征在于，所述数字移相器包括数字移相信号源、数模转换器和混频器，所述数字移相器可被配置用于产生数字移相信号，所述数模转换器可被配置用于将接收的所述数字移相信号转换为模拟移相信号，所述混频器可被配置用于利用所接收的所述模拟移相信号对所接收的所述初始模拟信号进行混频操作，以将所述初始模拟信号进行预设的移相操作。
  4. 如权利要求3所述的信号发射链路，其特征在于，所述数字移相信号源包括直接数字频率合成器，所述数模转换器为IQ数模转换器，所述混频器为IQ混频器。
  5. 根据权利要求1所述的信号发射链路，其特征在于，所述数字移相信号为单音信号，所述初始模拟信号为扫频信号；或者，
    所述数字移相信号为扫频信号，所述初始模拟信号为单音信号。
  6. 如权利要求1至5中任一项所述的信号发射链路，其特征在于，所述信号发射链路发射的是调频连续波信号。
  7. 一种信号发射链路，其特征在于，包括集成于同一集成电路中的信号发射主通路和信号校准链路；其中：
    所述信号校准链路被配置为用于对所述信号发射主通路进行校准以获取补偿信息；以及所述信号发射主通路被配置为用于根据所述补偿信息进行补偿操作后生成射频发射信号，以用于实现目标探测和/或通信。
  8. 根据权利要求7所述的信号发射链路，其特征在于，所述补偿信息包括谐波失真补偿参数、本振泄漏补偿参数和正交失衡补偿参数中的至少一个。
  9. 根据权利要求7所述的信号发射链路，其特征在于：
    所述信号发射主通路包括第一信号源和移相器；其中，所述第一信号源被配置为生成第一模拟信号；以及所述移相器被配置为对所述第一模拟信号进行频率搬移和/或移相，以形成射频发射信号。
  10. 根据权利要求9所述的信号发射链路，其特征在于：
    在所述移相器为非正交架构时，所述移相器包括第二信号源和发端混频器，其中所述第二信号源被配置为用于生成第二模拟信号，所述发端混频器被配置为对所述第一模拟信号和所述第二模拟信号进行混频处理，以形成所述射频发射信号；
    在所述移相器为正交架构时，其中所述移相器包括：第二信号源、数模转换模块和发端混频器；其中所述第二信号源被配置为生成第一数字信号；所述数模转换模块被配置为将所述第一数字信号转换为第二模拟信号；以及所述发端混频器被配置为基于所述第二模拟信号对所述第一模拟信号进行频率搬移和/或移相，以形成所述射频发射信号。
  11. 根据权利要求10所述的信号发射链路，其特征在于，所述发射主通路还包括补偿电路，其中所述补偿电路的信号输入端与所述第二信号源相连，信号输入端与所述移相器相连，所述补偿电路用于将补偿信号和第二信号源输出的信号进行合并处理后再输出。
  12. 一种信号收发链路，其特征在于，包括如权利要求1至11中任一项所述信号发射链路，以及信号接收链路；
    所述信号接收链路包括收端混频器、模数转换器和数字信号处理模块；
    其中，所述收端混频器可被配置为用于基于所接收的收端本振信号对所接收的回波信号进行下降频以得到模拟中频信号，所述模数转换器可被配置为用于对所接收的所述中频信号进行模数转换以得到数字中频信号，所述数字信号处理模块可被配置为用于对所数字中频信号进行处理以得到目标参数；
    所述回波信号为所述信号发射链路所发射信号被目标物反射和/或散射而形成的信号。
  13. 根据权利要求12所述的信号收发链路，其特征在于，所述收端混频器为实数混频器，所述模数转换器为实数模数转换器；或者，
    所述收端混频器为正交混频器，所述模数转换器为正交模数转换器。
  14. 根据权利要求12所述的信号收发链路，其特征在于，所述收端本振信号为扫频信号；或者，所述收端本振信号为单音信号。
  15. 一种信号校准链路，其特征在于，包括如权利要求12所述的信号收发链路；
    所述信号接收链路的接收天线连接端口连接至所述信号发射链路的发射天线连接端口，所述信号接收链路可被配置为用于对所述信号发射链路进行校准。
  16. 如权利要求15所述的信号校准链路，其特征在于，所述信号接收链路的本振信号与所述信号发射链路的本振信号之间具有预设的差频。
  17. 如权利要求16所述的信号校准链路，其特征在于，还包括BIST模块，所述BIST模块设置在本振信号源与所述收端混频器之间；
    其中，所述BIST模板可被配置为基于预设的频偏信号对接收的本振信号进行混频，以使得所述收端混频器接收到的本振信号与所述信号发射链路的本振信号之间具有预设的差频。
  18. 一种信号校准链路，其特征在于，包括如权利要求12所述的信号收发链路，以及BIST模块；
    所述信号接收链路的接收天线连接端口通过所述BIST模块连接至所述信号发射链路的发射天线连接端口，所述信号接收链路可被配置为用于对所述信号发射链路进行校准。
  19. 一种信号校准链路，其特征在于，包括两条信号接收链路、BIST模块、辅助电路单元和如权利要求1至11中任一项所述信号发射链路，以及；
    任一条所述信号接收链路均包括实数混频器、实数模数转换器和数字信号处理模块；所述实数混频器可被配置为用于基于所接收的本振信号对所接收的回波信号进行下降频以得到模拟中频信号，所述实数模数转换器可被配置为用于对所接收的所述中频信号进行模数转换以得到数字中频信号，所述数字信号处理模块可被配置为用于对所数字中频信号进行处理以得到目标参数；所述回波信号为所述信号发射链路所发射信号被目标物反射和/或散射而形成的信号；
    两条所述信号接收链路的接收天线连接端口依次通过所述辅助电路单元和所述BIST模块分别连接至所述信号发射链路的发射天线连接端口，所述信号接收链路可被配置为用于对所述信号发射链路的中频部分进行校准。
  20. 一种信号发射主通路的信号校准链路，其中所述信号发射主通路用于根据补偿系数对产生的信号进行补偿操作后生成射频发射信号，以用于实现目标探测和/或通信，其中：
    所述信号校准链路，被配置为用于获取当前补偿系数下信号发射主通路当前的观测信息；以及，在当前的观测信息满足迭代条件时，将当前的补偿系数作为所述信号发射链路补偿操作使用的补偿系数；否则，对当前的补偿系数进行迭代直到所得到的观测信息满足所述迭代条件为止。
  21. 根据权利要求20所述的信号校准链路，其特征在于，所述补偿系数包括谐波失真补偿参数、本振泄漏补偿参数和正交失衡补偿参数中的至少一个。
  22. 根据权利要求20所述的信号校准链路，其特征在于，所述信号发射主通路和所述信号校准链路集成于同一集成电路中。
  23. 一种信号补偿链路，其特征在于，包括如权利要求1至11中任一项所述信号发射链路，以及补偿单元，所述补偿单元可被配置为用于对所述信号发射链路的IQ失配、IQ失衡、信号泄漏、谐波失真缺陷中的至少一个进行补偿。
  24. 如权利要求23所述的信号补偿链路，其特征在于，所述补偿单元可被配置为用于基于如权利要求15至22中任意一项所述的信号校准链路所得到校准数据对所述信号发射链路进行补偿。
  25. 如权利要求24所述的信号补偿链路，其特征在于，所述信号校准链路中的所述信号接收链路作为辅助校准模块集成于待校准的发射链路附近区域，以在所述发射链路工作的间隙进行实时校准操作。
  26. 一种馈线不等长补偿方法，其特征在于，应用具有至少两个信号链路的电磁波传感器的天线阵列中，所述方法包括：
    将所述至少两个信号链路中馈线最短的作为基准链路，获取剩余各发射链路相对于该基准链路的时延差；以及
    基于所述时延差在数字域对所述天线阵列进行馈线不等长的补偿。
  27. 一种集成电路，其特征在于，包括依次连接的射频模块、模拟信号处理模块和数字信号处理模块；所述射频模块用于产生射频发射信号和接收射频接收信号；所述模拟信号处理模块用于对所述射频接收信号进行降频处理以得到中频信号；所述数字信号处理模块用于对所述中频信号进行模数转换以得到数字信号；以及
    所述射频模块包括如权利要求1至11中任一项所述的信号发射链路、如权利要求12至14中任一项所述的信号收发链路、如权利要求15至22中任一项所述的信号校准链路、如权利要求23至25中任一项所述的信号补偿链路；和/或
    所述数字信号处理模块基于权利要求26中所述的馈线不等长补偿方法在数字域进行补偿。
  28. 根据权利要求27所述的集成电路，其特征在于，还包括数据处理模块，所述数据处理模块用于对所述数字信号进行处理以实现目标探测和/或无线通信。
  29. 根据权利要求27或28所述的集成电路，其特征在于，所述集成电路为毫米波芯片。
  30. 根据权利要求27或28所述的集成电路，其特征在于，所述射频接收信号为所述射频发射信号被目标发射和/或散射所形成的回波信号，所述集成电路为传感器芯片。
  31. 一种电磁波传感器，其特征在于，包括：
    承载体；
    如权利要求27至30中任一项所述的集成电路，设置在所处承载体上；
    天线，设置在所述承载体上，或者所述天线与所述集成电路集成为一体器件设置在所述承载体上；
    其中，所述集成电路与所述天线连接，用于发射所述射频发射信号和/或接收所述射频接收信号。
  32. 一种设备，其特征在于，包括：
    设备本体；以及
    设置于所述设备本体上的如权利要求31所述的电磁波传感器；
    其中，所述电磁波传感器用于目标检测和/或通信，以向所述设备本体的运行提供参考信息。
  33. 一种非瞬时性计算机可读存储介质，其上存储有计算机可读指令，当所述指令被处理器执行时，使得所述处理器执行如权利要求26所述的方法。
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