JPS6236469B2 - - Google Patents

Description

DETAILED DESCRIPTION OF THE INVENTION The present invention relates to a constant voltage power supply device using a series resonant DC-DC converter that stabilizes the output DC voltage.

Conventional switching regulators generally use the PWM method, which stabilizes the output DC voltage by controlling the duty ratio of the on/off operation of the switching element. However, the disadvantage of the above method is that there is a period in which the direct current and voltage change sharply when the switching element is turned on and off, resulting in large switching losses and large unnecessary radiation noise. Therefore,
If we consider the above switching regulator as a power supply for audio equipment, we would insert a filter in the input/output section to greatly attenuate unnecessary radiation noise, and
Since noise countermeasures such as a completely sealed shield are required, there are problems such as increased costs and decreased reliability.

As a means to solve the above drawbacks, a series resonant DC-DC converter using a series resonant circuit composed of a capacitor and a coil has been proposed.
In this series resonant DC-DC converter, due to the series resonant circuit, the current waveform when the switching element is conducting becomes a sine wave, and the current and voltage intersect at zero when the switching element is turned on and off. Therefore, switching loss and unnecessary radiation noise are significantly reduced. However, in the above series resonant DC-DC converter, it is difficult to control input/output fluctuations to stabilize the output DC voltage without impairing the above characteristics.

Based on the above points, the circuit configuration and operation of a conventionally used series resonant DC-DC converter will be explained.

FIG. 1 is a basic circuit configuration diagram of a conventional series resonant DC-DC converter, and FIGS. 2a, b, and c are its operating waveform diagrams. In FIG. 1, switching elements (for example, transistors, thyristors, etc.) 3 and 4 that perform on/off operations are connected in series between both terminals of two input DC power supplies 1 and 2 connected in series. A resonance capacitor 7 and a primary winding 5 of a conversion transformer 5 are connected in series between the input DC power supplies 1 and 2 and the midpoints of the switching elements 3 and 4.
A is connected. Further, a rectifier circuit 8 and a smoothing capacitor 9 are connected to the secondary winding 5b of the conversion transformer 5, and a load 10 is connected to its output terminals a and b. Here, it is assumed that the resonant current flowing through the series resonant circuit composed of the resonant capacitor 7 and the effective leakage inductance of the conversion transformer 5 is i, and the charging voltage of the resonant capacitor 7 is _Vc . FIGS. 2a, b, and c show the on/off states of the switching elements 3 and 4, and operational waveform diagrams of the resonance current i and charging voltage _Vc . In FIG. ₂ , the switching element 3 is turned on at time t1, and the resonant current i is a sinusoidal current whose period is determined by the capacitance of the resonant capacitor 7 and the effective leakage inductance between time _t1 and time _t2 . becomes. During the above period, the charging voltage V _c changes from the initial charging voltage -V _c1 to V _c1 due to the resonance current i.

Next, at time _t2 , switching element 3 is turned off. During time t ₂ ≦ t ≦ t ₃ , both switching elements 3 and 4 are off, so the resonant current i becomes zero, and the charging voltage V _c1 of the resonant capacitor 7 also remains constant because there is no discharge path. . time t ₃
Then, when switching element 4 is turned on, time t ₁
The operating waveform is opposite in polarity to the operating waveform during ≦t≦t ₃ , and thereafter, when the switching element 3 is turned on again, the same operation is repeated. In addition, the resonant current i is
It is transmitted to the secondary side via the conversion transformer 5, and after rectification and smoothing, it is applied to the load 10 as a predetermined output DC voltage.
is supplied to

The above is the circuit configuration and operation of a conventional series resonant DC-DC converter. Above series resonant DC
- As means for controlling the DC converter, it has been proposed to control the capacitance value of the resonance capacitor 7 and to control the period during which both the switching elements 3 and 4 are in the OFF state, that is, the switching frequency. However, in either case, control was difficult because the amount of resonance current i did not change.

The present invention aims to provide a constant voltage power supply device that stabilizes the output DC voltage over a wide range of input/output fluctuations without impairing the characteristics of the resonance system of the above-mentioned series resonant DC-DC converter. It is something to do.

The present invention will be described below, but first a control transformer having a variable inductance function used in the present invention will be described. Figure 3 is a schematic configuration diagram showing an example, Figure 4 is its characteristic diagram, and Figure 5 is a diagram showing its characteristics.
The figure shows the equivalent symbols. Third
In the figure, AC windings N _a , N _b , N _c , N _d are provided on each leg of a combination of an E-type core and an I-type core, or a combination of two E-type cores, and a DC winding is provided on the central leg. A winding N _e is provided, and a DC current source I is connected between control terminals E and F of the DC winding N _e . Further, A and B are input terminals, and C and D are output terminals. The AC windings N _a and N _b are connected in series to form the first winding, and are wound in such a way that the magnetic flux induced in the center leg by the AC current from the input terminals A and B is canceled out. do. In other words, AC windings N _a , N _b
This is a state in which the magnetic fluxes φ ₂ and φ′ ₂ induced by the magnetic flux φ 2 and φ′ 2 are equal. Furthermore, the AC windings N _c and N _d are connected in series to form a second winding and are connected to the output terminals C and D, and a certain constant voltage is applied to the AC windings N _a and N _b . It is formed with a turns ratio.

Here, by flowing a DC current from the DC current source I, a magnetic flux φ ₁ is generated in the DC winding _Ne , and the inductance between the input terminals A and B changes. Figure 4 shows the change in inductance between input terminals A and B due to direct current. Therefore, the inductance between the input terminals A and B can be controlled in a decreasing direction by applying a direct current between the control terminals E and F. Equivalent symbols for the control transformer described above are shown in FIG. 5, and embodiments of the present invention using this will be described below with reference to FIG. 6 and subsequent drawings. FIG. 6 is a circuit diagram of the first embodiment of the present invention, in which components having the same functions as those explained in FIG. 1 are given the same reference numerals. Further, FIGS. 7a, b, c, d, e, and f are operational waveform diagrams in FIG. 6.

In FIG. 6, 11 is a control transformer as illustrated in FIG. 3, 12 is a rectifier circuit, 13 is a comparison circuit, 14 is a DC current control circuit, and 15 is a distribution circuit. Input terminal A of the control transformer 11,
B is the output terminal C,
D is connected to the rectifier circuit 12, and control terminals E and F are connected to the output terminal of the DC current control circuit 14. In addition, the output side of the rectifier circuit 12 is DC-DC
It is connected to output terminals a and b of the converter.
The comparator circuit 13 is supplied with the DC output voltage obtained at the output terminals a and b and a predetermined reference voltage E _s to each input terminal thereof, compares the respective voltage values, and calculates the difference. DC current control circuit 1
4 and the distribution circuit 15. The DC current control circuit 14 supplies the control transformer 11 with a DC current having a magnitude corresponding to the output signal from the comparison circuit 13 to the control terminals E and F of the control transformer 11.
This device changes the inductance between the first input terminals A and B (first winding), and constitutes the first control means. The distribution circuit 15 generates a frequency-modulated pulse train according to the magnitude of the output signal of the comparison circuit 13, and supplies this to the switching elements 3 and 4 to turn them on and off alternately. It constitutes a control means.

Next, the operation of the embodiment shown in FIG. 6 will be explained with reference to the operational waveform diagram shown in FIG. 7. Note that the explanation of the operation of the overlapping portions of the conventional example shown in FIG. 1 will be omitted. Resonant current i flowing in the resonant circuit
The period is determined by the effective leakage inductance of the conversion transformer 5, the capacitance of the resonance capacitor 7, and the inductance between the input terminals A and B of the control transformer 11. Further, the control current i _L1 flowing between the input terminals A and B of the control transformer 11 becomes a continuous sine wave, and is synchronized with the resonance current i.
Further, the charging voltage V _c of the resonance capacitor 7 increases in proportion to the sum of the resonance current i and the control current i _L . The above state is from time _t1 when switching element 3 is turned on to time _t'1 . At time t' ₁ , between the output terminals C and D of the control transformer 11 (second winding)
The voltage induced in The charging energy is transmitted to the output terminals a and b of the battery. This state is from time t' ₁ to t' ₂ . Further, the output current i _L2 of the control transformer 11 flowing during the above period becomes a sine wave, and its period is determined by the capacitance of the resonance capacitor 7 and the effective leakage inductance of the control transformer 11. Due to the above phenomenon, the charging voltage V _c decreases to V _c4 due to the flow of the output current i _L2 of the control transformer 11.
(time t′ ₂ ), and the next switching element 4
The control current i _L1 further decreases to V _c2 until time t ₃ when the voltage is turned on. Thereafter, the phenomenon of polarity reversal is repeated from time _t3 to time _t5 , and after the switching element 3 is turned on at time _t5 , the operating waveform becomes the same as that from time _t1 . Furthermore, since the factor that determines the amount of resonant current i is the initial charging voltage V _c of the resonant capacitor 7 , by changing the value of the initial charging voltage V _c , it is possible to
The amount of current transmitted to the next side changes, and the energy supplied to the output terminals a and b of the DC-DC converter can be controlled. The initial charging voltage V _c is
-V _c2 at time t ₁ and V _c2 at time t ₃ in FIG. The initial charging voltage (-V _c2 , V _c2 ) is related to the output current i _L2 of the control transformer 11 as described above, and the amount of output current i _L2 of the control transformer 11 is also related to the control current i _L1 . . Therefore, in order to change the output DC voltages obtained at the output terminals a and b, a control configuration that changes the amount of the control current i _L1 may be used.
In other words, the amount of control current i _L1 is inversely proportional to the inductance between input terminals A and B of the control transformer 11,
The present invention utilizes the fact that the switching frequencies of the switching elements 3 and 4 are also inversely proportional.

FIG. 8 is a circuit configuration diagram of a second embodiment of the present invention,
FIG. 9 is an operational waveform diagram of each part. In FIG. 8, parts having the same functions as those explained in FIG. 6 are given the same reference numerals. The difference between the circuit in FIG. 8 and the circuit in FIG. 6 is that the diodes 16 and 17 are replaced by switching elements 3 and 4.
The switching elements 3 and 4 are connected in parallel so as to conduct in a direction opposite to the conduction direction of the switching elements 3 and 4, that is, to be reverse biased with respect to the input DC power supplies 1 and 2. This embodiment is basically the same as the operating principle of the first embodiment shown in FIG. 1 or 2 as regenerative energy. Its operation will be explained with reference to FIG. The operation in which the output current i _L2 from the control transformer 11 flows is the same as that described in the first embodiment shown in FIG. 6, and therefore the description thereof will be omitted here. At time _t2 when the switching element 3 is turned off, the charging energy stored in the resonance capacitor 7 is transferred to the conversion transformer 5 and the diode 1.
6 to the input DC power supply 1 as a regenerative current. The above phenomenon occurs during the period from time _t2 to time t _{″ 2} in the waveform diagram of the resonant current i shown in FIG. The time t ₃
At this time, the charging voltage V _c5 of the resonance capacitor 7 changes greatly. Thereafter, the same phenomenon will be repeated.
The control operation by the control means is exactly the same as in the first embodiment shown in FIG.

A third embodiment of the invention is shown in FIG. This third embodiment further expands the control range for input/output fluctuations in the first embodiment shown in FIG. Components having similar functions are given the same reference numerals. In Figure 10,
Diodes 18 and 19 and regeneration coils 20 and 21 are connected in series. In this case, the diodes 18 and 19 and the regenerative coil 2
0 and 21 are connected in parallel to the series connection circuit of the switching elements 3 and 4 and the primary winding 5a of the conversion transformer 5, respectively, and are connected to the diodes 18 and 21, respectively.
19 are connected in a direction opposite to the conduction direction of the switching elements 3 and 4, that is, in a reverse biased manner with respect to the input DC power supplies 1 and 2.

This embodiment still utilizes the regenerative current from the resonant capacitor 7 described in the second embodiment (FIG. 8), but the regenerative current does not pass through the conversion transformer 5. Diode and regenerative coil connected in series (18 and 20 or 19 and 2)
1), it is regenerated to the input DC power supply 1 or 2, so it does not become the output energy of the DC-DC converter. Furthermore, regenerative coil 2
By changing the inductance of 0 or 21,
It is also possible to arbitrarily change the period of the regenerative current. The control operation by the control means is completely the same as in the first embodiment.

Although the above embodiments of the present invention have been described using a half-bridge configuration using two switching elements, similar effects can be obtained even when a full-bridge configuration using four switching elements is used. Furthermore, although the effective leakage inductance of the conversion transformer 5 was used as the inductance of the series resonant circuit in each of the above embodiments, another resonance coil may be inserted in series between the resonance capacitor and the primary winding of the conversion transformer. , it is also possible to configure it using the inductance of the resonance coil.
Furthermore, the control transformer having a variable inductance function used in the present invention is not limited to the configuration shown in FIG. Any wire can be used as long as it can vary the inductance of the first winding according to the conditions.

As described above, according to the present invention, the output DC voltage can be stabilized over a wide range of input/output fluctuations with a simple circuit configuration while taking advantage of the features of the series resonant DC-DC converter. , its industrial value is extremely high.

[Brief explanation of the drawing]

Fig. 1 is a circuit configuration diagram of a conventional series resonant DC-DC converter, Fig. 2 a, b, and c are its operating waveform diagrams, and Fig. 3 is a schematic diagram showing an example of the configuration of a control transformer used in the present invention. , FIG. 4 is its characteristic diagram, FIG. 5 is its equivalent symbol diagram, FIG. 6 is a circuit configuration diagram of the first embodiment of the present invention, and FIG. 7 is a, b, c, d, e, f.
8 is a circuit configuration diagram of the second embodiment of the present invention, and FIG. 9 is a, b, c, d, e,
f is its operating waveform diagram, and FIG. 10 is a circuit configuration diagram of a third embodiment of the present invention. 1, 2... Input DC power supply, 3, 4... Switching element, 5... Conversion transformer, 5a... Primary winding, 5b
... Secondary winding, 7... Resonance capacitor, 8... Rectifier circuit, 9... Smoothing capacitor, 10... Load, 11...
Control transformer, 12... Rectifier circuit, 13... Comparison circuit, 14... DC current control circuit, 15... Distribution circuit, 16, 17, 18, 19... Diode, 2
0, 21... Regenerative coil, a, b... Output terminal.

Claims (1)

[Claims] 1. At least a switching element and a conversion transformer that operate on and off with respect to an input DC power source.
A series connection circuit including a secondary winding and a resonant capacitor is connected, and a first rectifier circuit and a smoothing circuit are connected to the secondary winding of the conversion transformer to obtain a DC output voltage at an output terminal. DC
- a DC converter, a first winding connected in parallel to the resonance capacitor, and a second winding for output extraction; and the inductance of the first winding is controlled by the supplied electric signal a control transformer capable of changing the voltage, and a signal voltage obtained from a second winding of the control transformer rectified to
a second rectifier circuit supplying the output terminal of the DC-DC converter; a first control means for controlling the inductance of the control transformer as a function of the DC output voltage available at the output terminal of the DC-DC converter; A constant voltage power supply device comprising second control means for controlling the switching frequency (or period) of the switching element as a function of the DC output voltage obtained at the output terminal of the DC-DC converter. 2. A constant voltage power supply device according to claim 1, characterized in that a unidirectional element is connected in parallel to the switching element so as to conduct in a direction opposite to the conduction direction of the switching element. 3 In the statement of claim 1, a circuit in which a coil and a unidirectional element that conducts in a direction opposite to the conduction direction of the switching element are connected in series is defined as a circuit in which a coil and a unidirectional element that conducts in a direction opposite to the conduction direction of the switching element are connected in series.
A constant voltage power supply device characterized in that it is connected in parallel to the series connection circuit of the next winding.